A Full Ka-Band CMOS Amplifier Using Inductive Neutralization with a Flat Gain of 13 ± 0.2 dB

Abstract

This paper presents a CMOS wideband amplifier operating in the full Ka-band, with a low gain variation. An inductive neutralization is applied to the amplifier to compensate for the gain roll-off in the high-frequency region. Neutralization inductance is carefully determined considering the tradeoff between stability and gain. To achieve a low gain variation over the full Ka-band, the amplifier employs the frequency staggering technique in which impedance matching for three gain stages is performed at different frequencies of 26, 34, and 42 GHz. The experimental results show that a peak gain of 13.2 dB is achieved at 39.2 GHz. The 3 dB bandwidth is from 23.5 to 41.7 GHz, which fully covers the Ka-band. Especially, the gain ripple of the amplifier is only 13 ± 0.2 dB over a wide bandwidth from 26.2 to 40.2 GHz. The input and output return loss values are better than −10 dB from 26.3 to 40.1 GHz and from 25.3 to 50 GHz, respectively. The DC power consumption is 18.6 mW.

1. Introduction

Owing to the demand for high-speed data transfer, 5G communication based on the millimeter wave has been actively developed. The global spectrum allocated for the 5G millimeter wave is mainly distributed in the Ka-band (26.5–40 GHz) [1,2]. Therefore, a wideband flat-gain amplifier operating in the full Ka-band is needed for implementing multistandard or ultrawideband radio systems supporting the global spectrum.

Conventionally, wideband amplifiers operating at several gigahertz have been implemented using a resistive feedback technique [3,4,5]. The resistive feedback makes the input impedance more resistive, thereby allowing wideband input matching and enhancing the linearity [6]. However, a disadvantage of the resistive feedback is that the gain decreases in the upper end of the bandwidth [7]. Therefore, multiple amplifying stages must be cascaded to compensate for the gain roll-off, which increases the overall DC power consumption. Recently, a feedback technique based on a transformer balun was proposed to achieve a wideband gain response with a compact size [8,9]. However, the complicated structure of the transformer balun makes it challenging to optimize the transformer parameters such as the primary and secondary inductances and the coupling coefficient to achieve both the wideband input matching and flat-gain response at the same time.

This paper presents a full Ka-band amplifier with a low gain variation. In the first gain stage, the inductive neutralization is adopted to compensate for the gain roll-off in the upper end of the Ka-band, while the frequency staggering is employed in the subsequent gain stages for wideband and flat-gain responses. In Section 2, the analysis and design of the amplifier are presented. The experimental results are presented in Section 3. Finally, the conclusions are drawn in Section 4.

2. Circuit Analysis and Design

Figure 1 shows a schematic of the full Ka-band flat-gain CMOS amplifier using the inductive neutralization and frequency staggering techniques. Three common-source (CS) gain stages (M₁, M₂, and M₃) were cascaded to secure a sufficient gain. To accomplish a wideband flat-gain response over the full Ka-band, the inductive neutralization was applied to the first stage, while the frequency staggering was employed in the following interstage and output matching networks.

Figure 1. Schematic of the full Ka-band amplifier with flat gain.

2.1. Inductive Neutralization Technique

To achieve a high flat gain, the optimum transistor topology was determined first. Figure 2 shows the maximum stable gain (MSG) (or the maximum available gain (MAG)) [10] of various transistor topologies in the given 65 nm CMOS technology.

Figure 2. Comparison of MSG (or MAG) of different transistor topologies in a 65 nm CMOS technology.

For a fair comparison, the simulation conditions were set identical: the gate width and drain bias current of each transistor are 30 μm and 3.1 mA, respectively. Among the CS, common-gate (CG), and cascade topologies, the CS topology provides the highest gain beyond 38 GHz. However, the gain of the CS still gradually decreases as the frequency increases further than 38 GHz. To achieve a flat gain over the full Ka-band, a gain-boosting technique should be used to compensate for the gain roll-off at the upper end of the Ka-band.

In this research, we adopted the inductive neutralization technique in the CS to boost the gain. It is well known that the gate-drain capacitance C_gd degrades the forward gain of the transistor [11]. Therefore, by connecting a neutralization inductor (L_neut) between the gate and drain, C_gd is resonated with the inductor, and thus the MSG (or MAG) can be boosted [12,13]. In Figure 2, the MSG (or MAG) of the CS topology with the inductive neutralization (L_neut = 1.3 nH) is shown, which is higher than that of other topologies beyond 31 GHz. Moreover, the inductive neutralization changes the behavior of the MSG (or MAG) over the frequency. It is observed in Figure 2 that the peak of the MSG (or MAG) is pushed to a higher frequency region and, thus, can be located almost at the upper end of the Ka-band. This offsets the high loss of passive matching components and the resulting gain roll-off at the high frequencies.

While boosting the gain by resonating out the feedback capacitance C_gd, the inductive neutralization may disrupt the stability of the transistor. This is because positive feedback can be reinforced when the C_gd–L_neut network operates far from the resonance. Therefore, the value of L_neut should be carefully determined considering a tradeoff between the gain and the stability. Figure 3 shows the simulated MSG (or MAG) and stability factor K at 39 GHz versus L_neut. The MSG (or MAG) has two peaks at L_neut = 1.23 and 1.44 nH, respectively. On the other hand, the K has a single peak at L_neut = 1.3 nH and decreases rapidly as L_neut deviates from 1.3 nH. The K becomes even below unity when L_neut < 1.23 nH or L_neut > 1.44 nH. Hence, we chose L_neut of 1.3 nH, which results in MAG of 15.1 dB and K of 11.2 at 39 GHz. Notably, both MAG and K are boosted by inductive neutralization. The inductive neutralization with L_neut of 1.3 nH was applied to the first gain stage (M₁), which presents a high gain at the upper end of the Ka-band. It should be noted that the second and third gain stages (M₂ and M₃) have no inductive neutralization because they should present a high gain at lower frequencies, as described in Section 2.2.

Figure 3. Simulated MSG (or MAG) and stability factor K at 39 GHz versus L_neut.

The input matching of the amplifier should operate in a wide bandwidth to suppress mismatch loss that occurs in the interconnection with other circuit blocks. The inductive neutralization of the first gain stage alters the input impedance of M₁; therefore, it should be considered in the input matching design shown in Figure 4. Specifically, the C_gd–L_neut network brings the load impedance of M₁ (Z_L1) to the input of M₁ (Z_in1) when the operation is off the resonance frequency. Z_L1 includes the drain bias network of M₁ (L₃ and C_bypass), the input impedance looking into M₂, and the interstage matching between M₁ and M₂. In Figure 5, the impedances of the input matching network are shown as the frequency is swept from 26 to 40 GHz. The Z_in1 with no Z_L1 and L_neut included is dominantly determined by the gate capacitance and resistance, as expected, shown in line (a). However, Z_in1 with Z_L1 included, shown in line (b), and with both Z_L1 and L_neut included, shown in line (c), are substantially different from line (a). Fortunately, L_nuet induces a resonance behavior in Z_in1, thereby providing a wideband characteristic to the input reflection coefficient. Finally, the input impedance at the port (Z_in) is matched to 50 Ω with L₁, C₁, and L₂, shown in line (d).

Figure 4. Input matching network of the first gain stage.

Figure 5. Loci of the input impedances from 26 to 40 GHz.

2.2. Frequency Staggering Technique

To achieve a flat gain over a wide frequency range, the center frequencies of three gain stages (M₁, M₂, and M₃) are intentionally staggered. Through the interstage and output impedance matching described below, the peak-gain frequencies of the first, second, and third stages are tuned to 42, 25.5, and 30.5 GHz, respectively, as shown in Figure 6.

Figure 6. Gain of the three individual stages and the whole amplifier.

The ac equivalent circuit of the amplifier is drawn in Figure 7a. For the first interstage matching between M₁ and M₂, the impedances seen looking into the output of M₁ (Z_out1) and the input of M₂ (Z_in2) are simulated as Z_out1 = 48.3 − j89.2 Ω and Z_in2 = 14.2 − j71.9 Ω at 42 GHz. For conjugate matching, Z_in2 is transformed to Z_out1^* by an L-section network consisting of L₃, L₄, and TL₁. As shown in Figure 7b, the series inductive components (L₄ and TL₁) bring Z_in2 to Z_in2,1, which is located on a constant conductance circle of Re{1/Z_out1^*}. Then, a shunt inductor (L₃) transforms Z_in2,1 to Z_in2,2, which completes the conjugate matching to Z_out1^*. It should be noted that the value of L₃ determines the frequency at which the conjugate matching is satisfied. In Figure 7b, the locus of Z_in2,2 at 42 GHz is depicted as L₃ decreases from infinity to 0.1 nH. According to the plot, L₃ is chosen as 0.8 nH to achieve conjugate matching at 42 GHz. The final matching point of Z_in2,2 slightly deviates from Z_out1^* after fine optimization of the whole amplifier. The conjugate interstage matching between M₁ and M₂ at 42 GHz results in the gain peak of the first gain stage at the same frequency as that shown in Figure 6.

Figure 7. Interstage matching: (a) AC equivalent circuit of the amplifier; (b) first interstage matching at 42 GHz; (c) second interstage matching at 26 GHz.

The second interstage matching between M₂ and M₃ is performed in a similar way to the first interstage matching. However, the matching frequency is intentionally changed to 26 GHz. As shown in Figure 7c, the input impedance of M₃ (Z_in3) is transformed to the conjugate of the output impedance of M₂ (Z_out2^*). The center frequency of the matching is tuned to 26 GHz by choosing the value of a shunt inductor (L₅) as 0.3 nH. It is noted that the second interstage matching is supposed to have the matching performance sensitive to the adjacent stages more than the first interstage matching. This is because the adjacent impedances are seen through C_gd of M₂ and M₃, while C_gd of M₁ is diminished by the neutralization inductor L_neut. Therefore, an extra shunt inductor (L₆) is added as an additional degree of design freedom in the second interstage matching while minimizing the influence on the adjacent matching. The second interstage matching results in the gain peak of the second stage at 25.5 GHz, as observed in Figure 6.

The output matching is fulfilled by L₈ and L₉ at 34 GHz, which leads to the peak gain of the final stage at 30.5 GHz shown in Figure 6. It is noted that there is a slight difference between the matching frequency and the peak gain frequency of each individual stage. This is because the gain and bandwidth of the whole amplifier were re-optimized after combining the individual stages.

Through the frequency staggering, the overall gain of the whole amplifier exhibits a flat response of 12.8 ± 0.5 dB from 25.6 to 42.4 GHz, covering the full Ka-band, as shown in Figure 6. After full electromagnetic simulation with Keysight ADS Momentum, the values of the matching components are slightly reoptimized considering the undesired magnetic and electric coupling between matching components and interconnection lines. The design parameter values of the full Ka-band amplifier are listed in Table 1.

Table 1. Design parameter values of the full Ka-band amplifier.

Parameters	Values	Parameters	Values
L1	0.6 nH	C1	0.41 pF
L2	0.68 nH	R1	200 Ω
L3	0.8 nH	R2	1000 Ω
L4	0.45 nH	TL1	265 μm
L5	0.3 nH	TL2	120 μm
L6	0.7 nH	M1–M3	1 × 30 μm
L7	0.2 nH	Vg1	1.1 V
L8	0.65 nH	Vg2	1.1 V
L9	0.27 nH	Vg3	0.9 V
Lneut	1.3 nH	Vd1	1.1 V
Cbypass	1.36 pF	Vd2	1.1 V
Cdc	1.36 pF	Vd3	1.1 V

3. Experimental Results

The full Ka-band amplifier was fabricated using a 65 nm CMOS technology. The chip micrograph is shown in Figure 8. The whole chip area including the probing pads and the core chip area were 1 × 0.7 and 0.5 × 0.75 ${\mathrm{mm}}^2$, respectively.

Figure 8. Chip micrograph of the full Ka-band amplifier.

The measured S-parameters are shown in Figure 9. A peak gain of 13.2 dB was measured at 39.2 GHz. The 3 dB bandwidth was 18.2 GHz from 23.5 to 41.7 GHz, which covered the entire Ka-band frequencies. Specifically, from 26.2 to 40.2 GHz, the gain exhibited a flat response with a small variation, i.e., 13 ± 0.2 dB. The measured |S₁₁| and |S₂₂| were smaller than −10 dB from 26.3 to 40.1 GHz. The stability K-factor was calculated from the measured S-parameters, as shown in Figure 10. It was observed that the Ka-band amplifier was stable over the frequencies from DC to 50 GHz, which include both the operation and the out-of-band frequencies. The measured group delay is shown in Figure 11. The group delay was fairly flat in the frequency range between 22 and 46 GHz. The power measurement of the amplifier at 34 GHz is shown in Figure 12. The input 1 dB compression point (IP1dB) was −13.2 dBm. Figure 13 shows the measured IP1dB over the frequency. The IP1dB ranged from −13.2 to −16.6 dBm between 26 and 40 GHz.

Figure 9. Measured S-parameters of the full Ka-band amplifier.

Figure 10. Stability factor based on the measured S-parameters.

Figure 11. Measured group delay.

Figure 12. Measured output power and gain versus input power at 34 GHz.

Figure 13. Measured IP1dB versus frequency.

In Table 2, the measured performances are summarized and compared with those of the previously reported CMOS Ka-band amplifiers. The amplifier in this study exhibited a superior gain flatness in the full Ka-band frequencies to the other state-of-the-art amplifiers.

Table 2. Performance comparison with previous CMOS-based Ka-band amplifiers.

Ref.	Technology	Peak Gain (dB)	|S21|3-dB BW (GHz)	|S11|−10 dB BW (GHz)	Gain Flatness over Frequency	IP1dB (dBm)	DC Power Consumption (mW)	Core Chip Area ** $\left(\mathbf{m}{\mathbf{m}}^2\right)$
This work	65-nm CMOS	13.2	18.2 (23.5–41.7)	13.8 (26.3–40.1)	13 $\pm$ 0.2 dB over 26.2–40.2 GHz	−13.2	18.6	0.37
[8]	65-nm CMOS	13.5	23.4 (19.2–42.6)	22.1 (16.8–38.9)	12.65 $\pm$ 0.85 dB over 21–41 GHz *	−9.8	6.36	0.13
[14]	65-nm CMOS	22.1	8.5 (24–32.5)	10 (30–40 *)	20.6 $\pm$ 1.5 dB over 24–32.5 GHz	−19	19.3	0.12
[15]	22 nm SOI CMOS	12.6	17 (23–40)	6 (25–31 *)	20.6 $\pm$ 1.5 dB over 24–32.5 GHz	−6	13	0.12
[16]	22 nm SOI CMOS	18.2	19 (24–43)	24 (20–44 *)	18 $\pm$ 1.1 dB over 24–29 GHz	−20.4	12.1	0.21
[17]	28-nm CMOS	18.6	4.3 (28.2–32.5 *)	7 (30–37)	17.1 $\pm$ 1.5 dB over 28.2–32.5 GHz	−25	9.7	0.23
[18]	40-nm CMOS	18.4	9.3 (25–34.3 *)	14 (26–40)	16.9 $\pm$ 1.5 dB over 25–34.3 GHz *	−13.4	21.5	0.26
[19]	45 nm SOI CMOS	20	20 (27–47.5 *)	19 (27–48)	19 $\pm$ 1.5 dB over 27–47.5 GHz	−19	58	0.2
[20]	90 nm SOI CMOS	13.8	15 (29–44)	21 (29–50 *)	12.3 $\pm$ 1.5 dB over 29–44 GHz	−10	18	0.48 ***

* Estimated from a plot of the article; ** excluding probing pads; *** including probing pads.

4. Conclusions

In this research, a full Ka-band amplifier with a flat-gain response was fabricated using a 65 nm CMOS technology. To compensate for the gain roll-off in the upper end of the bandwidth, an inductive neutralization technique was used in the first stage. Neutralization inductance was carefully determined by considering gain and stability. To achieve a wideband flat gain, three CS stages designed at different center frequencies of 26, 34, and 42 GHz were cascaded. Using the frequency staggering technique, we found that the whole amplifier exhibited a flat-gain response of 13 ± 0.2 dB over the full Ka-band frequencies.