Full-Bridge T-Type Three-Level LLC Resonant Converter with Wide Output Voltage Range

Abstract

Traditional LLC resonant converters face significant challenges in wide-output-voltage-applications, such as limited voltage gain, efficiency degradation under wide-gain range, and increased complexity in magnetic component design. For example, in electric vehicle charging power modules, achieving wide output voltage typically relies on changing the transformer turns ratio or switching the series-parallel circuit configuration via relays, which prevents real-time dynamic adjustment. To overcome these limitations, this paper proposes a wide-gain-range control method based on a full-bridge T-type three-level LLC resonant converter, capable of achieving a voltage gain range exceeding six times. By integrating a T-type three-level bridge arm with PWM modulation and employing a variable-topology and variable-frequency control strategy, the proposed method achieves synergistic optimization for wide-output-voltage-applications. PWM modulation enables wide-range voltage output by dynamically adjusting both the converter topology and switching frequency. Finally, the proposed method is validated through circuit simulations and experimental results based on a full-bridge T-type three-level LLC converter prototype, demonstrating its effectiveness and feasibility.

1. Introduction

As a highly efficient power conversion scheme, the LLC resonant converter is widely used in various types of power electronic devices, particularly in medium- and high-power applications that require high efficiency, low loss, and low electromagnetic interference (EMI). Its most significant advantage lies in the implementation of soft-switching operation, which enables the converter to maintain zero-voltage-switching (ZVS) or zero-current-switching (ZCS) states during both conduction and turn-off of the switching devices. This capability effectively reduces switching losses and electromagnetic interference [1,2,3]. The soft-switching feature allows the LLC resonant converter to maintain high efficiency even at high frequencies, making it especially suitable for high-frequency, high-power-density applications such as high-efficiency power supplies and electric vehicle charging systems [4,5].

However, the frequency regulation range of LLC resonant converters is typically narrow. When operating far from the resonant frequency, conventional adjustments to the duty cycle or phase shift angle may fail to maintain soft-switching, leading to increased switching currents in the power devices and a significant drop in efficiency [6,7]. This limitation poses challenges for achieving wide output voltage ranges, especially under large load variations, where system efficiency can degrade substantially [8].

Many existing wide gain control methods primarily rely on adjusting the gain near the resonance point [9,10], but this approach limits the converter’s output voltage range to some extent. To address this issue, researchers have proposed various topology design schemes, with full-bridge LLC and half-bridge LLC being the most common structures. These topologies achieve a wider output voltage range within a certain range by adjusting the frequency, duty cycle, and phase shift angle [11,12,13,14,15]. However, they still have a limited gain range and pose significant design challenges under efficient operating conditions [16,17].

To address this issue, multilevel topologies have gained increasing attention in recent years. By increasing the number of levels, multilevel topologies can significantly reduce the voltage stress on the switching devices and provide a stable output across a wider voltage range [18,19,20]. For instance, converters with three or more levels can not only extend the gain range but also enhance both the efficiency and power density of the system [21,22,23]. However, the adoption of multilevel topologies often comes with more complex control strategies and higher cost burdens, particularly in high-power applications [24,25]. Additionally, adjusting the transformer turns ratio is a common method to extend the gain range. By altering the turns ratio or modifying the topology, the voltage gain can be adjusted within a specific range, thus enabling a wider output voltage [26,27,28,29,30]. However, this approach is constrained by factors such as transformer size, cost, and magnetic materials, making it unsuitable for all application scenarios [31].

Another three-level topology is the T-type three-level LLC converter. This topology typically includes four switching tubes, though only three are required when the T-type bridge is replaced by a bidirectional switch, which helps achieve low conduction losses and reduced switching stress [32]. The T-type bridge has also evolved into a hybrid bridge topology, which meets the wide gain requirement through frequency shifting [33]. Furthermore, recent research on the T-type three-level LLC converter has focused on widening the gain range of the T-type bridge LLC converter by achieving a 0.25-fold gain through alternating modulation of the topology [34,35,36]. Reference [34] indicates that the T-type bridge structure exhibits lower on-resistance and voltage drop compared to stacked half-bridge and NPC three-level LLC converters [37,38]. As illustrated in Figure 1, the switch network of the T-type bridge demonstrates reduced conduction resistance and voltage drop. Specifically, for power semiconductor devices with the same current rating and technology generation, the conduction resistance of a 1200 V SiC MOSFET is lower than the combined resistance of two 650 V SiC MOSFETs connected in series. Likewise, the conduction voltage drop of a 1200 V IGBT is less than that of two 650 V IGBTs in series. These characteristics indicate that the T-type three-level LLC converter achieves lower conduction losses, thereby improving overall system efficiency [39,40,41]. Reference [42] indicate that compared with T-type LLC resonant converters, dual-resonant cavity structures are more prone to current imbalance. In addition, the use of multiple transformers further increases the complexity of design and debugging. Additionally, the T-type three-level LLC converter is more cost-effective when using three independent drive supplies. The T-type bridge can also be replaced with bidirectional switches, further reducing the overall cost.

To expand the gain of the LLC converter while utilizing the advantages of the T-type bridge, this paper proposed a wide-range control method based on a full-bridge T-type three-level LLC resonant converter. The T-type three-level bridge arm reduces the voltage stress on the switching devices and extends the voltage gain range through multilevel modulation. Meanwhile, the wide-output-voltage design, which incorporates variable-frequency and variable-topology control strategies, ensures soft-switching characteristics across the entire load range.

The main contributions of this paper are as follows:

(1)

This paper proposes a wide-range control method based on a full-bridge T-type three-level LLC resonant converter and analyzes the working principle and wide output voltage characteristic.

(2)

A variable-topology strategy is employed to tackle the challenge of achieving both a wide output voltage range.

(3)

A PWM modulation method is proposed to achieve a wide output voltage range through variable-topology and variable-frequency control strategies.

The structure of this paper is as follows: Section 2 describes in detail the conventional control methods to realize wide output voltage and their defects and shortcomings; Section 3 presents the variable-topology control strategy; Section 4 presents experimental results and comparative analyses; and Section 5 concludes the paper.

2. Conventional Control Method for Full-Bridge T-Type Three-Level LLC Resonant Converter

2.1. Operating Principle of the Full-Bridge T-Type Three-Level LLC Converter

As shown in Figure 2, the topology full-bridge T-type LLC resonant converter utilizes two interleaved half-bridge T-type three-level converters operating with a 180° phase shift between phases. Each phase contains four power switches (S_a1–S_a4), where S_a1/S_a3 and S_a2/S_a4 operate as complementary pairs with a 50% duty cycle. The voltage divider capacitors (C_dc1, C_dc2) ensure balanced voltage distribution (Vc_dc1 = Vc_dc2), while the LLC resonant tank—comprising the resonant inductor (L_r), resonant capacitor (C_r), and the transformer’s magnetizing inductance (L_m)—supports soft-switching operation. The secondary-side full-bridge rectifier (D₁–D₄), along with the output capacitor (C_o), provides a regulated DC output.

2.2. Analysis of Conventional Control Methods

Conventional control methods often achieve a wide voltage output range by varying the duty cycle at frequencies higher than the resonant frequency. For example, Figure 3 shows the modulated waveform of a wide voltage output achieved by varying the duty cycle at frequencies greater than the resonant frequency.

Phase [t₀–t₁]: During this phase, S_a1 and S_a2 of phase A conduct, while S_b3 and S_b4 of phase B conduct. The input voltage to the resonant cavity is equal to the input voltage of the system, and the resonant current is greater than the magnetizing current. The magnetizing inductance is clamped by the output voltage, causing the magnetizing current to rise linearly.

Phase [t₁–t₂]: At time t₁, S_a1 and S_b4 turn off. At this moment, the resonant current is greater than zero and is sustained by the body diodes of S_a3 and S_b2 (i.e., S_a2 and S_b3). The input voltage to the resonant cavity becomes zero, allowing for ZVS when S_a3 and S_b2 turn on. Once S_a3 and S_b2 are on, the resonant current decreases, while the magnetizing current continues to rise linearly. When the magnetizing current equals the resonant current, the magnetizing inductance and resonant capacitance jointly resonate with the resonant inductor and capacitor. At this point, the primary side of the transformer is decoupled from the secondary side, and the magnetizing inductance resonates with the resonant capacitor and resonant inductor.

Phase [t₂–t₄]: The converter enters the next half-cycle. When the voltages across C_dc1 and C_dc2 are equal, the waveform of the converter in the [t₂–t₄] phase becomes symmetrical to that of the [t₀–t₂] phase.

A wide voltage output range is often achieved by varying the switching frequency below the resonant frequency, as shown in Figure 4, which illustrates the modulation waveform at frequencies below the resonant frequency.

Phase [t₀–t₁]: At time t₀, the switching tubes S_a3 and S_b1 are turned off. Since the excitation inductance L_m is relatively large and the duration of this mode is short, it can be approximated that the resonant inductance current remains unchanged. At this point, the resonance and excitation currents are equal. When the body diodes of S_a1 and S_b4 conduct, these transistors can be turned on at zero voltage. To ensure the zero-voltage turn-on of S_a1 and S_b4, they must be switched on before the current i_Lr changes from negative to positive. As the source-drain voltages of S_a1 and S_b4 drop to zero, zero-voltage turn-on is achieved. At this point, the voltage between points A and B is V_in, and the rectifier diode D₁ conducts. The excitation inductance is then clamped at V_o/n, ceasing to participate in the resonance, while the excitation inductance current increases linearly. Meanwhile, the resonant network consisting of L_r and C_r begins to resonate, with the voltage across it being V_in − V_o/n.

Phase [t₁–t₂]: At time t₁, the current in the resonant inductor equals the excitation inductor current. At this moment, the current on the transformer secondary side is zero, effectively disconnecting the load from the resonant network. The rectifier diode D₁ is turned off, ensuring ZCS without any reverse recovery issues. The voltage across the excitation inductor L_m is no longer clamped at V_o/n, and the resonant components L_r, L_m, and C_r continue to resonate and operate.

Phase [t₂–t₄]: During this phase, the operating principle of the converter is similar to the second half of the previous operating cycle, so it will not be repeated here.

When the full-bridge T-type three-level LLC resonant converter operates at f_s > f_r, a wide voltage output range is achieved by varying the duty cycle. Additionally, when operating below the resonant frequency, a wide voltage output range is typically achieved using Pulse Frequency Modulation (PFM) converter control. The voltage gain function of the converter can be derived through fundamental approximation equivalent analysis:

$$G={\displaystyle \frac{\sin D\pi }{\sqrt{{\left(1+\frac{1}{k}\left(1-\frac{1}{f{n}^2}\right)\right)}^2+\left(Q^2{\left(fn-\frac{1}{fn}\right)}^2\right)}}}$$

(1)

where $k=L_m/L_r$, $f_n=f_s/f_r$, $f_s$ is the switching frequency, the resonant frequency is $fr=1/2\pi \sqrt{L_r{C}_r}$, the quality factor is $Q=\sqrt{L_r{C}_r}/{\mathrm{R}}_{eq}$, and $n$ is the transformer transformation ratio.

As shown in Figure 5, the traditional method achieves a wide voltage gain range by controlling the converter at frequencies both below and above the resonant frequency, primarily by varying the duty cycle. Although theoretical analysis indicates that a near-zero output voltage can be attained by reducing the duty cycle, this results in certain operating ranges where ZVS soft switching is not achievable. In these conditions, the turn-off current increases significantly, which leads to lower efficiency and greater EMI. Therefore, new methods for wide output regulation need to be explored. The 0.75× and 0.25× voltage gain modulation methods proposed in this paper offer a balance between wide voltage regulation and efficient operation. The proposed variable topology control strategy defines three discrete gain ratios (0.75, 0.5, and 0.25) at the resonant frequency. Along with the 1× fundamental ratio, this results in three distinct operating modes. Pulse Frequency Modulation (PFM) further enhances voltage regulation, and together these techniques enable a wide output voltage range.

3. Proposed Modulation and Control Method for Wide Gain

3.1. Proposed 0.75 Gain Operating Mode

The modulation method proposed in Section 2 achieves a wide output voltage range through duty ratio variation. However, at small duty ratios, excessive resonant current spikes occur, leading to efficiency degradation. To achieve a wide gain range, the full-bridge T-type LLC converter employs variable-topology modulation, enabling multiple normalized gain points at the resonant frequency. The first part of this chapter presents the variable-topology modulation scheme that achieves a 0.75-times normalized gain, where the original reference normalization is unity gain. This introduces a 0.75-times voltage gain mode for the full-bridge T-type LLC converter operating at the resonant frequency. The 0.75-times gain mode consists of two frequency-controlled sub-modes (Mode 1 and Mode 2). Assuming that the voltages across C_dc1 and C_dc2 remain balanced at 0.5 V_in with sufficient capacitance, Figure 6a,b show the equivalent circuit and operational waveforms for Mode 1, respectively.

When the converter operates in Mode 1 of the 0.75-times gain mode, S_a2 remains on, while S_a4, S_b2, and S_b3 stay off. S_a1 and S_a3 operate complementarily with a 50% duty cycle, and S_b1 and S_b4 operate complementarily with a 50% duty cycle, with a 180° phase difference between S_b1 and S_a1. The converter exhibits the following operational phases within one cycle:

Phase [t₀–t₁]: During this phase, S_a1 and S_b4 conduct. The resonant tank input voltage V_AB equals V_in, and the resonant inductor and capacitor resonate in series. The magnetizing inductance is clamped to V_o/n, causing its current to increase linearly.

Phase [t₁–t₂]: When the magnetizing current equals the resonant current, the primary and secondary sides of the transformer decouple, allowing the magnetizing inductance to resonate with the resonant inductor and capacitor.

Phase [t₂–t₃] (Dead Time): At t₂, S_a1 and S_b4 turn off. Since the resonant current remains positive, the circulating current flows through the body diodes of S_a3 and S_b1, changing the resonant tank input voltage to −0.5 V_in. With the body diodes of S_a3 and S_b1 conducting, ZVS turn-on is achieved at t₃.

Phase [t₃–t₄]: With S_a3 and S_b1 conducting, the resonant tank input voltage is −0.5 V_in. Since the resonant current is lower than the magnetizing current, the resonant inductor and capacitor continue to resonate in series. The magnetizing inductance is clamped to −V_o/n, causing its current to decrease linearly.

Phase [t₄–t₅]: When the resonant and magnetizing currents equalize, the transformer decouples again, allowing the magnetizing inductance to resonate with the resonant components.

Phase [t₅–t₆] (Dead Time): At t₅, S_a3 and S_b1 turn off. With negative resonant current, the current flows through the body diodes of S_a1 and S_b4, changing V_AB to V_in. ZVS is achieved when S_a1 and S_b4 turn on after the dead time.

Mode 2 of the 0.75-times gain mode is symmetrical to Mode 1: S_a1, S_b2, and S_b3 remain off, while S_a3 stays on. S_a2 and S_a4 operate complementarily at a 50% duty cycle, as do S_b1 and S_b4. The corresponding waveforms and equivalent circuits are shown in Figure 6b and Figure 7a, respectively. Given the high symmetry between the modes, where only C_dc2 replaces C_dc1 as the unilateral power source, the operational process of Mode 2 is omitted here.

Due to the presence of negative-voltage half-cycle intervals in 0.75-times gain Mode 1, the resonant network is powered solely by the upper input capacitor, C_dc1, during these periods. Similarly, in 0.75-times gain Mode 2, there is a duration where only the lower input capacitor, C_dc2, supplies power.

To maintain voltage balance between the input capacitors, the converter should alternate between both modes while operating in 0.75-times gain mode. This alternation prevents excessive voltage droop caused by prolonged operation of a single capacitor. Ideally, the converter operates in Mode 1 for a fixed duration (typically several dozen switching cycles in high-frequency operation), then switches to Mode 2 for an equal duration, repeating this sequence.

Thanks to the four-diode rectifier bridge on the secondary side, mode transitions have minimal impact on the output load. Consequently, the input voltage V_AB of the resonant network can be approximated as a constant 0.75 V_in during this process. In summary, the voltage-normalized gain formula for the full-bridge T-type LLC converter in 0.75-times gain mode is derived as follows:

$$G={\displaystyle \frac{0.75}{\sqrt{{\left(1+\frac{1}{k}\left(1-\frac{1}{f{n}^2}\right)\right)}^2+\left(Q^2{\left(fn-\frac{1}{fn}\right)}^2\right)}}}$$

(2)

3.2. Proposed 0.25 Gain Operating Mode

This section proposes a mode of operation for the full-bridge T-type LLC resonant converter that achieves a voltage gain of 0.25 at the resonant frequency, referred to as the 0.25-fold gain mode. This mode consists of two sub-modes, each controlled by frequency conversion. Assuming that the voltages across C_dc1 and C_dc2 are both 0.5 V_in and that the capacitance is sufficiently large, the control method for Mode 1 of the 0.25-fold gain mode is as follows: switches S_a1, S_b1, S_b2, and S_b3 remain off, while switches S_a3 and S_b4 remain on, and switches S_a2 and S_a4 operate complementarily with a 50% duty cycle. The corresponding circuit waveforms and equivalent circuits are shown in Figure 8a and Figure 8b, respectively.

Phase [t₀–t₁]: During this stage, S_a2 is conducting, and the resonant cavity input voltage (V_AB) is 0.5 V_in. Initially, the resonant inductor and resonant capacitor resonate in series, and the excitation inductance is clamped to V_o/n, with the current increasing linearly. Since f_s < f_r, the excitation inductance and resonant inductance resonate together towards the end of this stage when the excitation inductance equals the resonant inductance.

Phase [t₁–t₂]: The time interval between t₁ and t₂ is the dead time. At t₁, S_a2 turns off, and the resonant current becomes greater than zero. The resonant current loop is refreshed through the body diode of S_a4, and the resonant cavity input voltage becomes 0 V. After the dead time, at t₂, S_a4 can achieve ZVS turn-on.

Phase [t₂–t₃]: During this stage, S_a4 turns on. The resonant cavity input voltage remains at 0 V, and the resonant current is smaller than the excitation current. The resonant inductor and resonant capacitor continue to resonate in series. The excitation inductance is clamped to −V_o/n, causing the excitation current to decrease linearly. When the resonant current equals the excitation current, the transformer’s primary and secondary sides are decoupled. At this point, the excitation inductance resonates with the resonant inductor and capacitor.

Phase [t₃–t₄] Stage: At t₃, S_a4 turns off, and the resonant current becomes negative. The resonant current is restored by the body diode of S_a2 and capacitor C_dc2, and S_a2 reverses conduction. The resonant cavity input voltage becomes V_in, after which S_a2 can achieve ZVS turn-on.

Mode II of the 0.25-times gain mode of the full-bridge T-type LLC converter is symmetrical to Mode I. When operating in Mode II, switches S_a2 and S_b1 remain continuously on, while switches S_a4, S_b2, S_b3, and S_b4 remain off. Switches S_a1 and S_a3 operate complementarily with a 50% duty cycle. The corresponding waveforms and equivalent circuits for this mode are shown in Figure 9a and Figure 9b, respectively.

Using the fundamental harmonic approximation (FHA) method, it can be shown that the normalized DC voltage gain function of the full-bridge T-type LLC converter in both 0.25-times gain modes can be expressed by Equation (3). From Equation (3), it is evident that the voltage gain of the converter in both 0.25-times gain modes is 0.25 when the switching frequency equals the resonant frequency, representing only 0.25 times the voltage gain of the full-bridge mode with identical circuit parameters and switching frequency.

Since the initial voltages of C_dc1 and C_dc2 may not be balanced in practice, and since operating in a single mode would cause the capacitor voltage to drift, Mode 1 and Mode 2 of the 0.25-times gain mode should operate alternately. This alternation ensures that the voltage across the dividing capacitors remains balanced within the desired range by controlling the duration of each mode.

$$G={\displaystyle \frac{0.25}{\sqrt{{\left(1+\frac{1}{k}\left(1-\frac{1}{f{n}^2}\right)\right)}^2+\left(Q^2{\left(fn-\frac{1}{fn}\right)}^2\right)}}}$$

(3)

3.3. Conventional 1 and 0.5 Gain Operating Mode

The conventional 1-times gain modulation method, introduced in Section 2, utilizes variable duty cycle and frequency modulation to achieve one times the normalized gain. This method is not repeated here. The full-bridge T-type three-level LLC converter, shown in Figure 10, operates in half-bridge mode, where S_a2 and S_a3 remain on, S_a1, S_a4, S_b2, and S_b3 remain off, and S_b1 and S_b4 operate with a 50% duty cycle.

The full-bridge T-type three-level LLC resonant converter operating in half-bridge mode undergoes the following stages during one cycle:

[t₀–t₁]: During this stage, S_b4 conducts, and the resonant cavity input voltage is 0.5 V_in.

[t₁–t₂]: At t₁, S_b4 is turned off. Since the resonant current is greater than zero, it forces the body diode of S_b1 to reverse conduction, renewing the current. After the current is renewed, the resonant cavity input voltage becomes −0.5 V_in. The time interval from t₁ to t₂ is the dead time. At t₂, S_b1 turns on, and ZVS is achieved due to the reverse conduction during the turn-on.

[t₂–t₄]: The converter then enters the next half-cycle of operation, which is symmetric to the previous one (from t₀ to t₂). ZVS is also realized for S_b4 at t₄.

Through the analysis of the 0.5-times gain mode, it can be seen that the alternate conduction of the upper and lower bridges not only reduces the voltage stress on the switching devices by half but also helps balance the input capacitors (Vc_dc1 = Vc_dc2), mitigating the effects of uneven voltage distribution. The normalized voltage gain in this modulation mode is as follows:

$$G={\displaystyle \frac{0.5}{\sqrt{{\left(1+\frac{1}{k}\left(1-\frac{1}{f{n}^2}\right)\right)}^2+\left(Q^2{\left(fn-\frac{1}{fn}\right)}^2\right)}}}$$

(4)

3.4. Control Strategy of Full-Bridge T-Type Three-Level LLC Resonant Converter

In summary, based on the gain function of the full-bridge T-type three-level LLC converter operating in different modes, the corresponding gain curves can be plotted, as shown in Figure 11. As seen in Figure 11, compared to the original operating modes, the proposed 0.75-times and 0.25-times gain modes enable the full-bridge T-type LLC converter to achieve an extended voltage gain range with minimal switching frequency variation, thereby enabling a wide output voltage range.

The full-bridge T-type three-level LLC resonant converter operates in four distinct modes, each of which is selected based on the gain range, as shown in Figure 12. The PI controller, G_V, adjusts the switching frequency according to the rated output voltage, V_ref. In this figure, V_ref represents the reference output voltage, V_O denotes the actual output voltage, and V_err indicates the voltage error. Additionally, G_pre refers to the predicted voltage gain, and f_s represents the output frequency of the primary side inverter bridge section.

To enhance the accuracy of the converter’s gain prediction, the mode prediction is incorporated into the mode selection process. Based on the output voltage, the required gain of the converter is determined, and the appropriate operation mode is selected according to the voltage gain. The relationship between voltage gain and the mode of operation is described by a specific relational equation, as shown in the mode selection diagram in Figure 12. This approach enables the converter to achieve a wide output voltage range through mode selection and variable frequency control, resulting in a voltage gain of more than six times the normalized voltage.

4. Simulation and Experimental Results

4.1. Simulation of Different Modulation Modes for Full-Bridge T-Type Three-Level LLC Converter

To verify the feasibility of the proposed modulation method to achieve the goal of wide voltage output, a complete simulation model of a full-bridge T-type three-level LLC resonant converter is constructed in this section based on the MATLAB/Simulink simulation platform. The main parameters of the simulation are shown in

Table 1. Parameter table for simulation experiment of full-bridge T-type three-level LLC resonant converter.

Component	Value	Component	Value
Vin	200 V	Lr	7.8 μH
Vout	100 V–400 V	Cr	900 nF
fr	60 kHz	Lm	31.2 μH
n	2	Rload	60 $\mathrm{\Omega }$
Cdc1/Cdc2	800 μF

.

For the modulation method proposed in Section 3, simulation experiments were conducted to verify its effectiveness. As shown in Figure 13, with an input voltage of 200 V and a transformer turns ratio of 1:2 on both the primary and secondary sides, the system operates at a resonant frequency of 60 kHz. The output voltage was switched between 400 V and 100 V three times, as shown in the simulation waveforms. Figure 13 clearly demonstrates the feasibility of this modulation method, which meets the requirement for a four-fold output voltage ratio under resonant frequency conditions. Moreover, during the switching process, no overvoltage or overcurrent conditions occurred, preventing significant damage to the converter. The input filter capacitor voltage remained stable at approximately 100 V, with only a small deviation of about 0.5 V. This deviation can be attributed to the converter’s balancing of capacitance charging and discharging during the switching between sub-modes.

Figure 14 shows the simulated waveforms of the converter in the unity gain mode, with an input voltage of 200 V and operating at the resonant frequency. When a complementary drive signal is detected at unit gain and a dead time is applied, the reverse-connected T-type devices remain in the off state. At this point, the output voltage stabilizes around 400 V with a ripple of approximately 0.25%. Meanwhile, the secondary current follows a continuous half-sine waveform, enabling continuous energy transfer from the primary side to the secondary side.

As shown in Figure 15, the simulated experimental waveform of the transition from sub-mode 1 to sub-mode 2 in 0.75-times gain mode is presented. It can be observed that the gate drive signal V_gs1 changes from a 50% duty cycle to a constant off state, while V_gs3 changes from a 50% duty cycle to a constant on state. With an input voltage of 200 V, the output voltage fluctuates around 300 V with a ripple rate not exceeding 1.67%. The switching process lasts 1 to 2 cycles, with minimal energy loss, no overcurrent in the resonant and excitation currents, and very little current jitter. This effectively achieves the transition between sub-modes while maintaining the voltage balance of the input capacitor. A 0.75-times modulation method is implemented at the resonant frequency, resulting in an output voltage that is 0.75 times the normalized gain.

As shown in Figure 16, the steady-state waveform of the bridge arm operation in 0.5× gain mode is presented. In this mode, the switching tubes S_a2 and S_a4 operate complementarily at the resonant frequency. The output voltage is half of the output voltage in the doubled gain mode, stabilizing around 200 V, with a ripple rate not exceeding 1.5%. The secondary-side current follows a continuous half-sinusoidal waveform, and energy is continuously transferred from the primary to the secondary side. Both the resonant current and excitation current remain within the permissible current range, ensuring that the converter achieves an output voltage half of the normalized gain at the resonant frequency.

As shown in Figure 17, the steady-state waveform of the switching process from sub-mode 2 to sub-mode 1 in the 0.25-times gain mode is presented. It can be observed that the gate drive signal V_gs1 changes from a 50% duty cycle to a constant off state, while V_gs3 changes from a 50% duty cycle to a constant on state. The input voltage is 200 V, and the output voltage fluctuates around 100 V, with a ripple rate not exceeding 3%. The switching process lasts for 1 to 2 cycles, with minimal energy loss, no overcurrent in the resonant and excitation currents, and minimal current jitter. This effectively enables switching between sub-modes while maintaining the voltage balance of the input capacitor. The resonant frequency is modulated by a factor of 0.25, resulting in an output voltage that is 0.25 times the normalized gain.

4.2. Experiments on Different Modulation Modes of Full-Bridge T-Type Three-Level LLC Converter

To verify the feasibility of the output voltage in the proposed 0.75-fold and 0.25-fold gain modes and to achieve a wide output voltage range, a full-bridge T-type three-level LLC resonant converter is constructed, as shown in Figure 18. The key parameters are as follows: Lr = 8.2 μH (including 0.4 μH transformer leakage inductance), Cr = 0.88 μF, Lm = 31.2 μH, fr = 58 kHz, and the transformer turns ratio n = 2. The MOSFET switching device is the STW70N60DM2 from STMicroelectronics. The rectifier diode is the VS-E5PX3006L-N3, and the controller used is the DSP F28377D.

As shown in Figure 19 and Figure 20, the experimental waveforms are obtained under the traditional 1-times gain and 0.5-fold gain modes. The input voltage is a DC 200 V, and the switching frequency is 58 kHz. The output voltages are 400 V and 200 V, respectively, corresponding to the unity and 0.5-fold normalized voltage gain. The peak resonant currents are 26 A and 13 A, respectively, with the output load set to R_load = 120 Ω. The experimental waveforms show that the output voltages are stable at 400 V and 200 V, with very small ripple. Additionally, the resonant inductor current and excitation inductor current align with the theoretical analysis and simulation results, confirming the validity and feasibility of the modulation scheme.

The dynamic experimental waveforms of the proposed 0.75-fold gain modulation mode, switching from sub-mode 2 to sub-mode 1, are shown in Figure 21. In sub-mode 2, the voltages of V_AB are 0.5 times and −1 times the input voltage. After switching to sub-mode 1, the voltages of V_AB become 1 and −0.5 times the input voltage. Both the theoretical analysis and simulation experiments validate the experimental waveforms, confirming their feasibility and correctness. During the switching process, there is no overcurrent in the resonant or excitation currents, which ensures the safety and reliability of the converter’s operation. The input voltage remains at DC 200 V, and the output voltage is 300 V with minimal ripple. The output load is set to R_load = 120 Ω, and the switching frequency is 58 kHz, with a peak resonant current of 20.7 A.

Figure 22 shows the steady-state experimental waveform after switching to sub-mode 2. In this sub-mode, the output voltage is 0.5 times and −1 times the input voltage, achieving 0.75 times the normalized gain of 300 V. The experimental waveforms align with the theoretical analysis and simulation results, proving that the modulation method is correct and feasible. This approach is conducive to achieving a wide voltage range output.

The dynamic experimental waveforms for the proposed 0.25-fold gain modulation mode, switching from sub-mode 2 to sub-mode 1, are shown in Figure 23. In sub-mode 2, the voltage of V_AB is −0.5 times the input voltage. After switching to sub-mode 1, the voltage of V_AB becomes 0.5 times the input voltage. Both theoretical analysis and simulation experiments validate the experimental waveforms, confirming their feasibility and correctness. During the switching process, neither the resonant current nor the excitation current exceeds the current limit, ensuring the safety and reliability of the converter’s operation. The input voltage is maintained at 200 V DC, and the output voltage is 100 V with minimal ripple. The output load is set to R_load = 120 Ω, the switching frequency is 58 kHz, and the peak resonant current is 9.2 A.

Figure 24 shows the steady-state experimental waveforms after switching to sub-mode 1. In this sub-mode, the output voltage is 0.5 times the input voltage, resulting in a normalized gain of 0.25 times 100 V. The experimental waveforms align with the theoretical analysis and simulation results, demonstrating the correctness and feasibility of the modulation method. This approach facilitates the realization of a wide voltage range output.

As shown in Figure 25, the heavy load waveform operates in 0.75-times gain mode with an input voltage of 100 V. Figure 26 presents the dynamic waveform of the output voltage, with the input voltage fixed at 100 V. Figure 26a shows the dynamic change of the voltage from 240 V to 180 V in 1-times gain mode, Figure 26b shows the dynamic change of the voltage from 180 V to 140 V in 0.75-times gain mode, Figure 26c shows the dynamic change of the voltage from 130 V to 90 V in 0.5-times gain mode, and Figure 26d shows the dynamic change of the voltage from 60 V to 50 V in 0.25-times gain mode. From the waveforms in these four sub-figures, it is evident that there is no over-voltage or over-current in the input voltage and resonant current during the voltage transition. This demonstrates that the converter can achieve a wide voltage range output by selecting different operating modes for voltage variation.

The efficiency test results of the converter operating in the 0.75× gain mode are shown in Figure 27 with an input voltage of 200 V. At an output power of 1800 W and a switching frequency of 60 kHz, the converter achieves a conversion efficiency of 93.02%. These efficiency measurements for the proposed 0.75× gain modes at different output powers and frequencies verify the feasibility and effectiveness of the proposed method.

5. Conclusions

In this paper, a wide-output-voltage-range control method based on a full-bridge T-type three-level LLC resonant converter is proposed. The proposed 0.75-fold and 0.25-fold voltage gain modulation techniques effectively extend the output voltage range, addressing the challenges encountered by conventional LLC resonant converters in wide voltage applications. By combining the T-type three-level bridge arm with PWM modulation and employing a variable topology and frequency control strategy, the method successfully broadens the voltage gain range. Additionally, the PWM modulation method enhances the output voltage range by dynamically adjusting both the topology and frequency, effectively overcoming the limitations of traditional LLC converters. Simulation models and experimental prototypes developed in this study validate the effectiveness and feasibility of the proposed method. Experimental results demonstrate that the method achieves an input voltage of 100 V and an output voltage with a gain exceeding six times, reaching 40 V–240 V. This provides a reliable solution for applications requiring a wide output voltage range, such as new energy power generation systems and electric vehicle charging equipment. In the future, this converter can be applied to practical systems, such as electric vehicle charging or renewable energy interfaces.