Design and Implementation of Novel DC-DC Converter with Step-Up Ratio and Soft-Switching Technology

Abstract

This paper focuses on the development of a high-conversion-efficiency DC/DC boost converter, which features high-voltage boost ratio conversion and employs soft-switching technology to reduce conversion losses. In the proposed design, the conventional energy storage inductor used in traditional boost converters is replaced with a coupled inductor, and an additional boost circuit is introduced. This configuration allows the converter to achieve a higher voltage conversion ratio under the same duty cycle, thereby enhancing the voltage gain of the converter. Additionally, a resonance branch is incorporated into the converter, and by applying a simple switching signal control, zero-voltage switching (ZVS) of the main switch is realized. To decrease the switching losses typically found in hard-switching high-voltage boost ratio converters, the proposed design enhances overall power conversion efficiency. The operation principle of this novel high-voltage boost ratio soft-switching converter is first examined, followed by the component design process. The converter’s effectiveness is then confirmed through simulation in PSIM. Finally, experimental testing using the TMS320F2809 digital signal processor demonstrates that the main switch achieves ZVS, validating the practical viability of the design. The converter operates under a full load of 340 W, achieving a conversion efficiency of 92.7%, demonstrating the excellent conversion performance of the developed converter.

1. Introduction

In recent years, renewable energy has gained increasing attention, with photovoltaic power generation systems becoming one of the most widely applied forms of renewable energy. These systems are characterized by sustainable energy supply and minimal environmental impact. With advancements in technology, the installation and operational costs of photovoltaic power generation systems have continuously decreased, and their long-term maintenance costs are also relatively low, making them an ideal energy supply option. Additionally, with the rapid development of power electronic technology, converters play a crucial role in photovoltaic power generation systems, as they can significantly improve the energy conversion efficiency of the system. The primary function of a converter is to convert the direct current (DC) generated by the photovoltaic module array into alternating current (AC) or other DC values for use in the power grid or energy storage. Due to the relatively low-voltage output of photovoltaic modules and the need for subsequent applications in high-voltage systems, boost converters [1,2,3] are commonly used in photovoltaic power generation systems. However, boost converters also have certain drawbacks. When the switch is operated within a reasonable duty cycle, in order to prevent overheating and damage to the switch during prolonged operation, the voltage gain of the converter is limited, thus failing to meet the requirement for high output voltage. Moreover, if the switching frequency is too high when operating the switch, it results in significant switching losses. Although soft-switching technologies [4,5,6,7] can reduce switching losses, they increase the size and cost of the converter and still cannot achieve the required higher output voltage. The overall efficiency and economic viability of the photovoltaic power system are impacted by these challenges.

In recent years, many researchers have begun to explore high-voltage boost ratio converters [8,9,10,11] to address the drawbacks of traditional boost converters. High-voltage boost ratio converters can convert the relatively low output voltage from a photovoltaic module array to a higher voltage, thereby reducing energy losses and improving system efficiency. However, this technology also faces several challenges. The design of high-voltage boost ratio converters is more complex and requires advanced electronic components and control techniques, which leads to increased manufacturing costs and greater risks to system reliability. Additionally, high-voltage boost ratio converters operate in high-voltage environments, posing potential voltage breakdown and insulation issues. Therefore, special insulation designs and protection measures are necessary. The boost converter presented in reference [8] demonstrates excellent step-up performance. However, due to the use of a three-winding coupled inductor, which increases the voltage gain through the turns ratio, the converter’s physical size becomes significantly larger. In addition, the main switch in this converter does not achieve soft switching, resulting in higher switching losses during transitions between the on and off states. Furthermore, as the output voltage increases, the main switch must withstand higher voltage and current levels, significantly reducing the switch’s lifespan. Reference [9] employs an integrated transformer and voltage multiplier technique to achieve a higher voltage gain and smoother input current; however, it suffers from significant copper losses in the magnetic components at high-voltage boost ratios and requires a more complex control strategy. Reference [10] presents an inductively coupled soft-switching bidirectional converter, which offers a high-voltage boost ratio and reduces input voltage ripple. However, it requires five switch components and six capacitors, which increases the size and cost of the physical hardware. Reference [11] introduces an interleaved high-voltage ratio boost converter with coupled inductor. However, during the interleaved operation, the inactive converter still transmits some power, which reduces the overall efficiency and causes distortion in the average current. Reference [12] proposed an inductively coupled soft-switching bidirectional converter that also achieves a high voltage boost ratio. However, its main switch components must withstand higher voltage switching stresses, requiring switches with higher voltage ratings.

In response to the limitations of the converters discussed above, this study introduces a high-voltage boost ratio converter with soft-switching capabilities. This converter uses a coupled inductor to replace the conventional energy storage inductor and incorporates a boost circuit, enabling a higher voltage conversion ratio under the same duty cycle, thereby improving the voltage gain of the converter. In addition, this paper incorporates a resonance branch into the converter and uses a simple switch signal control to achieve ZVS of the main switch, reducing switching losses in hard-switching high-voltage boost ratio converters and improving conversion efficiency. First, the circuit architecture of the proposed converter was established using PSIM (Version 9.1) simulation software [13], and its feasibility was verified through simulation analysis. Finally, the implementation was carried out using a TMS320F2809 digital signal processor [14] from Texas Instruments (Dallas, TX, USA), which demonstrated the superior conversion performance of the high-voltage boost ratio soft-switching converter.

This paper is divided into several sections, beginning with Section 2, which elaborates on the proposed high-voltage boost ratio DC-DC converter circuits—both in hard-switching and soft-switching forms—detailing their architecture, operation, and analytical modeling. Section 3 explains the design methodologies and parameter selection for the main components of the converter. Section 4 utilizes PSIM software to perform simulation analyses, verifying the converter’s performance under different load conditions. Section 5 describes the hardware implementation using a TMS320F2809 digital signal processor for experimental testing and provides a performance comparison between the hard-switching and soft-switching high-voltage boost ratio DC-DC converters. Finally, Section 6 summarizes the research findings and suggests potential directions for future work.

2. The Proposed High-Voltage Boost Ratio Converter

2.1. Operating Principle of the High-Voltage Boost Ratio Hard-Switching Converter

Figure 1 depicts the circuit architecture of the proposed high-voltage boost ratio hard-switching converter. This design replaces the traditional energy storage inductor with a coupled inductor and incorporates an additional boost circuit formed by $D_1$ and $C_1$. By leveraging the turns ratio of the coupled inductor and the boost circuit, the voltage conversion ratio is significantly enhanced. Furthermore, the converter features a simple circuit structure and ease of control, making it advantageous for practical applications. The working principle of the converter includes two operating modes, corresponding to the switch being either on or off. During the on state, the duty cycle D over one switching period T is described in Equation (1).

$$D\triangleq {\displaystyle \frac{t_{\mathrm{on}}}{T}}={\displaystyle \frac{t_{\mathrm{on}}}{t_{\mathrm{on}}+t_{\mathrm{off}}}}$$

(1)

where $t_{\mathrm{on}}$ represents the duration of when the switch is in the conducting state within a single switching period, and $t_{\mathrm{off}}$ denotes the interval during which the switch is turned off.

(1)

Switch on ($0\le t\le DT$)

When switch S is conducting, diode $D_2$ is also conducting, while diode $D_o$ and $D_1$ are in the off state. Figure 2 presents the equivalent circuit, and the turns ratio N of the coupled inductor is specified in Equation (2). At this point, inductor voltages $v_{L1}$ and $v_{L2}$ are expressed by Equations (3) and (4), respectively, while voltage $v_{C2}$ across energy storage capacitor $C_2$ is given by Equation (5).

$$N\stackrel{\mathrm{\Delta }}{=}{\displaystyle \frac{N_2}{N_1}}$$

(2)

$$v_{L1}=V_i$$

(3)

$$v_{L2}={\displaystyle \frac{N_2}{N_1}}{V}_i=N{V}_i$$

(4)

$$v_{C2}=v_{L2}+v_{C1}=V_i\left(N+{\displaystyle \frac{1}{1-D}}\right)$$

(5)

(2)

Switch off ($DT\le t\le T$)

When switch S is off, diode $D_2$ is in the off state, while diodes $D_o$ and $D_1$ are conducting. Figure 3 illustrates the corresponding equivalent circuit, and the inductor voltages $v_{L1}$ and $v_{L2}$ are given by Equations (6) and (7), respectively.

$$v_{L1}=V_i-v_{C1}=-V_i{\displaystyle \frac{D}{1-D}}$$

(6)

$$v_{L2}=V_i+v_{C2}-v_{L1}-V_o=N{V}_i+2{\displaystyle \frac{V_i}{1-D}}-V_o$$

(7)

Based on the volt–second balance theorem for inductor $L_2$, Equations (4) and (7) lead to Equation (8). After simplification, the voltage conversion ratio between output voltage $V_o$ and input voltage $V_i$ can be expressed by Equation (9).

$$N{V}_{i}DT+(N{V}_i+2{\displaystyle \frac{V_i}{1-D}}-V_o)(1-D)T=0$$

(8)

$$G={\displaystyle \frac{V_o}{V_i}}={\displaystyle \frac{2+N}{1-D}}$$

(9)

Based on Equation (9),

Table 1. Relationship between voltage gain and duty cycle for the high-voltage boost ratio hard-switching converter.

	Turns Ratio	N = 2	N = 3	N = 4	N = 5	N = 6
Duty Cycle
D = 0.1		G = 4.4	G = 5.6	G = 6.7	G = 7.8	G = 8.9
D = 0.2		G = 5	G = 6.3	G = 7.5	G = 8.8	G = 10
D = 0.3		G = 5.7	G = 7.1	G = 8.6	G = 10	G = 11.4
D = 0.4		G = 6.7	G = 8.3	G = 10	G = 11.7	G = 13.3
D = 0.5		G = 8	G = 10	G = 12	G = 14	G = 16
D = 0.6		G = 10	G = 12.5	G = 15	G = 17.5	G = 20
D = 0.7		G = 13.3	G = 16.7	G = 20	G = 23.3	G = 26.7
D = 0.8		G = 20	G = 25	G = 30	G = 35	G = 40

summarizes the relationship between the converter’s voltage gain and the duty cycle. As illustrated in Table 1 and Figure 4, increasing the turns ratio of the coupled inductor leads to a higher voltage conversion ratio at the same duty cycle.

2.2. Operating Principle of High-Voltage Boost Ratio Soft-Switching Converter

The high-voltage boost ratio hard-switching converter described above can increase the converter’s voltage conversion ratio by utilizing a coupled inductor and an additional boost circuit. However, if the voltage or current cannot be reduced to zero before switching occurs, switching losses will arise, which in turn lowers the overall efficiency of the converter. To improve the conversion efficiency, this paper proposes a high-voltage boost ratio soft-switching converter with inductive coupling. The circuit architecture of this converter is shown in Figure 5, while Figure 6 illustrates the switching control signals for the converter [15]. The converter achieves a high-voltage conversion ratio through inductive coupling and an additional boost circuit. It also incorporates a resonant branch composed of $L_r$ and $S_r$, where $C_1$ is the capacitor originally used in the hard-switching converter circuit. By controlling the switching signals, the converter enables ZVS for the main switch, thus improving efficiency and reducing switching losses. This paper will analyze the operating modes of the proposed soft-switching converter, which is divided into nine distinct modes. The corresponding switching waveforms for the components are illustrated in Figure 7. Prior to the mode analysis, the following assumptions are introduced:

(1)

The converter operates in continuous conduction mode (CCM), with the circuit assumed to be in a steady-state condition.

(2)

All components are assumed to be ideal, meaning that during conduction, they are treated as short circuits, and during cutoff, as open circuits. Consequently, the voltage drop across the switching devices during conduction is considered zero.

(3)

The input and output voltages are maintained at constant values.

(4)

The currents of energy storage inductors $L_1$ and $L_2$ are considered constant (i.e., $i_{L1}=I_{L1}$ and $i_{L2}=I_{L2}$).

(1) 

Mode 1 ($t_0~t_1$)

Figure 8 presents the equivalent circuit of the converter while operating in Mode 1. In this mode, main switch S is in the off state, while auxiliary switch S_r is turned on first. As a result, resonant capacitor $C_1$ begins to discharge, and the voltage across resonant inductor $L_r$ is $v_{C1}$. The current through resonant inductor $i_{Lr}$ rises from zero. Therefore, resonant inductor $L_r$ and resonant capacitor $C_1$ form a resonant tank, and the circuit equations can be expressed by Equation (10). After solving this, $i_{Lr}$ and $v_{C1}$ are given by Equation (11), while resonant impedance $Z_o$ and resonant angular frequency ${\omega }_o$ are given by Equation (12). When voltage $v_{C1}$ across the resonant capacitor drops to $V_i$, diode $D_1$ switches from the off state to the on state. The circuit then transitions to Mode 2. The time for this transition is given by Equation (13).

$$\left\{\begin{array}{l}{i}_{Lr}(t_o)=0\\ v_{C1}(t_o)={\displaystyle \frac{V_i}{1-D}}\\ i_{C1}(t)=-i_{Lr}(t)\\ v_{C1}(t)=v_{Lr}(t)\end{array}\right., t_0\le t\le t_1$$

(10)

$$\left\{\begin{array}{l}{i}_{Lr}(t)={\displaystyle \frac{V_i}{(1-D)Z_o}}\sin {\omega }_o(t-t_o)\\ v_{C1}(t)={\displaystyle \frac{V_i}{1-D}}\cos {\omega }_o(t-t_o)\end{array}\right., t_0\le t\le t_1$$

(11)

where resonance impedance $Z_o\triangleq \sqrt{{\displaystyle \frac{L_r}{C_1}}}$ and resonance frequency

$${\omega }_o\triangleq {\displaystyle \frac{1}{\sqrt{L_r{C}_1}}}$$

(12)

$$T_1=t_1-t_o=\sqrt{L_r{C}_1}{\cos }^{-1}\left(1-D\right)$$

(13)

(2) 

Mode 2 ($t_1~t_2$)

Upon entering Mode 2, the corresponding equivalent circuit is illustrated in Figure 9. At this point, the voltage across resonant capacitor $v_{C1}$ decreases to $V_i$. At this point, auxiliary switch S_r remains on, and both resonant inductor $L_r$ and resonant capacitor $C_1$ continue to form a resonant circuit. The current through resonant inductor $i_{Lr}$ continues to rise, while resonant capacitor $C_1$ undergoes continuous discharge. The circuit equations for this mode can be expressed as Equation (14). From Equation (14), $i_{Lr}(t)$ and $v_{C1}(t)$ can be derived as shown in Equation (15). When the voltage across the resonant $v_{C1}$ capacitor drops to zero, at which point ${\omega }_o(t-t_1)={\displaystyle \frac{\pi }{2}}$, resonant inductor current $i_{Lr}$ can be calculated from Equation (15) as shown in Equation (16). The operating time for this mode can be derived as shown in Equation (17). At this point, the anti-parallel diode connected to main switch S starts conducting, and the converter proceeds to operate in Mode 3.

$$\left\{\begin{array}{l}{i}_{Lr}(t)+C_1{\displaystyle \frac{d{v}_{C1}(t)}{dt}}=I_{L1}\hfill \\ \begin{array}{l}{L}_r{\displaystyle \frac{d{i}_{Lr}(t)}{dt}}=v_{C1}(t)\\ v_{C1}(t_1)=V_i\end{array}\hfill \end{array}\right., t_1\le t\le t_2$$

(14)

$$\left\{\begin{array}{l}{i}_{Lr}(t)=I_{L1}+{\displaystyle \frac{V_i}{Z_o}}\sin {\omega }_o(t-t_1)\\ v_{C1}(t)=V_i\cos {\omega }_o(t-t_1)\end{array}\right., t_1\le t\le t_2$$

(15)

$$i_{Lr}(t_2)=i_{Sr}(t_2)=I_{L1}+{\displaystyle \frac{V_i}{Z_o}}$$

(16)

$$T_2=t_2-t_1={\displaystyle \frac{\pi }{2}}\sqrt{L_r{C}_1}$$

(17)

(3) 

Mode 3 ($t_2~t_3$)

During Mode 3, the circuit behaves as illustrated in the equivalent model shown in Figure 10. At this point, the voltage across resonant capacitor $v_{C1}$ drops to a very small negative voltage, which causes the anti-parallel diode of main switch S to forward conduct. As a result, the voltage across main switch S is zero. The control of main switch S is then switched from the off state to the on state, achieving ZVS for main switch S. At the same time, auxiliary switch S_r is switched off. The circuit equations for this mode can be expressed as Equation (18). To ensure ZVS for the main switch, delay time $t_d$ must satisfy the condition in Equation (19), and typically, $t_d$ is 5% to 10% of switching period T. To guarantee the zero-voltage switching (ZVS) of main switch S under heavy load conditions and account for the turn-off delay of auxiliary switch S_r, time delay $t_{\alpha }$ is introduced. Accordingly, conduction time $t_{DSr}$ of the auxiliary switch is defined by Equation (20).

$$\left\{\begin{array}{l}{i}_S(t_2)=I_{L1}-i_{Lr}(t_2)=-{\displaystyle \frac{V_i}{Z_o}}\\ i_{Lr}(t_2)=I_{L1}+{\displaystyle \frac{V_i}{Z_o}}\end{array}\right., t_2\le t\le t_3$$

(18)

$$t_d\ge T_1+T_2=\sqrt{L_r{C}_1}\left({\displaystyle \frac{\pi }{2}}+{\cos }^{-1}\left(1-D\right)\right)$$

(19)

$$t_{DSr}\ge t_d+t_{\alpha }=\sqrt{L_r{C}_1}\left({\displaystyle \frac{\pi }{2}}+{\cos }^{-1}\left(1-D\right)\right)+t_{\alpha }$$

(20)

(4) 

Mode 4 ($t_3~t_4$)

In Mode 4, main switch S is conducting, while auxiliary switch S_r switches off. Diodes $D_1$ and $D_2$ are forward-biased, and the equivalent circuit is shown in Figure 11. In this mode, to allow energy storage capacitor $C_2$ and resonant inductor $L_r$ to form a resonant tank, the capacitance of $C_2$ is chosen to be close to that of $C_1$. Additionally, resonant inductor $L_r$ and common-mode inductance $L_2$ both discharge into the energy storage capacitor $C_2$. At this point, the current through resonant inductor $L_r$ is $I_{L1}+{\displaystyle \frac{V_i}{Z_o}}$. The circuit equation for this mode can be expressed as Equation (21). Solving for $i_{Lr}(t)$ and $v_{C2}(t)$ gives Equation (22). In this working mode, the current through resonant inductor $i_{Lr}$ decreases from $I_{L1}+{\displaystyle \frac{V_i}{Z_o}}$ to zero, and diodes $D_1$ and $D_2$ switch off. The circuit then transitions into Mode 5.

$$\left\{\begin{array}{l}{L}_r{\displaystyle \frac{d{i}_{Lr}\left(t\right)}{dt}}=v_{L2}(t)-v_{C2}(t)\hfill \\ C_2{\displaystyle \frac{d{v}_{C2}\left(t\right)}{dt}}=i_{Lr}(t)\hfill \\ v_{L2}(t)=N{v}_{L1}(t)=N{V}_i\hfill \\ v_{C1}\left(t_3\right)=0\hfill \\ i_{Lr}(t_4)=0\hfill \end{array}\right., t_3\le t\le t_4$$

(21)

$$\left\{\begin{array}{l}{i}_{Lr}(t)=\left(I_{L1}+{\displaystyle \frac{V_i}{Z_o}}\right)\cos {\omega }_o(t-t_3)+{\displaystyle \frac{N{V}_i}{Z_o}}\sin {\omega }_o(t-t_3)\\ v_{C2}(t)=N{V}_i\left[1+\cos {\omega }_o(t-t_3)\right]+\left(I_{L1}+{\displaystyle \frac{V_i}{Z_o}}\right)\sin {\omega }_o(t-t_3)\end{array}\right.$$

(22)

(5) 

Mode 5 ($t_4~t_5$)

During Mode 5, Figure 12 illustrates the corresponding equivalent circuit. At this point, main switch S remains conducting. The circuit equation for this mode can be expressed as Equation (23). This mode continues until main switch S transitions from conducting to off.

$$\left\{\begin{array}{l}{i}_S(t)=I_{L1}\hfill \\ v_{C1}(t)=0\hfill \end{array}\right., t_4\le t\le t_5$$

(23)

(6) 

Mode 6 ($t_5~t_6$)

In Mode 6, main switch S is turned off and the equivalent circuit is shown in Figure 13. As a result, current $I_{L1}$ charges resonant capacitor $C_1$, causing the voltage across resonant capacitor $v_{C1}$ to gradually increase. The corresponding circuit behavior in this mode can be described using Equation (24). By solving Equation (24), $v_{C1}(t)$ can be obtained as Equation (25). When resonant capacitor voltage $v_{C1}$ reaches $V_i$, this mode ends and transitions into Mode 7. As a result, the duration of this mode can be calculated using Equation (26).

$$C_1{\displaystyle \frac{d{v}_{C1}(t)}{dt}}=I_{L1}, t_5\le t\le t_6$$

(24)

$$v_{C1}(t)={\displaystyle \frac{I_{L1}}{C_1}}(t-t_5)$$

(25)

$$T_6=t_6-t_5={\displaystyle \frac{V_i{C}_1}{I_{L1}}}$$

(26)

(7) 

Mode 7 ($t_6~t_7$)

In Mode 7, the voltage across resonant capacitor $v_{C1}$ is $V_i$, and main switch S remains in the off state, while auxiliary switch S_r is also off. As a result, coupled inductor $L_2$ and energy storage capacitor $C_2$ transfer energy to the load via diode $D_o$. The circuit equation for this mode is given by Equation (27), and the equivalent circuit is shown in Figure 14. By solving Equation (27), voltage $v_{C1}$ across resonant capacitor $C_1$ can be obtained as Equation (28). At this point, currents flowing through the coupled inductor $I_{L1}$ and $I_{L2}$ gradually decrease, and when currents $I_{L1}$ and $I_{L2}$ become equal, the circuit transitions into Mode 8.

$$\left\{\begin{array}{l}{C}_1{\displaystyle \frac{\mathrm{d}{v}_{C1}}{\mathrm{d}t}}=I_{L1}-I_{L2}\\ v_{C1}(t_6)=V_i\end{array}\right., t_6\le t\le t_7$$

(27)

$$v_{C1}(t)={\displaystyle \frac{I_{L1}-I_{L2}}{C_1}}(t-t_6), t_6\le t\le t_7$$

(28)

(8) 

Mode 8 ($t_7~t_8$)

In Mode 8, voltage across the resonant capacitor $v_{C1}$ is $V_i/(1-D)$, and diode $D_1$ transitions from the conducting to the off state. Resonant capacitor $C_1$ stops charging, and both main switch S and auxiliary switch S_r remain off. Consequently, the coupled inductor and energy storage capacitor $C_2$ continue to transfer energy to the load. The equivalent circuit for this mode is shown in Figure 15, and the circuit equation is given by Equation (29). This mode concludes when the voltage across energy storage capacitor $C_2$ reaches zero, triggering the circuit to shift into Mode 9.

$$\left\{\begin{array}{l}{I}_{L1}=I_{L2}\\ v_{C1}(t_7)={\displaystyle \frac{V_i}{1-D}}\end{array}\right., t_6\le t\le t_7$$

(29)

(9) 

Mode 9 ($t_8~t_9$)

In Mode 9, voltage across the energy storage capacitor $v_{C2}$ is zero, and diode $D_o$ transitions from conducting to the off state. Both main switch S and auxiliary switch S_r remain off. The equivalent circuit for this mode is shown in Figure 16, marking the completion of the entire switching cycle analysis. The cycle will repeat once auxiliary switch S_r turns on again, returning to Mode 1 of the next cycle.

In order to validate the performance of the proposed high-voltage boost ratio soft-switching converter, it is compared with several other high-voltage boost ratio soft-switching topologies. The comparison is based on voltage gain, switch voltage ratings, the number of switching components, the number of diode components, the number of inductive components, and the number of capacitive components. A summary of the results is presented in

Table 2. Component specifications of the high-voltage ratio soft-switching converter adopted.

Converter Topology	Converter in [4]	Converter in [5]	Converter in [6]	Converter in [12]	Proposed Converter
Voltage Gain	${\displaystyle \frac{2N+2-ND}{1-D}}$	${\displaystyle \frac{2+N}{1-D}}$	${\displaystyle \frac{2+N}{1-D}}$	${\displaystyle \frac{2N+2-ND}{1-D}}$	${\displaystyle \frac{2+N}{1-D}}$
Voltage Stress on MOSFETs	${\displaystyle \frac{V_O}{2N+2-ND}}$	${\displaystyle \frac{V_O}{2+N}}$	${\displaystyle \frac{V_O}{2+N}}$	${\displaystyle \frac{V_O-N{V}_i}{1+2N}}$	${\displaystyle \frac{V_i}{1-D}}$
MOSFETs	2	4	2	2	2
Diodes	3	0	2	3	3
Inductors	1	1	2	2	2
Capacitors	4	4	4	3	3

. As shown in the table, the proposed converter achieves a high voltage gain with fewer components and utilizes simple signal control for soft switching, highlighting its key advantages.

3. The Component Design of the Proposed High-Voltage Boost Ratio Converter

The output power of the high-voltage boost ratio hard-switching converter proposed in this paper is 340 W. The relevant circuit parameters and specifications are provided in

Table 3. Parameter specifications of the high-voltage boost ratio hard-switching converter.

Parameter	Value
Input Voltage Vi	72 V $\pm$ 10%
Output Voltage Vo	430 V
Output Power Po	340 W
Switching Frequency f	25 kHz
Turns Ratio of Coupling Inductor N	2

.

Assuming the ideal operation of all components, the input power is expected to be equal to the output power, i.e.,

$$V_i{I}_{L1}={\displaystyle \frac{V_o^2}{R}}$$

(30)

$$I_{L1}={\displaystyle \frac{V_o^2}{V_i^2}}{\displaystyle \frac{V_i}{R}}$$

(31)

Substituting Equation (9) into Equation (31) yields, as expressed in Equation (32).

$$I_{L1}={\left({\displaystyle \frac{2+N}{1-D}}\right)}^2{\displaystyle \frac{V_i}{R}}$$

(32)

where ${\displaystyle \frac{2+N}{1-D}}$ represents the voltage gain of the converter.

When main switch S is turned on, $v_{L1}$ can be expressed as

$$v_{L1}=V_i=L_1{\displaystyle \frac{\mathrm{d}{i}_{L1}}{\mathrm{d}t}}$$

(33)

From Equation (33), it can be observed that when the main switch is turned on, inductor current $i_{L1}$ increases linearly, with the conduction time denoted as $t_{\mathrm{on}}=DT$. The variation rate of the inductor current is determined by Equation (34).

$$\mathrm{\Delta }{i}_{L1(\mathrm{close})}={\displaystyle \frac{V_i}{L_1}}DT$$

(34)

By applying Equations (32) and (34), the peak and valley values of inductor current $i_{L1}$ can be obtained, as shown in Equations (35) and (36) [16].

$$I_{L1(\max )}=I_{L1}+{\displaystyle \frac{\mathrm{\Delta }{i}_{L1}}{2}}={\left({\displaystyle \frac{2+N}{1-D}}\right)}^2{\displaystyle \frac{V_i}{R}}+{\displaystyle \frac{V_{i}D}{2L_{1}f}}$$

(35)

$$I_{L1(\min )}=I_{L1}-{\displaystyle \frac{\mathrm{\Delta }{i}_{L1}}{2}}={\left({\displaystyle \frac{2+N}{1-D}}\right)}^2{\displaystyle \frac{V_i}{R}}-{\displaystyle \frac{V_{i}D}{2L_{1}f}}$$

(36)

If $I_{L1(\min )}=0$, the inductor can operate at the boundary between CCM and discontinuous conduction mode, which yields

$${\left({\displaystyle \frac{2+N}{1-D}}\right)}^2{\displaystyle \frac{V_i}{R}}={\displaystyle \frac{V_{i}D}{2L_{1(\min )}f}}$$

(37)

From Equation (37), the value of $L_{1(\min )}$ can be calculated as

$$L_{1(\min )}={\displaystyle \frac{DR}{2f}}{\left({\displaystyle \frac{1-D}{2+N}}\right)}^2$$

(38)

Since the coupled inductor structure is similar to the coupled transformer structure, the maximum value of $I_{L2}$ can be shown as [16]

$$I_{L2(\max )}=-{\displaystyle \frac{I_{L1(\max )}}{1+N}}$$

(39)

3.1. Design of Coupled Inductors

The given scenario involves ensuring that a converter operates in CCM across all duty cycles, with a maximum output power of 340 W, a fixed output voltage of 430 V, and a rated load of 550 Ω. From the relationship between duty cycle D and the ${\displaystyle \frac{D}{2}}{\left({\displaystyle \frac{1-D}{2+N}}\right)}^2$ function curve shown in Figure 17, it is observed that function ${\displaystyle \frac{D}{2}}{\left({\displaystyle \frac{1-D}{2+N}}\right)}^2$ reaches its maximum value when D = 1/3. By substituting load resistance R = 550 Ω, the turns ratio of coupled inductors N = 2, switching frequency f = 25 kHz, and duty cycle D = 1/3 into Equation (38), minimum primary inductance $L_{1(\min )}$ of the coupled inductor is calculated to be 102 μH. To ensure that coupling inductor $L_1$ operates in CCM under both light and heavy load conditions, the calculated inductance value is further multiplied by a safety factor of 1.25. Hence, the selected value of coupling inductor $L_1$ is 127 μH.

3.2. Design of Capacitors $C_1$ and $C_2$

From Figure 1 of the high-voltage boost ratio hard-switching converter, it can be observed that capacitor $C_1$, coupled inductor $L_1$, main switch S, and diode $D_1$ form a traditional boost converter. When the duty cycle of main switch S is 0.8, the voltage across capacitor $C_1$ is approximately 360 V. From Equation (5), under the same operating conditions, the voltage across capacitor $C_2$ is around 500 V. Therefore, the rated voltage of capacitor $C_1$ is selected as 400 V, while the rated voltage of capacitor $C_2$ is 600 V. Based on the analysis of operating Mode 3 in Equation (18) and Figure 7, it is evident that the capacitance value of $C_1$ influences the reverse current flowing through main switch S. Considering ripple size and component availability, the chosen capacitance value for $C_1$ is 0.33 μF/400 V. Additionally, from the explanation of operating Mode 4 of this high-voltage boost ratio soft-switching converter, it is understood that $C_2$ should have a capacitance value similar to that of $C_1$, so $C_2$ is selected as 0.33 μF/600 V.

3.3. Design of Resonant Inductor $L_r$

Since the conduction time of the auxiliary switch in a typical soft-switching converter is usually designed to be between 5% and 10% of the switching period, let $t_d=10\%T=4 \mathrm{\mu }\mathrm{s}$ ($T={\displaystyle \frac{1}{f}}={\displaystyle \frac{1}{25 \mathrm{kHz}}}=40 \mathrm{\mu }\mathrm{s}$) and $t_{\alpha }=2\%T=0.8 \mathrm{\mu }\mathrm{s}$ represent these values, respectively. From Equation (20), if duty cycle D is set within the range of 0.1 to 0.8, the resonance inductance value can be calculated to vary between 8.08 μH and 17.08 μH. Based on this, the chosen resonance inductance for the converter is 18 μH.

3.4. The Selection of Main Switch S Auxiliary Switch S_r

This paper adopts TK49N65W (650 V/49 A) silicon MOSFETs for both main switch S and auxiliary switch Sr. Although wide-bandgap devices such as GaN and SiC offer advantages like higher switching speed and lower losses at high frequencies (>100 kHz) [10,17,18], the proposed converter operates at a moderate switching frequency of 25 kHz. Under this condition, the ZVS operation significantly reduces switching losses, making the performance of traditional MOSFETs comparable to that of GaN/SiC devices without the added cost and complexity. Moreover, GaN/SiC devices would be underutilized in this frequency range and would not yield a significant efficiency improvement. Therefore, considering cost, availability, and the effective implementation of ZVS, conventional MOSFETs are a practical and sufficient choice for the proposed converter.

After the design of the aforementioned components, the specifications of the components used in the proposed high-voltage boost ratio soft-switching converter are listed in

Table 4. Component specifications of the high-voltage boost ratio soft-switching converter.

Component	Specifications
Coupled Inductor L1	127 μH
Resonant Inductor Lr	18 μH
Main Switch S	MOSFET-TK49N65W (650 V/49 A)
Auxiliary Switch Sr	MOSFET-TK49N65W (650 V/49 A)
Fast Diodes (Do, D1, D2)	IQBD30E60A1 (600 V/30 A)
Filtering Capacitor Co	340 μF/900 V
Resonant Capacitor C1	0.33 μF/400 V
Energy Storage Capacitor C2	0.33 μF/600 V

.

4. Simulation Results

In this paper, PSIM simulation software [13] was utilized to model the proposed high-voltage boost ratio soft-switching converter, aiming to validate its enhanced performance under both light and full load conditions. Figure 18 shows the waveform of the duty cycle, input voltage, and output voltage when the converter operates at full load (p = 340 W). The output voltage of 430 V is obtained when the input voltage is 72 V and the duty cycle is 0.33, as shown in Figure 18. As a result, the converter achieves a high-voltage conversion ratio, which aligns with the voltage gain specified in Table 1. Figure 19 and Figure 20 illustrate the simulated electrical waveforms of main switch S and auxiliary switch S_r under full load operation (p = 340 W). Figure 21 and Figure 22 show the simulation waveforms of the electrical quantities for main switch S and auxiliary switch S_r when the converter operates under light load (p = 100 W). From Figure 19 and Figure 21, it can be observed that at both full load (p = 340 W) and light load (p = 100 W), the voltage on main switch S falls to zero prior to the switch turning on. This demonstrates that main switch S achieves ZVS, confirming that the proposed converter enables soft switching for the main switch.

5. Experimental Results

After confirming the proposed converter’s feasibility through PSIM simulations, a practical prototype of the high-voltage boost ratio soft-switching converter was developed, using the TMS320F2809 digital signal processor [14] as its control core. The overall hardware circuit setup is shown in Figure 23, and the test platform is depicted in Figure 24. Figure 25 shows the waveforms of the trigger signal for the main switch, S, input voltage, and output voltage of the high-voltage boost ratio soft-switching converter under full load operation p = 340 W. From Figure 25, it can be observed that when the duty cycle of the main switch is 0.33 and the input voltage is 72 V, the output voltage reaches 430 V, achieving the expected voltage gain, as shown in Table 1. Figure 26, Figure 27, Figure 28 and Figure 29 show the waveforms of the trigger signals, voltages, and currents for main switch S and auxiliary switch S_r at load conditions of p = 100 W and p = 340 W, respectively. The experimental results demonstrate that the proposed high-voltage boost ratio soft-switching converter achieves ZVS for main switch S under both light load (p = 50 W) and full load (p = 340 W) conditions. This helps reduce the switching loss, thereby improving the overall conversion efficiency. Additionally, the measured waveforms of main switch S and auxiliary switch S_r match the simulated waveforms. Figure 30 shows a comparison of the efficiency between the proposed high-voltage boost ratio soft-switching and hard-switching converters, with loads ranging from p = 50 W to p = 340 W. The figure demonstrates that the conversion efficiency improves by 3–4% across different load levels.

The experimental prototype, including the heat sink and the voltage and current sensing circuits for the photovoltaic module array, occupies approximately 180 mm × 110 mm of board area, which is about 10–20% smaller than typical multi-winding or interleaved converter designs. The estimated cost of the main components is also approximately 20% lower than the multi-stage or multi-switch topologies reported in [4,5,6,12], making it well suited for compact photovoltaic system applications.

In Figure 26 and Figure 28, the oval-shaped area indicates that when the two-terminal voltage of the main switch drops to zero, the control signal is turned on, allowing the main switch to achieve zero-voltage switching.

6. Conclusions

This paper presents high-voltage boost ratio soft-switching converter topology.

Through both simulation analysis and experimental results, it has been demonstrated that the converter achieves zero-voltage switching (ZVS) for the main switch. By replacing the traditional energy storage inductor with a coupled inductor and adding a boost circuit, the overall voltage gain of the converter is enhanced, enabling it to achieve a higher voltage output. Additionally, a resonant branch is incorporated to ensure the main switch operates with ZVS, thereby reducing switching losses and switch stress, which in turn improves conversion efficiency. The proposed converter offers advantages such as ease of control via triggering signals, a simple circuit topology, and quantifiable component design parameters. Under different load conditions, the conversion efficiency improves by 3–4%, with a peak efficiency of approximately 93%. Thus, the converter demonstrates excellent conversion performance. This converter is promising for practical applications in photovoltaic module arrays, where it can be used for maximum power point tracking to enhance power generation efficiency.

Furthermore, since the switching frequency of this converter is 25 kHz, which is well below the frequency range where electromagnetic interference (EMI) becomes a critical issue (typically above 150 kHz), it inherently mitigates high-frequency radiated noise problems. At this stage, no dedicated EMI input/output filters have been implemented; however, future work is planned to conduct EMI measurements using a line impedance stabilization network and a spectrum analyzer. This acknowledges the current limitations of this study and clearly outlines future research directions to ensure the converter can better meet practical requirements and comply with international electromagnetic compatibility standards.