Differential-Fed Wideband Circularly Polarized SIW Cavity-Backed Slot Antenna Array

Abstract

This paper presents a wideband circularly polarized (CP) substrate-integrated waveguide (SIW) cavity-backed slot antenna array arranged in a 2 × 2 configuration with differential feeding structures. The design features arc-shaped microstrips within the SIW cavity to excite the ${\mathrm{TE}}_{011}^x$/${\mathrm{TE}}_{101}^y$ and ${\mathrm{TE}}_{211}^y$/${\mathrm{TE}}_{121}^x$ modes. By overlapping the center frequencies of the two modes, wideband CP radiation is achieved. The introduction of four modified ring couplers composes a simple but efficient differential feeding network, eliminating the need for balanced resistors like baluns, making it more suitable for millimeter wave or even higher frequency applications. Experimental results show that the antenna array achieves a −10 dB impedance bandwidth of 32.6% (from 17.28 to 24.00 GHz), a 3 dB axial ratio (AR) bandwidth of 13.8% (from 17.05 to 19.57 GHz), a 3 dB gain bandwidth of 41.8% (from 15.39 to 23.51 GHz) and a peak gain of 10.6 dBi, with results closely matching simulation data. This study enhances the development of differential CP SIW cavity-backed slot antenna arrays, offering a potential solution for creating compact integrated front-end circuits in the millimeter wave or Terahertz frequency range.

1. Introduction

Circularly polarized (CP) antennas are useful for many systems due to their advantages in reducing polarization mismatch loss, multipath distortion, etc. [1,2]. As the operating frequency rises and the integrated system becomes more compact, CP antennas with broadband, muti-band, high gain, low profile and low cross-polarization have a broader application prospect. The cavity-backed structure can improve the performance of CP antennas [3]. However, the substantial thickness of the metallic cavity and high manufacturing costs presents challenges for integration with compact circuits [4]. To address these limitations, the substrate-integrated waveguide (SIW) presents a more viable approach for integration with cavity-backed CP antennas.

The SIW structure is renowned for its seamless integration with planar circuits, compact design and a high Q-factor [5]. Due to its low conduction loss and compatibility with PCB technology, it offers lower machining costs compared to metal waveguides, which is particularly beneficial for millimeter-wave and higher-frequency applications [5,6,7]. When integrated with a CP antenna instead of a metallic cavity, the SIW structure exhibits multiple advantages [8,9,10,11]. For instance, Ref. [8] reports a wideband CP SIW antenna incorporating a pair of S-dipoles, achieving a 43% −10 dB impedance bandwidth and a 36% 3 dB axial ratio (AR) bandwidth. In [9], a dual-band CP SIW antenna was realized by introducing two annular exponential slots. By exciting two distinct modes, this design achieves dual-band CP performance at 37.5 GHz and 47.8 GHz. Ref. [10] presents a high-gain CP SIW antenna with strip-shaped parasitic patches placed on the surface of the SIW structure, delivering a single-element gain of 10.53 dBi. Additionally, Ref. [11] develops a low-profile CP SIW antenna featuring two orthogonal slots. Leveraging a compact SIW cavity-backed structure, the profile of this antenna is only 0.035 λ₀ (where λ₀ denotes the free-space wavelength at the center frequency).

The above-mentioned single antennas all have excellent performance in some respects. However, in applications involving millimeter waves or even higher frequencies, significant propagation loss and atmospheric absorption will pose challenges and further restrict the communication range [12]. High gain antennas can help to solve these problems, which could be achieved by composing the arrays. In recent years, numerous CP SIW antenna arrays with outstanding performance have been reported, employing techniques like S-dipoles [8], SIW-based sequential rotation [13,14], parasitic patches [10], slot-coupled magneto-electric dipoles [15], SIW-based filtenna [16], SIW resonant cavity [17], among others. These techniques enable the achievement of CP radiation with characteristics like wideband, dual-band, high gain and low profile, respectively.

Differential circuits provide many benefits for microwave applications, when incorporating differential circuits into RF front-end designs, a balun is frequently required to achieve a 180° phase shift [18]. However, this component leads to additional losses and increases circuit complexity. Differential antennas have the capability to remove the balun and can be directly integrated with the front-end circuits. When the CP SIW antennas combine differential circuits, they can further minimize the connection losses and lower the cross-polarization levels [19].

At present, there are some reports on differential CP SIW antenna elements and arrays [20,21,22,23,24]. A new type of laminated resonator CP SIW antenna element was proposed in [20]. By incorporating an anisotropic polarization grid, this antenna element achieves an overlapping bandwidth of 26.16%. In [21], the proposed single antenna element can achieve an effective CP bandwidth of 52% and a low profile of 0.096 λ₀ by employing a laminated resonator design along with two wideband balun feeding networks. In [22], the designed dual-band single antenna element using differential feeding method exhibits the port isolation larger than 75% for CP radiation. Furthermore, a quadri-polarization-agile SIW patch antenna array was proposed in [23], in which CP radiation can reach an AR bandwidth more than 9.8%. According to [24], a high-gain antenna array featuring parabolic reflectors can achieve a maximum gain of 20.5 dBic using a differential approach.

While there are several publications discussing differential-fed antennas, there has been a lack of research specifically addressing the differential CP SIW antenna array. The antenna elements mentioned in [20,21,22] are not appropriate for array construction because they either have multiple ports or require a complicated feeding network for four ports. The established array in [23] has a limited AR bandwidth for CP radiation, or the constructed array in [24] is excessively large. We would like to propose a wideband, high-gain, miniaturized differential antenna array with easy-to-assemble antenna elements and easy-to-integrate feeding networks.

To push forward for the above issues, this paper proposes a novel differential-fed CP SIW cavity-backed slot antenna array that features a wide bandwidth, high gain and compact size. The radiation structure is designed to produce CP radiation through a pair of arc-shaped microstrips. This innovative design allows for the activation of two distinct modes, which helps to expand the frequency range. Furthermore, a modified ring coupler is utilized to generate differential signals. Unlike traditional baluns that rely on balanced resistors for differential signal production, the proposed array can be effectively used in the millimeter wave or Terahertz frequency bands.

The remainder of this paper is organized as follows: Section 2 and Section 3 provide an analysis of the configuration, evolution and principles underlying the antenna element and the array, respectively. Section 4 presents the experimental measurements and corresponding discussions. Section 5 offers the concluding remarks.

2. Antenna Element

2.1. Configuration

The configuration of the proposed CP SIW cavity-backed slot antenna element is shown in Figure 1. The antenna element is made up of two layers of Rogers RO4350 material (ε_r = 3.66, tanδ = 0.004). Both the top layer (Substrate I) and the bottom layer (Substrate II) are of equal thickness, denoted as h₁ = h₂. For the main radiation parts, a pair of arc-shaped microstrips with length l₁ and width w₁ are inserted into Substrate II. They are arranged with a space d₁. Two short pins with a diameter d_s connect the arc-shaped microstrips to enable differential feeding from below. A through hole with a diameter d_h is placed in the center of the antenna element to help keep a good impedance match. A circle of metallic vias with a diameter d and space p apart, are included throughout two layers of the substrates to form the SIW cavity with a diameter C₁. An aperture with a diameter d_R is etched on the top surface of Substrate I for promoting sufficient radiation. Two matching apertures with a diameter d_M are formed in the bottom layer of Substrate II to align the impedance of the feeding ports. The differential ports are labeled as Port + and Port − for feeding purposes. The overall configuration is summarized in

Table 1. Configuration of the proposed antenna element (unit: mm).

Parameter	D1	C1	h1	h2	l1	w1	p
Value	14	10	1.524	1.524	6.47	0.3	1.53
Parameter	d	d1	dR	dh	ds	dM	-
Value	1	5	4.2	1	0.5	1.1	-

.

2.2. Evolution Process

Figure 2 illustrates the evolution process of the proposed antenna element and the various shapes of its radiation parts. Ant. I closely resembles the laminated resonator antenna described in [25]. By leveraging this structure, Ant. I can achieve linear polarized (LP) radiation. The impedance matching of Ant. I is primarily affected by three parameters, the space d₁ between two straight-shaped microstrips, the length l₁ of these microstrips and the thicknesses h₁ and h₂ of the substrates. As depicted in Figure 3a, Ant. I exhibits a narrow impedance bandwidth, and there is limited room for horizontal adjustment. To address this limitation, we proposed using slanted microstrips to broaden the impedance bandwidth, resulting in the development of Ant. II, as shown in Figure 2b. By fine-tuning the length of the slanted microstrips, the bandwidth of Ant. II is significantly enhanced compared to Ant. I. However, as Figure 3b indicates, Ant. II still only provides LP radiation. To simultaneously achieve superior AR characteristics, we introduced a pair of arc-shaped microstrips to replace the slanted ones, leading to the introduction of Ant. III. This new structure enables Ant. III to have a wider impedance bandwidth. Moreover, its AR performance is improved without sacrificing much of the gain bandwidth. To further optimize impedance matching, a through hole was placed at the center of Ant. III [26].

2.3. Working Principle

In the following section, an analysis of the working principle of Antenna III will be presented. Figure 4 displays the horizontal distribution of the E-field inside the SIW cavity at various time intervals corresponding to the two points, f_c1 = 19.0 GHz and f_c2 = 22.0 GHz, obtained using Ansys HFSS.

At the frequency of 19.0 GHz, assuming that the E-field distribution begins in the horizontal plane at phase = 0° when t = 0, the field is concentrated at the ends of the two arc-shaped microstrips along with the plane at phase = 0°, demonstrating the activation of the ${\mathrm{TE}}_{011}^x$ mode, as indicated by the black arrows [25]. At t = T/4, the E-field distribution in the horizontal plane moves to a phase of 90°. The field remains focused at the ends of the two arc-shaped microstrips but now aligns with the plane at phase = 90°, indicating the activation of the ${\mathrm{TE}}_{101}^y$ mode. At t = T/2, the transverse E-field distribution in the SIW cavity resembles that of t = 0, but its direction is reversed, corresponding to the phase = 180°. Likewise, at t = 3 T/4, the transverse E-field distribution in the SIW cavity aligns with that of t = T/4, but again in the opposite direction, corresponding to the phase = 270°. These two sets of orthogonal modes facilitate the generation of left-handed CP (LHCP) radiation.

At the frequency of 22.0 GHz, a different mode can be activated. At time t = 0, the E-field distribution in the horizontal plane is established at a phase of 270°. This is in contrast to the E-field distribution at 19 GHz, where the field is concentrated along two separate arc-shaped microstrips. The E-field is oriented in both positive and negative directions at the same time, which is different from its orientation at 19 GHz, indicating that the ${\mathrm{TE}}_{211}^y$ mode has been activated. At t = T/4, the E-field in the horizontal plane shifts to a phase of 0°, the field is still concentrated along two separate arc-shaped microstrips, but aligns with plane at phase = 0°, signifying the activation of the ${\mathrm{TE}}_{121}^x$ mode. At t = T/2 and t = 3 T/4, the transverse E-field distributions correspond to those at t = 0 and t = T/4, respectively, but they are oriented in the opposite direction. As a result, these two orthogonal modes enable the production of LHCP radiation at this alternate operating frequency.

Figure 5 illustrates the simulated E-field distributions in the CP SIW cavity at frequencies f_c1 = 19.0 GHz and f_c2 = 22.0 GHz, respectively. Due to the circular shape of the cavity, ${\mathrm{TE}}_{011}^x$ and ${\mathrm{TE}}_{101}^y$ modes cannot be differentiated. Consequently, the mode distribution within the cavity is more accurately identified as aligned with the EHM₁₁₁ mode [26,27], as depicted in Figure 5a. At the higher frequency of 22.0 GHz, the ${\mathrm{TE}}_{211}^y$ and ${\mathrm{TE}}_{121}^x$ modes generate the E-field distributions shown in Figure 5b. Similarly, since the cavity of the antenna is circular, it is also difficult to distinguish between the two modes. It is more appropriate to refer to them as the EHM₂₁₁ mode or EHM₁₂₁ mode. The design allows for the simultaneous excitation of two modes within the CP SIW cavity, and these modes interact with one another, resulting in an increased bandwidth of the antenna element.

2.4. Parametric Study

To investigate the impact of antenna element configurations, we carried out a targeted study on these parameters. The analysis presented in the previous section indicates that the arc-shaped microstrips are critical to the performance of this antenna element. As a consequence, the dimensions of the arc-shaped microstrips, specifically the length l₁ and width w₁, as well as the thickness h₁ and h₂ of substrates I and II, which determine the current flow path of the arc-shaped microstrips, are selected.

Figure 6 shows the simulated differential S-parameters of the Port + and Port −, S_dd11, ARs and LHCP gains in the plane defined by phi = 0° and theta = 0° under different arc-shaped microstrips length l₁. As the value of l₁ increases, the impedance bandwidth also increases, depicted in Figure 6a. Specifically, when l₁ is equal to 6.47 mm, the antenna element achieves the largest AR bandwidth from Figure 6b. Hence, when considering the impedance bandwidth, AR bandwidth and LHCP gain bandwidth collectively, the value of l₁ = 6.47 mm demonstrates superior performance, which has the 34.4%−10 dB impendence bandwidth (15.78–22.33 GHz), 20.6% 3 dB AR bandwidth (17.30–21.28 GHz) and 39.9% 3 dB LHCP gain bandwidth (15.07–22.57 GHz), and the overlaid bandwidth is 20.6% from 17.30 to 21.28 GHz.

Figure 7 depicts the simulated S_dd11, ARs and LHCP gains in the plane defined by phi = 0° and theta = 0° under different arc-shaped microstrips width w₁. Unlike the changes seen in length l₁, the impedance bandwidth and AR bandwidth show a trend of first increasing and then decreasing as the width w₁ is altered. In terms of the LHCP gain bandwidth, it diminishes with an increase in the width w₁. However, when evaluating the overlaid bandwidth, the antenna element demonstrates better performance at w₁ = 0.3 mm.

Figure 8 illustrates the simulated S_dd11, ARs and LHCP gains in the plane defined by phi = 0° and theta = 0° for different substrate thicknesses (h₁ and h₂). Based on the principles of the SIW resonant cavity, the height of the cavity can be calculated using the equation presented in [21],

$$f_0={\displaystyle \frac{v_{\mathrm{c}}}{2\sqrt{{\epsilon }_{\mathrm{r}}}}}\sqrt{{\left({\displaystyle \frac{1}{C_1}}\right)}^2+{\left({\displaystyle \frac{1}{h_1+h_2}}\right)}^2}$$

(1)

where v_c is the velocity of light in the free space. Consequently, the overall substrate thickness can be calculated, although it must be evaluated independently for h₁ and h₂. As depicted in Figure 8a, when the ratio of h₁ to h₂ is 1:1, the antenna element can achieve the widest impedance bandwidth. As shown in Figure 8b, the antenna element can achieve the broadest AR bandwidth under this thickness distributions. In contrast, the AR bandwidth deteriorate for the remaining two conditions, even more so for h₁ to h₂ equals 1:3, resulting in the loss of CP radiation characteristics. Regarding the LHCP gain bandwidth, changes in the thickness distribution of the substrate show that an increase in h₁ and a decrease in h₂ lead to a gradual decline in the LHCP gain bandwidth. Adjustments to the thickness distribution impact the LHCP gain bandwidth more significantly than changes to l₁ and w₁.

2.5. Performance

Following the parameter sweep, the antenna element exhibited its optimal performance with a −10 dB impedance bandwidth of 34.4% (from 15.78 to 22.33 GHz), a 3 dB AR bandwidth of 20.6% (from 17.30 to 21.28 GHz) and a 3 dB LHCP gain bandwidth of 39.9% (from 15.07 to 22.57 GHz). The overlaid bandwidth, which accounts for the overlapping sections of the three bandwidth types, is 20.6% from 17.30 to 21.28 GHz.

To verify the radiation properties of the antenna element, Figure 9 presents the radiation patterns in both the E-plane and H-plane at the designated frequencies of 17.8, 19.0 and 20.9 GHz. This includes the LHCP co-polarization and the right-handed CP (RHCP) cross-polarization. The configuration results in symmetrical radiation patterns in both planes, maintaining this symmetry at the specified frequency points, with RHCP levels remaining below −25 dB. The use of a pair of differential feeds enhances the performance of the antenna by reducing electromagnetic interference, which leads to lower cross-polarization and better resolution.

Table 2. Comparisons of the proposed antenna element and the state of the art.

Ref.	Feeding Method	Center Frequency (GHz)	−10 dB Imp. BW (%)	3 dB AR. BW (%)	3 dB Gain BW (%)	Overlaid BW (%)	Size (${\mathbf{\lambda }}_0^3$)	Cross-Pol. (dB)	Peak Gain (dBi/dBic)
[8]	Single	30	43	36.1	38 (1 dB)	36.1	1.0 × 1.0 × 0.315	<−15 *	7.5
[9]	Single	37.5; 47.8	1.1; 1.4	1.1; 1.5	n.a.	n.a.	n.a. × 1.27 × 0.081	<−15 *	5.7
[10]	Single	29.17	24.37	26.14	33.85	20.36	0.83 × 0.83 × 0.20	n.a.	10.53
[20]	Diff	5.09	30.27	26.16	>30.27	26.16	0.53 × 0.53 × 0.389	<−10 *	9.82
[21]	Diff	5.0	65.91	64	52	52	1.33 × 1.33 × 0.115	<−20 *	6.97
[22]	Diff	10.5	>34	n.a.	n.a.	n.a.	0.53 × 0.53 × 0.137	<−10 *	6.0
This work	Diff	19.0	34.4	20.6	39.9	20.6	0.89 × 0.89 × 0.19	−25	8.14

Note: λ₀: The wavelength of the center frequency in the free space. n.a.: Not available in this paper. *: The cross-polarization is not given in this paper directly. It is estimated according to the depicted radiation patterns.

shows the comparisons of the proposed antenna element and the state of the art. It should be noted that the data in this work and those in Refs. [8,10,20,22] presented in the table are obtained through simulations, while the data from other publications for comparison are derived from experiments. The proposed antenna element demonstrates strong performance in terms of bandwidth, size and cross-polarization levels. The use of a differential feeding strategy significantly improves the cross-polarization performance. Additionally, the simple design and compact size of the proposed antenna make it easier to assemble into an array, which are key advantages over the other structures listed in Table 2.

3. Antenna Array

3.1. Configuration of the Antenna Array

Utilizing the proposed antenna elements, a 2 × 2 antenna array was constructed with the aim of improving the gain, as depicted in Figure 10. According to the configuration principle, the spacing L_a between elements in an antenna array should be set at 0.5 λ₀ to λ₀ at the operational frequency [28]. Here we chose the separation distance L of 11.1 mm between the two elements, which is approximately equal to 0.7 λ₀ at the center frequency of 19.0 GHz, ensuring higher gain while lowering the sidelobes in the array.

3.2. Differential Feeding Networks

In order to implement differential feeds, four modified ring couplers, which compose the differential feeding networks, are incorporated into the antenna array, as shown in Figure 10b,c; the exact dimensions of the feeding networks are listed in

Table 3. Configuration of the proposed antenna array and the corresponding feeding networks.

Parameter	h1	h2	h3	dp	dM2	dh	dscrew
Value (mm)	1.524	1.524	0.254	0.7	1.6	1.0	2.0
Parameter	wf1	wb1	La	Lb1	Lb2	Lb3	Lb4
Value (mm)	0.54	28.0	11.1	46.888	10.894	2.1	25.1
Parameter	Lf1	Lf2	Lf3	Lf4	Rf1	Rf2	Rf3
Value (mm)	5.0	2.3	1.61	2.59	2.47	2.18	1.0
Parameter	θf1	θf2	θf3	θf4	-	-	-
Value (deg)	60	90	60	120	-	-	-

. The ring couplers are etched on the bottom side of Substrate III, which has a thickness h₃ of 0.254 mm. The upper surface of Substrate III is coated with copper, functioning as a shared ground for both the antenna array and the ring couplers. To enhance the connectivity between the antenna array and the feeding networks, a copper pad with a diameter d_p of 0.7 mm is incorporated, featuring a concentric circle of the short pin for feeding purposes. In light of the impact of the introduced pads on impedance bandwidth and AR bandwidth, as well as the challenges associated with machining, the diameter of the matching aperture d_M2 is adjusted to 1.6 mm. The diameter of the short pin d_s is maintained at 0.5 mm without any alterations. All the layers are fastened together using plastic screws with a diameter of d_screw.

Since inputting a signal into Port T1 enables the equal power division of two outputs, Port 1+ and Port 1− can keep a phase difference of 180°, and the unbalanced phase difference can be maintained within ±5° across the operation band of the array, as demonstrated through simulations. In the absence of balanced resistors such as a balun, this feeding method can be utilized for wideband operation with a more straightforward design [29].

Figure 11 presents S_dd11, along with ARs and LHCP gains for the conditions involving configurations with and without the feeding networks. The impedance bandwidth has undergone a decrease from 34.4% (15.78–22.33 GHz) to 32.6% (17.28–24.00 GHz), representing a reduction of 1.8%. The 3 dB AR bandwidth has experienced a reduction of 4.2%, from 20.6% (17.30–21.28 GHz) to 16.1% (18.73–22.00 GHz). The LHCP gain bandwidth demonstrates an observed decrease compared with the conditions with and without feeding networks, diminishing from 47.8% (14.14–23 GHz) to 42.9% (14.84–22.94 GHz), indicating a reduction of 4.9%, and the peak gain shifts from 14.3 dBi at 21.50 GHz to 10.4 dBi at 22.13 GHz.

The integration of feeding networks greatly influences the resonance properties of the antenna array [10,24]. However, in the case of the proposed antenna array, these effects can be kept within a reasonable range, enabling efficient feeding of each differential antenna element.

4. Experiments and Discussion

4.1. Prototype and Measurements

A prototype of the proposed antenna array was manufactured, along with the required feeding networks, to verify its performance. Figure 12a displays the components and assembly photos of the manufactured antenna array. In addition, the AR, LHCP gain and radiation patterns of the antenna array were measured in an anechoic chamber, with the measurement setup depicted in Figure 12b. Each antenna element in the proposed array is fitted with a modified ring coupler, and the whole array is powered through a one-to-four power divider during the measurements.

A 4-port Ceyear Network Analyzer 3672D is used to measure the reflection coefficients. As the differential ports of each antenna element are integrated into the array, it is only possible to measure the reflection coefficient of each individual port. For example, measuring Port T1, the reflection coefficient for this port is evaluated while the other three ports are terminated with a 50-ohm load.

Figure 13a displays the simulated and measured reflection coefficients for the excitation of Port T1 in the antenna array. The curves indicate that, while the overall trend is consistent between the simulation and the measurement, there are variations in their positions, as shown by the resonance points and the associated decreases in resonant frequencies. Figure 13b shows the simulated and measured curves for the AR bandwidth and the LHCP gain. The simulated and measured 3 dB AR bandwidths are 16.1% (18.73 to 22.00 GHz) and 13.8% (17.05 to 19.57 GHz), respectively. The simulated and measured 3 dB LHCP gain bandwidths are 42.9% (14.84–22.94 GHz) and 41.8% (15.39 to 23.51 GHz), respectively. The measured peak gain of the proposed antenna array is 10.6 dBi at 23.02 GHz.

Figure 14 shows the simulated and measured radiation patterns of the proposed antenna array in both the E-plane and H-plane at frequencies of 18.7, 19.0 and 19.3 GHz. The measurements for LHCP align more closely with the simulations, while the results for RHCP are higher than the simulated values. The co-polarization appears to be largely symmetric at the 0° radiation direction across all three frequency points. The results indicate that the array can effectively support circular polarization, with the optimal radiation direction occurring in the plane defined by phi = 0° and theta = 0°.

4.2. Inaccuracy Analysis

An analysis of the differences in Figure 13 and Figure 14 between the experimental data and the simulation results was performed. Initial data points were shifted, likely due to inaccuracies in the machining process. As shown in Equation (1), the resonant point was affected by the cavity diameter C₁; even a slight increase in this diameter will lead to a decrease in the resonant frequency. Additionally, although the plastic screws were tightened to their fullest extent during the measurements, the small size of the antenna caused air gaps between the laminated layers. This could lead to an increase in the relative dielectric constant between these layers, thus impacting the resonance point positions. Furthermore, the feeding networks of the antenna array are highly sensitive to bending, which is due to the thinness of the Substrate III. Therefore, it is crucial to handle the antenna array with great care during testing, which added to the complexity of the experiments. Considering these factors, there are discrepancies between simulations and measurements.

4.3. Comparison and Discussion

Table 4. Comparisons of the proposed antenna array and the state of the art.

Ref.	Feeding Method	Center Frequency (GHz)	−10 dB Imp. BW (%)	3 dB AR. BW (%)	3 dB Gain BW (%)	Overlaid BW (%)	Size (${\mathbf{\lambda }}_0^3$)	Peak Gain (dBic/dBi)
[8]	Single	30	27.6	32.7	30	27.6	5.90 × 5.90 × 0.4725	25.2
[10]	Single	29	21.94	22.07	29.03	21.72	6.68 × 6.68 × 0.27	26.1
[13]	Single	10	21	18	9.3	n.a.	3.70 × 3.70 × 0.11	13.2
[14]	Single	20	15.9	13.8	>15.9	13.8	12 × 12.67 × 0.13	25.9
[15]	Single	61	18.2	16.4	>18.2	16.4	6.22 × 6.92 × 0.50	26.1
[17]	Single	6.6175	3.17	0.98	>0.98	0.98	3.69 × 3.69 × 0.035	20.1
[23]	Diff	60.5	11.6	>9.8	>11.6	>9.8	n.a. × n.a. × 0.45	12.2
This work	Diff	19.0	32.6	13.8	41.8	13.8	2.98 × 1.78 × 0.21	10.6

Note: λ₀: The wavelength of the center frequency in the free space. n.a.: Not available in this paper. Bonding films are not included.

summarizes the performance and key metrics of the proposed antenna array in comparison to the state of the art. Among the feeding methods, only the antenna array design in [23] and the one discussed in this paper utilize differential feeds. The design developed in this study shows an enhanced impedance bandwidth, a wider AR bandwidth, a greater gain bandwidth and a lower profile, which were achieved using a simple microstrip differential feeding technique. Although the structures reported in [13,14,17], they all have the higher gains, the bandwidth is narrower than the structure proposed in this paper. The structures reported in [8,10,15], they have wider AR bandwidths and higher gains, but they all have larger array size, which may have the limited space when they are applied in the compact front-end circuits.

In summary, this design employs a differential feeding approach that enhances the performance of the antenna array, leading to a simpler structure, improved bandwidth and a lower profile. It is important to mention that in practical applications, the ring coupler responsible for splitting power and generating differential signals can be eliminated, enabling direct integration into the front-end circuits.

5. Conclusions

This research advances the development of differential CP SIW cavity-backed slot antenna arrays. Compared to existing studies mentioned in this paper, the proposed array demonstrates broad performance, high gain, a straightforward feeding approach and a compact, low-profile design, making it particularly well-suited for advancements in compact integrated circuits, specifically, such as 5G/6G millimeter-wave communication, high-resolution imaging antennas for spaceborne/airborne synthetic aperture radar (SAR), satellite communication and navigation systems, the Internet of Things (IoT) and intelligent sensor networks, as well as special scenarios with strict electromagnetic compatibility (EMC) requirements. Furthermore, adjusting the configuration of the arc-shaped microstrips enables the achievement of dual-band CP radiation and reconfigurable radiation, among other features. This adaptability allows for the selection of operational modes tailored to various application scenarios, representing a significant advantage of this antenna type. This adaptability enables this type of antenna to select a working mode suitable for the above application scenarios, which is also a significant advantage of this antenna type.