Low-Profile Metasurface Antenna for Broadband RCS Reduction and Omnidirectional Radiation

Abstract

A low-profile, low radar cross-section (RCS) omnidirectional metasurface antenna is investigated and proposed in this letter. The antenna consists of a top circular patch, a three-layer dielectric substrate, a full metal ground, a multi-layer polarization conversion metasurface, and four short vias for connecting the top patch to the ground. Wideband impedance matching is achieved by modifying an F-shaped feeding structure. The broadband RCS reduction is realized by loading the antenna with the polarization conversion metasurface (PCM) in an appropriate array configuration. The antenna prototype has been fabricated and measured in an anechoic chamber. Experimental results illuminated that the antenna features a low profile of 0.051λ₀ (λ₀ is the wavelength at 2.35 GHz) and a 10 dB impedance bandwidth of 2.11–2.62 GHz (a fractional bandwidth of 21.56%). Significantly broadband RCS reduction is achieved from 7.05 to 16.96 GHz, with a maximum reduction of –28 dB and an average reduction of –12.51 dB.

1. Introduction

The omnidirectional antenna can be found in many areas such as aviation communication, indoor base stations, satellite communication, navigation system, and tracking applications due to their simple structure and vast radiation pattern. Yet, the traditional omnidirectional antennas could compromise the low-scattering characteristics of the platform, significantly degrading the overall stealth. Therefore, the antenna with low radar cross-section (RCS) and omnidirectional radiation has become increasingly important for fighter aircraft.

Up to now, numerous researchers have paid attention to the low-profile omnidirectional antennas [1,2,3,4,5,6,7,8,9,10,11,12,13,14,15,16,17,18,19,20,21,22,23,24,25,26]. A low-profile, omnidirectional circularly polarized (CP) patch antenna was proposed in [1]. This design utilizes two monopole modes of a circular patch, connected to a modified ground plane via a set of conductive vias to achieve wideband impedance matching. Curved branches introduced at the edge of the circular patch excite degenerate modes (TM₀₁ and TM₀₂), generating circularly polarized waves. This antenna features a low profile of 0.024λ (3 mm), a 10 dB impedance bandwidth of 19.8%, and a 3 dB axial ratio bandwidth of 19.3%. It is true that the radiation performance is easily affected by the carrier due to the modified ground plane. However, the metal ground planes in these antennas are defective, which easily leads to electromagnetic leakage in practical applications and affects system compatibility. Conversely, a full metal ground plane often results in a significant increase in RCS for an omnidirectional antenna.

As a momentous scheme, the metasurfaces have been introduced to reduce the RCS of antennas [27,28,29]. Metasurfaces are ultrathin, planar artificial materials composed of subwavelength-structured unit cells, which can flexibly manipulate the amplitude, phase, polarization, and propagation direction of electromagnetic waves with unprecedented precision [30,31,32,33,34]. In [27], several resistors were loaded into the metasurface to achieve absorption. The RCS reduction can be achieved by amplitude manipulation. In [28], a transmissive polarization conversion metasurface was properly arrayed and loaded above the antenna. A multilayer transmissive metasurface was designed in [29], where the bottom layer utilizes a frequency-selective surface to achieve high transmission efficiency in the operating band and RCS reduction in two out-of-band frequency ranges. Yet, these antennas all radiate a normal beam with low RCS. Different from the existing literature, a low-profile, low-RCS omnidirectional antenna has been proposed and designed with a full-metal ground. An F-shaped feeding structure was realized to achieve Wideband impedance matching. The antenna device has been fabricated based on a 6 mm PCB board. Measurement demonstrates that the proposed antenna achieves a low profile of 0.051λ₀ (λ₀ is the wavelength at 2.35 GHz) and a 10 dB impedance bandwidth of 2.11–2.62 GHz (a fractional bandwidth of 21.56%). Significant RCS reduction is obtained from 7.05 to 16.96 GHz with a maximum reduction of −28 dB and an average in-band reduction of −11.3 dB. The proposed metasurface antenna has potential applications on the stealth airborne platforms.

2. Antenna Design

2.1. Antenna Configuration

Figure 1 illustrates the structure of the omnidirectional antenna, which consists of three dielectric layers, a top circular patch, a full metal ground, three-layer polarization conversion metasurfaces, and four metalized vias for connecting the top patch to the ground. Figure 2 shows the configuration of the proposed polarization conversion metasurface (PCM) unit, composed of two dielectric layers and three metal patches. The proposed metasurface antenna is fabricated on a Rogers RT5880 substrate. The key material parameters are: a relative permittivity (εr) of 2.20 and a loss tangent (tanδ) of 0.001. The thickness of the PCM unit is equal to the total thickness of the antenna (H1 + H3 + H4 = H2 + H4 = 6 mm). Furthermore, the thickness of the upper dielectric layer of the PCM unit equals the combined thickness of the upper two dielectric layers of the antenna (H2 = H1 + H3 = 3.7 mm). Consequently, the upper two metallic layers of the PCM unit are, respectively, loaded onto the top and third dielectric layers of the antenna.

Figure 1c shows the top-layer structure of the antenna. In the center, there is a circular patch with radius R2. This patch is connected to the bottom ground plane through four metalized vias with a radius of R3. The top-layer metallic patches of the PCM unit are loaded around the central circular patch. As shown in Figure 1d,e, the central patches on both the second and third layers are trapezoidal metal structures, designated as Trapezoidal Patch-1 and Trapezoidal Patch-2, respectively. The length of these trapezoidal patches is L7. The other length of Trapezoidal Patch-1 is denoted as T1 and T2. Similarly, T3 and T4, respectively, denote the other length of Trapezoidal Patch-2. These two trapezoidal patches are connected by a feeding probe of an SMA port for the antenna. The second-layer metallic patches of the PCM unit are disposed around Trapezoidal Patch-2. The feeding port of the antenna is located at the bottom.

As shown in Figure 3, parametric sweeps of R2 and R3 were performed according to the variation in the antenna S11 parameter, and the final values were determined as R2 = 25.6 mm and R3 = 1.70 mm. Similarly, as illustrated in Figure 4, parametric sweeps of L1 and P for the PCM unit were conducted based on the variation of its polarization conversion ratio, and the optimized parameters were determined as L1 = 2.9 mm and P = 7.0 mm.

2.2. Feeding Structure

To clarify the F-shaped feeding structure of the antenna, its design procedure and corresponding results are analyzed, as specifically illustrated in Figure 5 and Figure 6. The traditional coaxial feeding structure is shown in Figure 5a. The feeding probe directly excites the top circular patch, resulting in poor impedance matching in Figure 5a. To achieve a wide bandwidth, a multi-layer configuration of Γ- and F-shaped feeding structures was explored, as depicted in Figure 5b,c. In the design, they are denoted as Structure 2 and Structure 3. The feeding probe excites the metal patches on the intermediate layers. As shown in Figure 6a, a comparison of the real and imaginary parts of impedance reveals that the inductive reactance of the antenna gradually decreases as the number of layers increases for the three feeding structures. Observation of the surface current distribution in Figure 7 reveals that the F-shaped feed effectively enhances the surface current intensity of the antenna. Figure 6b shows a comparison of the S₁₁ curves of the three structures. The results indicate that the employment of an F-shaped feed increases the resonant frequency of the antenna, converting it from a single-resonance mode to a cascade of multi-resonance modes, thereby broadening the radiation bandwidth. Therefore, F-shaped feeding of Structure 3 is employed in this paper.

2.3. PCM Array Design

The simulated co-polarized and cross-polarized reflection magnitude and phase of the unit are shown in Figure 8a. The magnitude of co-polarized reflection (Rxx and Ryy) is less than −10 dB, and that of cross-polarized reflection (Ryx and Ryx) is near 0 dB from 7.05 to 16.96 GHz, which indicates that the unit achieves polarization rotation in the band.

For broadband RCS reduction based on a chessboard array, a pair of units must be constructed so that the amplitude of the cross-polarized reflection is equal and their phase difference is 180°. According to Pancharatnam–Berry phase theory [35], when a right-handed circularly polarized (RHCP) wave is incident along the -z-axis, the incident electric field vector ${\overrightarrow{E}}^i(\omega )$ and the reflected electric field vector ${\overrightarrow{E}}^r(\omega )$ can be expressed as:

$${\overrightarrow{E}}^i(\omega )=E_0(\omega )\overrightarrow{e_x}-j{E}_0(\omega )\overrightarrow{e_y}$$

(1)

$${\overrightarrow{E}}^r(\omega )=R_x{E}_0(\omega )\overrightarrow{e_x}+R_y{E}_0(\omega )\overrightarrow{e_y}$$

(2)

R denotes the reflection efficiency of the unit, and Φ represents the phase of the incident wave. Subsequently, the reflection coefficients R_x and R_y can be expressed as follows:

$$\left\{\begin{array}{c}{R}_x=\left|r_{xx}\right|e^{-j{Ф}_{xx}}+\left|r_{yx}\right|e^{-j{Ф}_{yx}}\\ R_y=\left|r_{yy}\right|e^{-j{Ф}_{yy}}+\left|r_{xy}\right|e^{-j{Ф}_{xy}}\end{array}\right.$$

(3)

When the unit is rotated by φ with respect to the y-axis, the incident electric field vector ${\overrightarrow{E}}^{ir}$ and the reflected electric field vector ${\overrightarrow{E}}^{rr}$ after rotation can be expressed in the u-v relative coordinate system as follows:

$$\begin{array}{c}\left(\begin{array}{c}{\overrightarrow{E}}^{ir}(\omega )\\ {\overrightarrow{E}}^{rr}(\omega )\end{array}\right)=E_0(\omega )\left(\begin{array}{cc}1& -j\\ R_u& -j{R}_v\end{array}\right)\left(\begin{array}{cc}\cos \phi & -\sin \phi \\ \sin \phi & \cos \phi \end{array}\right)\left(\begin{array}{c}\overrightarrow{e_u}\\ \overrightarrow{e_v}\end{array}\right)\\ =E_0(\omega )\left(\begin{array}{cc}{e}^{-j\phi }& -j{e}^{-j\phi }\\ R_u{e}^{-j\phi }& -j{R}_v{e}^{-j\phi }\end{array}\right)\left(\begin{array}{c}\overrightarrow{e_u}\\ \overrightarrow{e_v}\end{array}\right)\end{array}$$

(4)

Since the incident wave is a circularly polarized wave, the variation trends of the reflection efficiency and reflection amplitude along the u-axis in the relative coordinate system are consistent with those along the x-axis in the original coordinate system. This indicates that the reflection amplitude R_u and phase Φ_u satisfy R_u = R_x and Φ_u = Φ_x. By analogy, the variation trends along the v-axis in the relative coordinate system are consistent with those along the y-axis in the original coordinate system. Similarly, the reflection amplitude R_v and phase Φ_v satisfy R_v = R_y and Φ_v = Φ_y. By transforming the relative coordinates back to the original coordinate system, the expression of the reflected wave ${\overrightarrow{E}}^{rr}$ after rotation in the x–y coordinate system can be obtained as follows:

$$\begin{array}{c}{\overrightarrow{E}}^{rr}(\omega )=E_0(\omega ){\left(\begin{array}{c}{R}_u{e}^{-j\phi }\\ -j{R}_v{e}^{-j\phi }\end{array}\right)}^T\left(\begin{array}{cc}\cos \phi & \sin \phi \\ -\sin \phi & \cos \phi \end{array}\right)\left(\begin{array}{c}\overrightarrow{e_x}\\ \overrightarrow{e_y}\end{array}\right)\\ ={\displaystyle \frac{E_0(w)}{2}}{\left(\begin{array}{c}\left(R_x(w)-R_y(w)\right)e^{-j2\phi }\\ R_x(w)+R_y(w)\end{array}\right)}^T\left(\begin{array}{c}\overrightarrow{e_x}+j\overrightarrow{e_y}\\ \overrightarrow{e_x}-j\overrightarrow{e_y}\end{array}\right)\end{array}$$

(5)

According to the above formula, it can be seen that ${\overrightarrow{E}}^r$ can be divided into two components, namely the left-handed circularly polarized component ${\overrightarrow{E}}_{(LHCP)}^r$ and the right-handed circularly polarized component ${\overrightarrow{E}}_{(RHCP)}^r$, as shown below:

$${\overrightarrow{E}}^r(\omega )={{\overrightarrow{E}}^r}_{(LHCP)}(\omega )+{{\overrightarrow{E}}^r}_{(RHCP)}(\omega )$$

(6)

$$\left(\begin{array}{c}{\overrightarrow{E}}_{(LHCP)}^r(\omega )\\ {\overrightarrow{E}}_{(RHCP)}^r(\omega )\end{array}\right)={\displaystyle \frac{E_0(\omega )}{2}}\left(\begin{array}{c}(\left|R_x(\omega )\right|e^{j{Ф}_x}-\left|R_y(\omega )\right|e^{j{Ф}_y})(\overrightarrow{e_x}+j\overrightarrow{e_y})e^{-j2\phi }\\ (\left|R_x(\omega )\right|e^{j{Ф}_x}+\left|R_y(\omega )\right|e^{j{Ф}_y})(\overrightarrow{e_x}-j\overrightarrow{e_y})\end{array}\right)$$

(7)

When the reflection efficiency and phase satisfy |R_x(ω)| = |R_y(ω)| = |R(ω)| and |ΔΦ| = |Φ_x − Φ_y| = π, Equation (7) can be expressed as follows:

$$\begin{array}{c}\left(\begin{array}{c}{\overrightarrow{E}}_{(LHCP)}^r(\omega )\\ {\overrightarrow{E}}_{(RHCP)}^r(\omega )\end{array}\right)={\displaystyle \frac{1}{2}}R(\omega )E_0(\omega )\left(\begin{array}{c}{e}^{-j2\phi }(1-e^{j(\pm \pi )})\left(\overrightarrow{e_x}+j\overrightarrow{e_y}\right)\\ (1+e^{j(\pm \pi )})\left(\overrightarrow{e_x}-j\overrightarrow{e_y}\right)\end{array}\right)\\ =R(\omega )E_0(\omega )\left(\begin{array}{c}\left(\overrightarrow{e_x}+j\overrightarrow{e_y}\right)e^{-j2\phi }\\ 0\end{array}\right)\end{array}$$

(8)

From Equations (1)–(8), it can be seen that when a right-handed circularly polarized (RHCP) wave is incident, only the left-handed circularly polarized component undergoes a phase change of −2φ.

Based on this principle, three array configurations are designed to reduce the radar cross-section (RCS) of antennas. As shown in Figure 6b, the phase difference is about 180° between the PCM unit cell and that with 90° rotation.

As shown in Figure 9, three configurations of omnidirectional antennas loaded with the PCM array are designed in this work, namely the chessboard array, the hexagonal array, and the octagonal array. Figure 10 illustrates that the characteristics of impedance matching, gain, and radiation patterns are hardly affected by PCM with the three array configurations. The monostatic RCS results of the antennas are presented in Figure 11. The calculated values of average RCS reduction between the reference and the metasurface antennas are listed in

Table 1. Average RCS reduction.

Configurations	Average RCS Reduction from 7.05 to 16.96 GHz
	x-Polarization	y-Polarization
Configuration 1	11.45	11.30
Configuration 2	10.57	10.48
Configuration 3	12.52	12.28

in the band from 7.05 to 16.96 GHz. The metasurface antennas achieved average RCS reductions of 11.45, 10.57, and 12.52 dB under x-polarized incidence and 11.30, 10.48, and 12.28 dB under y-polarized incidence. It can be found that the most RCS reduction can be obtained for the metasurface antenna with configuration 3. In addition, simulations were performed to investigate the RCS reduction performance of the proposed antenna at oblique incidences of 15° and 30°. As illustrated in Figure 12, favorable RCS reduction characteristics are still maintained under oblique incidence conditions. To verify that PCM has little effect on the radiation performance of the antenna, we compared the radiation efficiency of the reference antenna and configuration 3. As shown in Figure 13, the radiation efficiency of the omnidirectional antenna does not change significantly after loading PCM. As illustrated in Figure 14, the cross-polarized radiation pattern of the antenna at 2.2 GHz is adopted for validation. It is shown that PCM has no obvious effect on increasing the cross-polarization level of the antenna.

3. Measured Results

As shown in Figure 15, a prototype of the proposed metasurface antenna loaded with a PCM array of configuration 3 has been fabricated using PCB technology. The metasurface antenna with dimensions of π × 62.5² mm² × 6 mm was measured in an anechoic chamber. Figure 16a,b presents the measured and simulated results of S-parameters and gain of the antenna. The experimental results indicate that a working bandwidth covering 2.11–2.62 GHz (a fractional bandwidth of 21.56%) can be obtained with an amplitude of S11 less than −10 dB, and the gain is more than 2.5 dB for the proposed metasurface antenna. The measurement is in proper agreement with the simulation. The measured radiation patterns at the two resonance frequencies of 2.2 and 2.5 GHz are given in Figure 17. The measured results indicate that the antenna exhibits low cross-polarized components of less than −40 dB and the favorable omnidirectional characteristics with a gain variation of less than 3 dB. From the measured results of RCS reduction in Figure 18, we can obviously see that the RCS reduction above 8 dB can be achieved from 7.05 to 16.96 GHz for the metasurface antenna in normal incidences with x- and y-polarization.

A comparison is summarized in

Table 2. Comparison between the proposed metasurface antenna.

Ref.	Size (λ03)	Bandwidth (GHz)	Full Ground	RCS Reduction (GHz)	Omnidirectional Radiation	Polarization for RCS Reduction	Azimuthal Gain (dBi)
[36]	0.42 × 0.42 × 0.03	2.05–2.75 (29.3%)	√	×	√	×	1.5
[37]	π × 0.322 × 0.06	3.2–4.0 (22.2%)	√	3.0–7.0	√	Dual-polarization	4.5
[38]	π × 0.352 × 0.009	1.95–2.62 (29.3%)	√	×	√	×	3.2
[39]	0.85 × 0.85 × 0.1	VHF-UHF (1700%)	√	7.2–17.2	√	Dual-polarization	2.0
[40]	π × 0.552 × 0.04	Dual-band (18.5%/23.6%)	√	×	√	×	4.6/6.8
[41]	π × 0.362 × 0.05	6.38–17.67 (85.6%)	√	×	×	×	×
[42]	π × 0.152 × 0.04	1.66–5.04 (7.2%)	√	×	√	×	1.55
[43]	0.9 × 0.9 × 0.07	1.66–5.04 (95.8%)	√	×	√	×	5.8
[44]	π × 0.352 × 0.01	2.99–3.01 (1%)	√	2.99–3.01	√	Single-polarization	5.7
[45]	π × 1.222 × 0.45	1.6–3.8 (81.5%)	×	×	×	×	6.1
Our design	π × 0.512 × 0.05	2.11–2.6 (21.56%)	√	7.05–16.96	√	Dual-polarization	3.5

λ₀ is the wavelength at the center frequency.

between the proposed metasurface antenna and several existing low-profile, low RCS antennas [36,37,38,39,40,41,42,43,44,45]. Compared with antennas in [37,39,40,41,42,44], the proposed design achieves an ultra-low profile of 0.05λ₀. The full metal ground of the proposed antenna decreases the effect of the carrier compared with the antenna in [41]. As shown in Table 2, this metasurface antenna successfully realizes the omnidirectional radiation and RCS with the advantage of ultra-low profile, a relatively small gain variation, and low cross-polarized components.

4. Conclusions

In this letter, we present a low-profile, broadband, low RCS omnidirectional metasurface antenna. By feeding the metal patch on the intermediate layer of the antenna, a low-profile height of 6 mm is successfully achieved. The broadband RCS reduction is realized by loading the antenna with a polarization conversion metasurface array in an octagonal configuration. Measurement and simulation demonstrated that the proposed metasurface antenna exhibits advantages of omnidirectional radiation, low RCS, stable gain, and low cross-polarized components. Owing to its characteristics, this metasurface antenna offers a potential application on a stealth aircraft.