Design and Analysis of a Bearing-Integrated Rotary Transformer

Abstract

In this paper, a bearing- and transformer-integrated electric excitation synchronous motor excitation system (bearing-integrated rotary transformer) is proposed to support the motor rotor and energy transmission of excitation systems. Firstly, the working principle of the bearing-integrated rotary transformer is discussed. Secondly, the structure and electromagnetism of the bearing-integrated rotary transformer are designed through the processes and principles of pole slot matching, stator/rotor size, winding, and the magnetic regulating needle. Thirdly, the bearing-integrated rotary transformer undergoes an electromagnetic–thermal simulation. Finally, a prototype of the bearing-integrated rotary transformer is manufactured, and the electromagnetic and transmission characteristics are tested, verifying the correctness of the proposed scheme and providing additional ideas for the improvement of synchronous motor excitation systems.

1. Introduction

With the emergence of new energy vehicles in the car industry, the influence of internal motor vehicle running performance has become increasingly important for the stable operation of these vehicles. Therefore, drive systems using brushless technology have become an area of focus for researchers. Electrically excited synchronous motors are suitable for electric vehicles, mining machinery, and other fields because they do not use rare earth materials, are simple to control, and have excellent anti-vibration performance, among other advantages [1,2]. However, traditional electric excitation systems rely on their slip ring–brush structure for excitation energy transmission, which has problems, such as commutation spark, mechanical wear, and high maintenance costs; this seriously restricts the improvement of reliability and integration development of the motor.

Research into AC motor brushless technology can be roughly divided into three directions: One is using permanent magnet excitation technology to achieve a brushless mechanism, though this includes the risk of permanent magnet high-temperature demagnetization, material cost fluctuations, and weak magnetic control problems under wide-ranging speed regulation conditions [3]. The second direction is achieving a brushless mechanism through additional winding or an additional exciter [4], such as the two-stator motor, through the introduction of auxiliary excitation winding, though this can lead to a 20–30% increase in the axial length and significantly reduced power density. For example, T.F. Han proposed the realization of motor excitation based on the theory that the motor’s structure can combine the secondary magnetic field of the induction motor with the connection between the stator and the rotor winding [5]. The third excitation scheme draws on the adjustability of the magneto-electric excitation synergy magnetic field [6]. In one study [7], which evaluated a new type of hybrid excitation brushless motor, the scheme greatly improved the motor by reducing motor loss and the cost of using a permanent magnet while increasing mechanical strength. Mizuno. T proposed a new type of hybrid excitation motor with the structure for the two axial parts connected by a magnetic yoke. The motor rotor is also divided into two parts, and the magnetic pole is alternately distributed by the permanent magnet pole and the iron core pole [8]. Recently, the modulated brushless motor has made breakthroughs with spatial harmonics, such as the dual rotor structure proposed by Zheng Ping’s team to increase the torque density to 18 kN·m/m³; however, the number of permanent magnets still accounts for more than 15% of the motor’s total mass [9].

In the domain of contactless energy transmission, rotary transformer technology offers a novel approach to brushless energy transfer. Conventional rotary excitation transformers employ a split-type design with an independent bearing system and occupy an axial space of up to 25%. The disk partition structure proposed by Aydin. M reduces the axial dimension by 12%; however, the increase in air gap reluctance causes a rise in magnetic leakage by up to 18% [10].

Therefore, this project proposes replacing the brush and slip ring of the traditional electric excitation synchronous motor with a bearing-integrated rotary transformer. Firstly, through topological reconstruction, the magnetic adjustment module is integrated with the bearing support function, reducing the number of components by 40% compared to the split-type design presented in [10]. Secondly, three-dimensional magnetic field modulation technology is employed to decrease the harmonic distortion rate of the air-gap magnetic flux density, thereby enhancing transmission efficiency relative to the approach described in [7]. More significantly, this design eliminates the independent slip-ring structure entirely by reusing electro-mechanical functions and compressing the axial installation space to one-third of that required by the conventional brush system while maintaining energy transfer efficiency.

2. Characteristic Analysis of Bearing-Integrated Rotary Transformer

Firstly, the structure and excitation principle of the bearing-integrated rotary transformer is introduced. Secondly, based on the theory of the double rotating magnetic field and magnetic field modulation theory, we explore the equivalent magnetic circuit model, incomplete equivalent circuit model, and complete equivalent circuit model.

2.1. The Basic Structure and Excitation Principle of a Bearing-Integrated Rotary Transformer

In this paper, the electric excitation synchronous motor of electric vehicles is the research object. The third harmonic excitation principle of the motor is shown in Figure 1: The stator core and winding of the bearing-integrated rotary transformer are fixed at the stator end of the synchronous motor. The rotor core and its winding are assembled on the rotating shaft and rotate with the motor’s rotor. The excitation of the bearing-integrated rotary transformer uses a single-phase current. The excitation current and armature current share a set of DC buses. The excitation current is the third harmonic current and is provided by the two-level inverter, filter capacitor, bearing-integrated rotary transformer, and rectifier.

2.2. Equivalent Magnetic Circuit Model

To explain the relationship between the air gap magnetic field pole pairs, speed, and output voltage in the bearing-integrated rotary transformer, the following assumptions were made.

(1) The pole pairs of the stator air gap magnetic field, the number of needle rollers in the modulation ring, and the pole pairs of the rotor air gap magnetic field are p_s, p₀, and p_r, respectively.

(2) The stator air gap magnetic field speed is n_s. The rotor speed is n_r.

By introducing a single-phase sinusoidal current I₁ into the stator winding, the fundamental pulsating magnetomotive force established in the air gap is the following:

$$f_{\mathrm{p}}(\varphi ,t)=F_{\mathrm{pm}}\cos (p_{\mathrm{s}}\varphi )\cos ({\omega }_{\mathrm{p}}t)$$

(1)

$$F_{\mathrm{pm}}={\displaystyle \frac{4}{\pi }}{\displaystyle \frac{\sqrt{2}}{2}}{I}_1{N}_{\mathrm{c}}q{K}_{\mathrm{d}1}$$

(2)

In the formula, F_pm is the amplitude of the fundamental magnetomotive force; ω_p is the angular frequency; K_d1 is the winding coefficient; Nc is the number of turns performed in winding; q is the number of coils; t is time; and $\varphi$ is the mechanical angle.

Using the triangular formula, this can be changed into the following:

$$\begin{array}{l}f(\varphi ,t)\\ =F_{\mathrm{p}\mathrm{m}}\cos (p_{\mathrm{s}}\varphi )\cos ({\omega }_{\mathrm{p}}t)\\ ={\displaystyle \frac{1}{2}}{F}_{\mathrm{p}\mathrm{m}}\cos \left({\omega }_{\mathrm{p}}t-p_{\mathrm{s}}\varphi \right)+{\displaystyle \frac{1}{2}}{F}_{\mathrm{p}\mathrm{m}}\cos \left({\omega }_{\mathrm{p}}t+p_{\mathrm{s}}\varphi \right)\\ =f_{\mathrm{p}}(\varphi ,t)+f_{\mathrm{q}}(\varphi ,t)\end{array}$$

(3)

The first term on the right side of the equation is the forward rotating magnetic potential, and the second term is the reverse rotating magnetic potential. Therefore, the pulsating magnetic potential can be divided into two rotating magnetic potentials with the same rotational speed and opposite direction. The amplitude of each rotating magnetic potential is half the amplitude of the pulsating magnetic potential.

If the drop in magnetic pressure in the stator and rotor core is neglected without considering the influence of the stator and rotor cogging, the magnetic circuit has a certain ideal magnetoresistance characteristic, and the magnetic circuit is considered linear. In this magnetic circuit, the magnetic derivative per unit area remains constant in the inner and outer air gaps, while the permeability in the air gap between the stator and rotor teeth is 0.

The number of teeth in the rotor is N_r; the ratio of the width of the magnetic needle to the pitch is K (pole arc coefficient); the height of the air gap is h; and the function expression of the air gap permeability per unit area is the following:

$${\lambda }_{\mathrm{h}}(\varphi ,t)={\lambda }_0+{\lambda }_1·\cos \left[N_{\mathrm{r}}\left(\varphi -{\theta }_{\mathrm{r}0}-{\omega }_{\mathrm{r}\mathrm{m}}t\right)\right]$$

(4)

$${\lambda }_0=K{\displaystyle \frac{{\mu }_0}{h}}$$

(5)

$${\lambda }_1=2\left({\displaystyle \frac{\sin \left(K\pi \right)}{\pi }}\right){\displaystyle \frac{{\mu }_0}{h}}$$

(6)

In the formula, ${\lambda }_0$ is the air gap permeability fundamental component; ${\lambda }_{\mathsf{\nu }}$ is the $\nu$ order permeability harmonic amplitude; ${\theta }_{\mathrm{r}0}$ is the angle between the rotor and the synthetic magnetomotive force axis; ω_rm is the rotor speed; and ${\mu }_0$ is the vacuum permeability.

The magnetic flux density, B, is expressed as the product of the magnetomotive force and the air gap permeability function. The air gap flux density of the bearing-integrated rotary transformer is as follows:

$$\begin{array}{l}B(\varphi ,t)=f(\varphi ,t)·{\lambda }_{\mathrm{h}}(\varphi ,t)\\ =\left[f_{\mathrm{p}}(\varphi ,t)+f_{\mathrm{q}}(\varphi ,t)\right]·{\lambda }_{\mathrm{h}}(\varphi ,t)\\ =\left[{\displaystyle \frac{1}{2}}{F}_{\mathrm{p}\mathrm{m}}\cos \left({\omega }_{\mathrm{p}}t-p_{\mathrm{s}}\varphi \right)+{\displaystyle \frac{1}{2}}{F}_{\mathrm{p}\mathrm{m}}\cos \left({\omega }_{\mathrm{p}}t+p_{\mathrm{s}}\varphi \right)\right]\\ \times \left[{\lambda }_0+{\lambda }_1·\cos N_{\mathrm{r}}\left(\varphi -{\theta }_{\mathrm{r}0}-{\omega }_{\mathrm{r}\mathrm{m}}t\right)\right]\\ =B_{\mathrm{p}0}+B_{\mathrm{q}0}+B_{\mathrm{p}1(+)}+B_{\mathrm{p}1(-)}+B_{\mathrm{q}1(+)}+B_{\mathrm{q}1(-)}\end{array}$$

(7)

In this formula,

$$B_{\mathrm{p}0}={\displaystyle \frac{1}{2}}{F}_{\mathrm{p}\mathrm{m}}{\lambda }_0\cos \left({\omega }_{\mathrm{p}}t-p_{\mathrm{s}}\varphi \right)$$

(8)

$$B_{\mathrm{q}0}={\displaystyle \frac{1}{2}}{F}_{\mathrm{p}\mathrm{m}}{\lambda }_0\cos \left({\omega }_{\mathrm{p}}t+p_{\mathrm{s}}\varphi \right)$$

(9)

$$B_{\mathrm{p}1(+)}={\displaystyle \frac{1}{2}}{F}_{\mathrm{p}\mathrm{m}}\left({\lambda }_1/2\right)\times \cos \left[\left(p_{\mathrm{s}}+N_{\mathrm{r}}\right)\varphi -\left({\omega }_{\mathrm{p}}+N_{\mathrm{r}}{\omega }_{\mathrm{r}\mathrm{m}}\right)t-N_{\mathrm{r}}{\theta }_{\mathrm{r}0}\right]$$

(10)

$$B_{\mathrm{p}1(-)}={\displaystyle \frac{1}{2}}{F}_{\mathrm{p}\mathrm{m}}\left({\lambda }_1/2\right)\times \cos \left[\left(p_{\mathrm{s}}-N_{\mathrm{r}}\right)\varphi -\left({\omega }_{\mathrm{p}}-N_{\mathrm{r}}{\omega }_{\mathrm{r}\mathrm{m}}\right)t+N_{\mathrm{r}}{\theta }_{\mathrm{r}0}\right]$$

(11)

$$B_{\mathrm{q}1(+)}={\displaystyle \frac{1}{2}}{F}_{\mathrm{p}\mathrm{m}}\left({\lambda }_1/2\right)\cos \times \left[\left(N_{\mathrm{r}}-p_{\mathrm{s}}\right)\varphi -\left({\omega }_{\mathrm{p}}+N_{\mathrm{r}}{\omega }_{\mathrm{r}\mathrm{m}}\right)t-N_{\mathrm{r}}{\theta }_{\mathrm{r}0}\right]$$

(12)

$$B_{\mathrm{q}1(-)}={\displaystyle \frac{1}{2}}{F}_{\mathrm{p}\mathrm{m}}\left({\lambda }_1/2\right)\times \cos \left[\left(N_{\mathrm{r}}+p_{\mathrm{s}}\right)\varphi -\left({\omega }_{\mathrm{p}}+N_{\mathrm{r}}{\omega }_{\mathrm{r}\mathrm{m}}\right)t-N_{\mathrm{r}}{\theta }_{\mathrm{r}0}\right]$$

(13)

In the above formula, the magnetomotive force of the stator winding is divided into six components via the flux-adjusting needle’s modulation of the magnetic field, and their amplitudes are different; this is related to the number of poles in the stator, the frequency of the excitation current, the distribution of needle permeability, and the speed. These six magnetic flux density components with various speeds and poles induce six different electromotive forces into the rotor winding. It can be seen from the principle of the motor that B_p0 and B_q0 are independent of the motor speed and can only generate an induced electromotive force. The latter four magnetic flux density components are related to the motor speed, resulting in a moving electromotive force.

2.3. Equivalent Circuit Model

When the bearing-integrated rotary transformer runs without a load (the rotor side winding is open), stator winding is added with a sinusoidal excitation current I₁, and a pulsating magnetomotive force with an amplitude of F_pm is generated in the air gap. The frequency is f₁, and the amplitude is proportional to the current I₁. The rotor winding is cut to generate the induced potential E₂, which is expressed by the stator winding current I₁ and the excitation reactance X_m:

$$E_2=-j{I}_1{X}_{\mathrm{m}}=-j\sqrt{2}\pi f_1{N}_2{\varphi }_{\mathrm{m}}$$

(14)

Through the decomposition of the symmetrical component method, the magnetomotive force of the forward magnetic field and the reverse magnetic field is F_m/2, which is proportional to I₁/2. Both cut the rotor winding with frequency f₁, and the induced potential generated in the rotor winding is E_s1 and E_s2. When the rotor side is open, the amplitude of E_s1 and E_s2 is equal, the phase is consistent, and the frequency is f₁; these values are obtained through the expression of the stator winding current I₁ and the magnetizing reactance X_m,:

$$E_{\mathrm{s}1}=E_{\mathrm{s}2}=-j{\displaystyle \frac{I_1}{2}}{X}_{\mathrm{m}}$$

(15)

$$E_{\mathrm{s}}=E_{\mathrm{s}1}+E_{\mathrm{s}2}$$

(16)

From the perspective of the stator, the forward and reverse rotating magnetic fields are closely related. The sum of E_s1 and E_s2, plus the leakage impedance voltage drop of the stator winding, should be balanced with the applied voltage in accordance with the following:

$$U=I_1{Z}_{\mathsf{\sigma }}+E_{\mathrm{s}1}+E_{\mathrm{s}2}$$

(17)

From the perspective of the rotor, the role of forward and reverse-rotating magnetic fields should be considered separately. The rotor circuit equation is transformed, and the original equation of the forward magnetic field rotor circuit is as follows:

$$S{E}_{\mathrm{s}1}={\displaystyle \frac{1}{2}}[R_2^{\prime }+jS{X}_2^{\prime }]I_{\mathrm{r}1}$$

(18)

In this formula,

$$X_2^{\prime }=X_{2\mathsf{\sigma }}^{\prime }+X_{\mathrm{m}}$$

(19)

After the transformation,

$$E_{\mathrm{s}1}={\displaystyle \frac{1}{2}}[{\displaystyle \frac{R_2^{\prime }}{S}}+j{X}_2^{\prime }]I_{\mathrm{r}1}$$

(20)

The original equation of the reverse magnetic field rotor circuit is as follows:

$$(2-S)E_{\mathrm{s}2}={\displaystyle \frac{1}{2}}[R_2^{\prime }+j(2-S)X_2^{\prime }]I_{\mathrm{r}2}$$

(21)

After the transformation,

$$E_{\mathrm{s}2}={\displaystyle \frac{1}{2}}[{\displaystyle \frac{R_2^{\prime }}{(2-S)}}+j{X}_2^{\prime }]I_{\mathrm{r}2}$$

(22)

After the equivalent transformation, the equivalent circuit of the rotor side can be converted to the stator side. Finally, the complete equivalent circuit of the bearing-integrated rotary transformer is simplified, as shown in Figure 2. When the rotor winding flows through the current, a magnetic field is generated in the air gap, which affects the two rotating magnetic fields in the air gap; thus, the two magnetic field amplitudes, E_s1 and E_s2, are not equal during the operation of the bearing-integrated rotary transformer with the load.

3. The Design of a Bearing-Integrated Rotary Transformer

Figure 3 shows the basic structure diagram of the bearing-integrated rotary transformer. From outside to inside, it has a stator, outer sheath, needle roller, inner sheath, and rotor. The needle roller supports the bearing and magnetic field as a modulation ring. Figure 4 shows the shaft section diagram of the bearing-integrated rotary transformer. The modulation ring is composed of inner and outer sheaths and magnetic rollers. The air gap magnetic field is changed by modulation in the magnetic field to realize energy exchange.

3.1. Machinery Design

According to the principle of magnetic field modulation, the rotor structure directly affects the path of the main magnetic circuit. According to the topology of the bearing-integrated rotary transformer given in Figure 3, three rotor structures with different numbers of slots are given. The number of slots is 2/4/6, respectively. Through finite element calculation, a comparison diagram of magnetic density and tooth width was created, as shown in Figure 5.

It can be seen from Figure 5 that as the number of rotor slots increases, the magnetic density and width of the tooth gradually decrease. A lower number of slots on the rotor translates to an incomplete magnetic circuit, increased magnetic density, and saturated core. However, too many slots will make the core saturated, and the tooth width of the rotor decreases, which increases the difficulty of manufacturing. Therefore, the rotor of the bearing-integrated rotary transformer in this paper selects three pairs of poles and six slots.

3.2. Stator Structure Design

The number of modulation needle rollers is the sum of the number of stator poles and the number of rotor poles. Therefore, based on the finite element method, five different coordination schemes are simulated and analyzed, and the back-electromotive force waveforms under different schemes are obtained. It is known that the rotor has three pairs of poles, and the number of stator slots and modulation needle rollers are 12/9, 14/10, 16/11, 18/12, and 20/13, respectively. The no-load back-EMF waveforms of the five different schemes are shown in Figure 6, and the back-EMF amplitude comparison is shown in

Table 1. Comparison of back-electromotive amplitude.

Scheme	12/9	14/10	16/11	18/12	20/13
back- electromotive force	13 V	6.3 V	11 V	7.6 V	8.3 V

.

Figure 6 shows the sine degree of the back-electromotive force waveform of the bearing-integrated rotary transformer for Scheme 3, Scheme 5, Scheme 2, Scheme 1, and Scheme 4. The waveform sine degree of Scheme 3 is the best, and its back-electromotive force amplitude is also the largest. Therefore, we chose the scheme with the 16/11 number of stator slots and modulation needle roller groups; the number of stator poles was 8, the number of slots was 16, and the number of needle roller groups was 11.

There are 11 groups of adjustable magnetic needles. The structure of the modulation ring is shown in Figure 7. The needle is assembled between the inner sheath and the outer sheath, and the position is fixed by the raceway of the outer sheath. The modulation ring is assembled between the stator and the rotor.

3.3. Winding Design

The bearing-integrated rotary transformer designed in this paper has eight stator pole pairs, sixteen stator slots, three rotor pole pairs, and a rotor slot number of six. The stator and rotor have single-phase single-layer windings. The stator windings and rotor windings all adopt the whole pitch distribution, and the number of slots per pole and per phase is one. Figure 8 shows the winding expansion diagram.

3.4. Analysis of Polar Arc Coefficient Selection

To study the influence that the pole arc coefficient of the flux-adjusting needle has on the amplitude of the no-load back-EMF and the amplitude of the magnetic flux density, the model of Pr = 3, Ps = 8, and P0 = 11 was studied and analyzed. The pole arc coefficient of the bearing-integrated rotary transformer can be changed by altering the number of needle rollers in each flux-adjusting needle group and the angle between adjacent needle rollers. Through finite element calculation, Figure 9 shows the influence of the pole arc coefficient on the amplitude of no-load back-EMF, and Figure 10 shows the change in harmonic amplitude for the air-gap magnetic flux density with the pole arc coefficient.

In Figure 8 and Figure 9, the no-load back electromotive clearly increases with the pole arc coefficient before 0.5 and gradually decreases after 0.5. The change in the harmonic amplitude of the air gap flux density also increases initially before decreasing. Therefore, in this paper, the polar arc coefficient is 0.5, and the corresponding number of rolling needles in each group is three. According to the above analysis, the basic parameters of the bearing-integrated rotary transformer are the pole slot matching, size, magnetic needles, winding, and so on, for the stator and rotor, as shown in

Table 2. Bearing-integrated rotary transformer parameters.

Parameter	Sign	VAL
Output power (W)	P0	60
Reactive power	Qr	10.82
Rated frequency (Hz)	fs	150
Axial length (mm)	h	80
Stator outer diameter (mm)	Φs1	120
Bore diameter (mm)	Φs2	84
Number of stator slots	Zs	16
Stator pole pairs	Ps	8
Stator winding circles	N1	35
Stator sheath thickness (mm)	ds	1
Rolling needle radius (mm)	R0	2
Modulation ring pole number	P0	11
Number of rolling needles (a)	N0	33
Rotor sheath thickness (mm)	dr	1
Rotor outer diameter (mm)	Φr1	72
Rotor inner diameter (mm)	Φr2	24
Rotor slot number	Zr	6
Rotor pole pairs	Pr	3
Rotor winding turns	N2	15

; a three-dimensional structure diagram of the bearing-integrated rotary transformer is shown in Figure 11.

4. Simulation Analysis of Bearing-Integrated Rotary Transformer

After designing the size of the bearing-integrated rotary transformer, it is necessary to analyze its performance. The bearing-integrated rotary transformer has a unique structure, and its manufacturing and assembly are more complicated than traditional transformers. This study simulates the performance of the model based on the design. Because the bearing-integrated rotary transformer needs to operate in the vehicle motor, the working environment of the transformer requires excellent heat resistance. When designing its size, it is necessary to control the thermal load of the transformer. Before the electromagnetic analysis, the bearing-integrated rotary transformer needs to be modeled. The three-dimensional finite element model of the bearing-integrated rotary transformer is shown in Figure 12.

According to the principle of the finite element method, which analyzes and solves problems, mesh generation is needed before electromagnetic field calculation. The denser the mesh generation, the longer the simulation time and the higher the simulation accuracy. In general, according to design experience, the air gap in the rotary transformer should have the densest grid. The grid subdivision of the bearing-integrated rotary transformer is shown in Figure 13.

4.1. Electromagnetic Simulation

In this study, the electromagnetic simulation of the bearing-integrated rotary transformer was designed using transient field current source simulation. When the bearing-integrated rotary transformer was unloaded, a 7 A current was added to the stator winding, and the speed was set to 2500 rpm. Through post-processing, the magnetic field distribution of the bearing-integrated rotary transformer at a certain time under a no-load condition is shown in Figure 14.

In Figure 14, under no-load conditions, the magnetic flux density amplitudes of the stator teeth and yoke of the bearing-integrated rotary transformer are 1.05 T and 1.31 T, respectively, and the magnetic flux density amplitudes of the rotor teeth and yoke are 0.8 T and 0.74 T, both less than 1.8 T. The magnetic field line shows that the stator magnetic field has eight pairs of poles. The magnetic field passes through the stator, needle roller, and rotor to form a loop, and the magnetic flux leakage is minimal, which meets the design requirements.

The no-load back-EMF induced by the rotor winding is shown in Figure 15, with an amplitude of 11 V and an adequate sine degree for the waveform.

To assess the bearing-integrated rotary transformer when operating under load, the external circuit shown in Figure 16 was built. At this time, the stator magnetic field and the rotor magnetic field were combined to synthesize the load magnetic field, as shown in Figure 17, which is the magnetic field distribution of the bearing-integrated rotary transformer at a certain time under load. The magnetic flux density amplitudes of the stator teeth and yoke of the bearing-integrated rotary transformer under load operation were 0.53 T and 0.65 T, respectively, and the magnetic flux density amplitudes of the rotor teeth and yoke were 0.38 T and 0.46 T, respectively.

A 3Ω resistive load was added to the rotor winding: the back-electromotive force of the rotor winding is shown in Figure 18, and the load current after rectification is shown in Figure 19.

4.2. Temperature Field Simulation

Figure 20 shows the curve of core loss with time when the bearing-integrated rotary transformer was loaded. The total volume loss was 13.06 W, which was calculated according to the product of the total volume and the loss per unit volume.

Figure 21 shows the curve of copper loss with time when the bearing-integrated rotary transformer was loaded. The diagram shows that the winding ohmic loss of the bearing-integrated rotary transformer is 26.12 W with load.

The temperature distribution of the bearing-integrated rotary transformer was simulated and analyzed. The overall temperature distribution is shown in Figure 22 for when the bearing-integrated rotary transformer ran continuously for one hour to reach a steady state.

From the analysis in Figure 22, when the bearing-integrated rotary transformer works at the rated speed, the temperature of the rotor winding and the rotor core of the bearing-integrated rotary transformer is higher, reaching 46.41 °C. The rotor winding achieves a high temperature because most of the heat in the rotor winding is transmitted through the rotor. The rotor core has a high temperature due to its location in the central part of the bearing-integrated rotary transformer as a whole. The heat dissipation condition is relatively poor for the stator and the sheath. The lowest temperature is located in the stator yoke because the heat generated during the operation of the bearing-integrated rotary transformer is mainly distributed to the environment through the end face of the stator core. The temperature distribution of each part of the rotary transformer is shown in Figure 23.

5. Prototype Development and Experimental Analysis of Bearing-Integrated Rotary Transformer

To verify the rationality of the structural size, design method, and electromagnetic characteristics of the bearing-integrated rotary transformer, according to the size parameters of each component given in Section 3, the processing scheme and manufacturing drawings of the prototype were assessed and delivered to the motor processing plant for manufacturing. Figure 24 shows the schematic diagram of each part of the prototype. Figure 24a,b show the stator core and rotor core, Figure 24c,d represent the outer raceway and inner raceway, respectively. Figure 24e shows the modulating loop. Figure 24f,g depict the fully assembled stator and rotor, respectively. Figure 24h shows the bearing-integrated rotary transformer.

After the prototype is manufactured, the bearing-integrated rotary transformer prototype is tested. The test first needs to build the brushless excitation system of the bearing-integrated rotary transformer. The test platform in Figure 25 mainly includes an excitation power supply, drive motor, electronic load, oscilloscope, etc.

5.1. No-Load Test

When there is no load, the drive motor is not started. At this time, the speed of the bearing-integrated rotary transformer is 0 r/min. Different excitations are added to test the no-load back-electromotive waveform of the prototype under various voltages, and the experimental waveform and data are stored by an oscilloscope. As shown in Figure 26a–d, when the current is 4 A, 5 A, 6 A, 7 A, and the frequency is 150 Hz, the rotor winding test voltage waveform is compared with the simulated voltage waveform.

It can be seen from Figure 26 that when the bearing-integrated rotary transformer is stationary, different levels of amplitude current excitation are added, and the simulation results are approximately consistent with the experimental results. To meet the demand of the load motor, a current of 7 A was selected for excitation, and the test result was 8.24 V.

5.2. Load Test

The winding resistance value of the load motor is simulated with 3Ω load, the rated speed of the load motor is 2500 r/min, the drive motor is started, and currents of 7 A and 150 Hz are introduced into the stator winding. The load current waveform is shown in Figure 27, and the load voltage waveform is shown in Figure 28.

It can be seen from Figure 27 and Figure 28 that the load current is 3.12 A, and the load voltage is 9.84 V when three loads are added to the rotor winding.

5.3. Simulation and Experimental Results

Comparing experimental data with the simulation results, when the bearing-integrated rotary transformer is stationary, the finite element analysis calculates a no-load back-EMF of 8.9 V, while the measured value is 8.24 V. The waveforms match well with a 7.4% amplitude error, meeting system requirements.

When a 3Ω resistive load is connected to the rotor winding, the measured load current is 3.12 A, and the voltage is 9.84 V, aligning with the simulation results. This confirms the validity of the proposed design.

6. Conclusions

The bearing-integrated resolver proposed in this paper functions both as motor rotor support and the energy transmission of the excitation system, providing a new brushless excitation solution for electrically excited synchronous motors. The main conclusions of this study are as follows:

Electromagnetic and thermal simulations were performed on the bearing-integrated rotary transformer. Under no-load conditions, the peak magnetic flux density reached 1.05 T at the stator teeth and 0.8 T at the rotor teeth, confirming a reasonable magnetic circuit design without saturation. During the loaded operation, the flux distribution remained uniform with minimal leakage. The maximum temperature of the rotor windings reached 46.4 °C under load, satisfying the motor’s thermal specifications.

When testing the prototype of the bearing-integrated resolver, the rotor back EMF amplitude was 8.24 V at no load; compared with the simulation, the back EMF waveform was the same, and the amplitude error was 7.4%, meeting the system requirements. The load current was 3.12 A, and the load voltage was 9.84 V when three loads were added to the rotor winding, which is approximately consistent with the simulation results. This confirms the validity of the proposed design. The successful implementation of this scheme provides a new design for the brushless electric excitation synchronous motor, and its high integration has a certain reference value for the motor system.