Switchable Filter with Four Operating Modes Implemented on a 50-Ohm Microstrip Line

Abstract

In this paper, we implement a compact switchable bandpass filter on a 50 Ω microstrip line. The proposed structure consists of an input/output stage with one end terminated at 50 Ω, a C-shaped-open loop resonator, and two L-shaped-open loop resonators. The proposed filter operates in four different modes depending on the on/off combination of the five PIN diodes. Each mode includes a dual-band pass filter (DB-BPF) designed for the 1.4 GHz and 5.1 GHz bands, another DB-BPF covering the 2.4 GHz and 4.2 GHz bands, a wideband BPF with a bandwidth ranging from 2 to 4.5 GHz, and an all-pass filter (APF) that allows all frequencies to pass through. The proposed structure is extremely compact because it is implemented on a 50 Ω line without any additional space.

1. Introduction

As wireless technology convergence accelerates, the demand for radio frequency (RF) front-end components that can support multiple communication standards within a single platform is increasing [1]. The 5G NR Sub-6 GHz band requires a wide bandwidth of up to 100 MHz for high-speed data transmission [2]. In contrast, IoT services such as NB-IoT and IEEE 802.11 operate in a narrow bandwidth of several MHz to tens of kHz to achieve very low energy consumption [3]. If these heterogeneous requirements are addressed with multiple circuits, the circuit area increases, which is disadvantageous for terminal designs where miniaturization is important.

Accordingly, switchable filters that provide various frequency responses in a single circuit are attracting attention. Among various implementation methods, PIN diode-based reconfigurable filters are particularly suitable for multi-standard transceivers that must quickly switch between wideband 5G links and ultra-narrowband IoT channels due to their fast switching speed and simple circuit configuration [4,5].

In [6], a switchable filter with four distinct modes was proposed by integrating a bandstop filter (BSF)/all-stop filter (ASF) structure with a dual-mode resonator (DMR) in a parallel configuration. In [7], three modes of bandpass filter (BPF), BSF, and DB-BPF were implemented based on coupled-lines. In [8], a switchable filter capable of implementing two BPFs and one BSF was proposed, while in [9], a switchable filter designed to implement three modes—BPF, DB-BPF, and BSF—was proposed by combining one main path and an additional path. In [10], a PIN diode was used to switch between BPF and BSF, and a varactor diode was used to control their resonant frequencies. Similarly, in [11], a structure enabling switching between two modes, BPF and BSF, was proposed, with an absorptive circuit added to enhance the matching characteristics of both modes. In [6,10], two or more structures with different filter functions are positioned in parallel, and they operate in different modes depending on which structure is activated. Since sufficient length must be guaranteed for coupling between each structure, it has the disadvantage of being large in size and complex in design parameters. In the case of [7,8,9,11], a method is used to switch the filter mode by selecting one of several lines through a single pole multi-throw switch. However, this structure is larger than the method that utilizes coupling because it arranges multiple structures in parallel. In addition, since it is a method of controlling multiple PIN diodes, the complexity of the bias circuit increases and the insertion loss increases.

To address these issues, research on implementing filter structures directly on 50 Ω microstrip lines has been attracting attention [12,13,14,15]. In [12], a sext-band BPF was proposed by adding a symmetrical short stub outside the 50 Ω line. In [13], a filter capable of switching between three modes—a broadband filter, a dual-band filter, and a tri-band filter—was implemented by integrating a stub with a PIN diode into the 50 Ω transmission line. However, both studies had the problem of increasing the overall filter size due to the long stub. In [14], a structure capable of switching between BPF and DBPF was proposed using a switchable J-inverter. However, it still had the problem of large size, with a large IL (Insertion Loss) of 2.9–4.9 dB using 13 diodes. In [15], a dual-band notch filter with tunable frequency was implemented by applying an embedded split ring resonator (ESRR) structure loaded with a varactor inside a 50 Ω line. In [16], a simple and compact quadruple-band pass filter was implemented by inserting a C-shaped open-loop resonator inside a 50 Ω line, but the spacing between resonant frequencies is fixed, so it has limitations in operating in various environments.

In this study, the filter was designed by inserting a resonator inside a 50-ohm line, and its size was minimized to 0.4 × 0.05 λ_g². The proposed filter operates in the 1.35–1.45 GHz, 2.30–2.50 GHz, 4.00–4.30 GHz, 4.85–5.30 GHz, and 2.0–4.5 GHz bands, and all band-pass modes can cover 0–7 GHz. These frequency ranges include the 1.4 GHz extended band of NB-IoT, the 2.4 GHz and 5 GHz bands of IEEE 802.11, and the n77 (3.3–4.2 GHz), n78 (3.3–3.8 GHz), and n79 (4.4–5.0 GHz) bands of 5G NR. The proposed architecture demonstrates that it can operate in multiple band frequency ranges within a single circuit and can support operations in 5G and IoT of various standards.

2. Configuration

Figure 1 shows the basic shape of the filter implemented on a simple 50 Ω line. A substrate with a dielectric constant of 2.2 and a thickness of 62 mil is used. The line width at both ends of the basic filter structure is 4.8 mm, which corresponds to a characteristic impedance of 50 Ω. The proposed basic filter consists of a pair of stubs on the top and bottom, two L-shaped resonators, and one C-shaped resonator. The stub has a line width of w and its length of s. The line width of the C-shaped resonator is set to L_C, and the line width of the L-shaped resonator is also set to 0.4 mm. The spacing between the stub and the L-shaped resonator, the spacing between the stub and the C-shaped resonator, and the spacing between the 50-ohm line and the L-shaped or C-shaped resonator are set to d.

The basic structure shown in Figure 1 features a total of five gaps, each designated as g, where PIN diodes are applied to enable its function as a switchable filter. Considering that PIN diodes will be applied, all gaps are set to 0.1 mm. Depending on the various on/off combinations of the five PIN diodes, the proposed structure operates as a wideband BPF, a narrowband dual-band BPF, and an all-pass filter.

3. Wideband BPF and Dual-Band BPF

3.1. Operating Principle

Figure 2 shows four different filter structures and their simulation results according to the applied resonator shape inside the 50-ohm line. Based on these characteristics, a switchable filter will be considered in Section 4. The filter parameters are set as follows: L = 30 mm, s = 14.9 mm, L_C = 31.8 mm, w = 0.4 mm, g = 0.1 mm, and d = 0.1 mm.

The structure presented in Figure 1 is Type 3, and the structure with only two L-shaped resonators on the top in Figure 1 is Type 1, and the structure with only one C-shaped resonator on the bottom is Type 2. From the simulation results, it can be seen that Type 1 does not allow RF signals to pass through the entire band, and Type 2 and Type 3 operate identically. From these results, it can be inferred that the two L-shaped resonators on the top are meaningless dummy structures, and the dual-band filter characteristics are implemented by one C-shaped resonator on the bottom. Type 4 is the case where two identical C-shaped resonators are applied on the top and bottom. At this time, the proposed structure exhibits wideband filter characteristics.

3.2. Wideband BPF

Type 4 structure has the same structure at the top and bottom, so it is sufficient to consider only the structure in Figure 3a. As illustrated in Figure 3b, the wideband BPF is constructed using two pairs of coupled transmission lines, each with an electrical length of θ and characteristic impedances Z_0o and Z_0e for the odd and even modes, respectively. Owing to the filter’s symmetrical structure, its behavior can be effectively analyzed through odd- and even-mode techniques, with the corresponding equivalent circuits depicted in Figure 3c,d.

The even-mode input impedance of Figure 3c can be expressed by

$$Z_{ine}=-{\displaystyle \frac{j}{2}}\left(Z_{0e}+Z_{0o}\right)\cot \theta$$

(1)

and the input reflection is

$${\mathsf{\Gamma }}_e={\displaystyle \frac{Z_{ine}-Z_o}{Z_{ine}+Z_o}}$$

(2)

Likewise, the odd-mode input impedance can be obtained

$$Z_{ino}=j {\displaystyle \frac{{\left(Z_{0e}-Z_{0o}\right)}^2-{\left(Z_{0e}+Z_{0o}\right)}^2{\cos \theta }^2}{\left(Z_{0e}+Z_{0o}\right)\sin 2\theta }}$$

(3)

Similarly, the corresponding input reflection is

$${\mathsf{\Gamma }}_o={\displaystyle \frac{Z_{ino}-Z_o}{Z_{ino}+Z_o}}$$

(4)

Therefore, transmission and reflection coefficients of the proposed filter can be obtained as

$$S_{21}={\displaystyle \frac{{\mathsf{\Gamma }}_e-{\mathsf{\Gamma }}_o}{2}}$$

(5)

$$S_{11}={\displaystyle \frac{{\mathsf{\Gamma }}_e+{\mathsf{\Gamma }}_o}{2}}$$

(6)

Figure 4a shows the simulation results of Type 4 (wideband BPF mode) according to the change in stub length, s. The parameters for the simulation are the same as those mentioned above. From the simulation results, it can be confirmed that the stub length, s, affects f₂ and the transmission zero (TZ) frequency located on the right side of the passband in the wideband BPF mode. At this time, f₁ hardly changed. As the stub length, s, increases, the right skirt characteristic of the filter improves. In Figure 1, the longest s that can be implemented is 14.9 mm. Considering this fact, the stub length s is selected as 14.9 mm.

Figure 4b shows the electric field distribution at the TZ frequency of 6.57 GHz when s is 14.9 mm. The electric field is stronger as it approaches red and weaker as it approaches blue. In the simulation, the RF signal is set to the left for input and the right for output. Type 4 is a symmetrical structure based on the left and right, and the left and right sides operate as bandstop resonators of one wavelength each. As a result, the S₂₁ of the filter exhibits good characteristics of about −50 dB.

Figure 5 shows the simulation results of the Type 4 structure according to the total filter length L. From the simulation results, it can be seen that L has a significant effect on the first resonant frequency f₁, and f₁ decreases as L increases.

Figure 6 presents the simulation results of Type 4 by changing the length of one of the two C-shaped resonators. In Figure 6, we will explain why the proposed structure operates as a wideband filter for Type 4 and a dual-band filter for Type 3. Here, when L_C is 31.8 mm, it refers to the Type 4 structure. At this time, the 3 dB operating bandwidth is 1.98–5.18 GHz. The simulation results show that L_C gradually decreases and when it reaches 28 mm, two TZs start to appear in the passband of the wideband BPF.

As L_C decreases, the frequency of the two TZs increases and becomes more distinct. When the upper C-shaped resonator disappears completely (L_C = 0 mm), it becomes the same as Type 2 in Figure 2 and has dual-band filter characteristics. It has already been mentioned above that Type 2 has the same characteristics as Type 3. In this study, the first TZ of the filter is generated by a stub with a length of s, and when a C-shaped resonator is added, the stub and the C-shaped resonator operate as a coupled line. At this time, if the length of the C-shaped resonator is 28 mm or more, the TZ does not appear.

3.3. Dual-Band BPF

If the structure includes two L-shaped resonators and one C-shaped resonator, as in Figure 7a, which is the same as Type 3 in Figure 2a, it operates as a dual-band BPF. The dual-band filter has two TZs, one located between the two passbands (f₁ and f₂) and one located in the upper frequency band of the upper passband (f₂). The corresponding frequencies are referred to as f_TZ1 and f_TZ2, respectively. Figure 7b shows the simulation results of Type 3 as the stub length, s, changes. As s increases, f_TZ1 decreases. At this time, f_TZ2, which exists on the right side of f₂, shows a tendency to decrease slightly.

Figure 8a,b show the E-field distribution formed at the transmission zero frequency, respectively. As can be seen from the field distribution, the stubs are operating as λ/4 resonators and λ resonators, respectively. The field distribution shown in Figure 7b is identical to that in Figure 4b. When operating in dual-band filter mode, the second TZ is identical to the right TZ in the wideband filter mode.

Figure 9 shows the simulated S₂₁ response of the Type 3 structure as the length L_C changes. According to the simulation results, as the length of L_C decreases, f₂ shifts significantly to higher frequencies, and f₁ also increases, but the change is relatively small. This indicates that the change in the length L_C of the C-shaped resonator is more sensitive to the higher resonant frequency f₂.

Figure 10 shows the electric field distribution at two resonant frequencies f₁ and f₂ of Type 3 with dual-band BPF characteristics. The parameter settings for the simulation are as follows: L = 30 mm, s = 14.9 mm, L_C = 32.7 mm, g = 0.1 mm. As shown in the figure, in the case of Type 3, the RF signal is well transmitted along the lower path at both resonant frequencies. On the other hand, a strong electric field is formed between the gaps in the upper path, and the electric field cannot pass from left to right. At f₁, one null exists in the center, and at f₂, two nulls exist, operating at half-wavelength and one-wavelength, respectively.

Figure 11 shows the S₂₁ simulation results of the Type 3 structure according to the change in length L. The length, L, is a parameter that affects both the main resonant frequencies f₁ and f₂. According to the simulation results, as the length of L increases, the two resonant frequencies f₁ and f₂ shift to lower frequencies, respectively. This occurs because an increase in length, L, extends the structure’s effective electrical length, thereby lowering the resonant frequency.

4. Switchable Filter Design with Bais Circuit

Figure 12a shows a switchable filter with five diodes and a bias circuit to operate them. The configuration of the switchable filter is the same as Figure 1 except for the bias circuit and diodes. The two D₂s located on the left and right gaps and the two D₃s located on the upper and lower outer gaps are each configured as a pair. The bias circuit consists of an inductor (green), a capacitor (blue), and a resistor (black). As is well known, the thinner and shorter the bias line, the better. Accordingly, the line width was set to 0.1 mm and the length to 5.6 mm. To implement the bias circuit in a limited space, the capacitor is located on the back side where the GND is located. L_i1 = 3, L_i2 = 1 mm are set, and other parameters are maintained as optimized values obtained based on the parameter study performed in Section III. Figure 12b shows the equivalent circuit of the diode used. It acts as a resistor in the On state and as a capacitor in the Off state. R_on is 5.2 ohm, C_off is 0.025 pF, and L_S is about 0.5 nH.

When simulating the switchable filter including the bias circuit, the equivalent values of the diode are all considered. The bias voltages used to control the On/Off state of each diode are shown in

Table 1. Applied bias voltage conditions and operating modes.

Condition [V]					Diode State			Operating Mode
V1	V2	V3	V4	V5	D1	D2	D3
4	1	0	0	1	On	Off	Off	Wideband
1	3	6	0	1	Off	On	Off	Dual-band
2	4	0	0	1	Off	Off	Off	Dual-band
-	-	-	1	0	-	-	On	All pass

. In the case of D₃, the bias voltage is applied by connecting a bias tee to the ports at both ends of the line. The voltages applied to the bias tees are indicated as V₄ and V₅ in Figure 12a, respectively. Figure 12c shows the fabricated switchable filter.

Figure 13 shows the simulated filter characteristics when the values of R and L used in the bias circuit are changed. The values of R and L used in the bias circuit have a large effect when the proposed structure operates as a dual-band filter but have a relatively small effect when it operates as a wideband filter. For this reason, only simulation results applying the bias circuit to the dual-band filter are presented in this paper. As the inductor value increases, the insertion loss in the 1.5 GHz band, which is f₁, tends to decrease. At this time, the insertion loss in the 5.5 GHz band, which is f₂, showed almost no change. Since the inductor has a large impedance value as the frequency increases, the impedance is relatively large in the f₂ band, which is a relatively high frequency. Accordingly, the RF signal flowing through the bias circuit can be appropriately blocked. On the other hand, the impedance is relatively small in the f₁ band, so some RF signals flow into the bias circuit. This is because the insertion loss in f₁ improves as the inductor value increases.

However, when the inductor value is 90 nH, S₂₁ ripple is formed in the f₂ band. Among the three inductors, the insertion loss could be reduced by lowering the inductor value of the V₂ section to 68 nH. When no resistor is used in the bias circuit, the insertion loss is the smallest in the 1.5 GHz band, but resonance occurred at 0.6 GHz and strong ripple occurred in the 5.5 GHz band. When a 0.3 Kohm resistor is used, the insertion loss in the f₁ band decreases, but the insertion loss in the f₂ band increases.

Figure 14 shows the electric field distribution when the inductor values used in the bias circuit are 29 nH and 90 nH/68 nH, respectively. At this time, the simulation is performed at f₁. In the case of 29 nH, it can be confirmed from the simulation results that the electric field is also formed in the bias circuit. On the other hand, when a 68 nH inductor was used in the V2 section and 90 nH inductors were applied to the other two bias lines, the amount of RF signal leaking into the bias circuit was significantly reduced, as confirmed by the simulated electric field distribution. Based on these facts, it can be confirmed that the small inductor value is not sufficient to operate as an RF chock to prevent the electric field from passing through to the bias circuit.

Figure 15 shows the simulation results of the effect of the bias circuit applied position L_i1 on the filter performance. As the bias applied position changes, there is little effect in other modes, but the effect is very strong in the BSF mode where the D₂ diode is turned on. From the simulation results, as L_i1 increases, the insertion loss of S₂₁ tends to decrease in both resonant frequency bands. However, when L_i1 is 5 mm, there is a problem of S₂₁ ripple occurring in the 5.5 GHz band. Considering these facts, L_i1 is selected as 3 mm.

The switchable filter operates as two types of dual-band BPF, wideband BPF, and all-pass filter depending on various bias combinations. The simulation and measurement results for these four cases are presented in Figure 16. It is confirmed that some measured TZ occurred at a slightly lower frequency than the simulation. These inconsistencies may be attributed to slight inaccuracies in the lengths of the L- and C-shaped resonators caused by fabrication tolerances during the manufacturing process. In addition, the resonators are connected through PIN diodes, whose parasitic elements can alter the effective electrical lengths of the resonators. Considering these facts, the simulation and measurement results can be seen to be in good agreement overall. When D₁ is off, D₂ is on, and D₃ is off, the switchable filter operates as a DB-BPF, and the results are presented in Figure 16a. The measured 3 dB fractional bandwidths centered at f₁ = 1.4 and f₂ = 5.1 GHz are 23.49 and 14.42%, respectively. The measured insertion losses at this time are −1.33 and −1.3 dB, respectively. On the other hand, when D₁ is off, D₂ is off, and D₃ is off, the switchable filter operates as a DB-BPF, and the operating bandwidths are f₁ = 2.4 and f₂ = 4.2 GHz, respectively, and the measured 3 dB fractional operating bandwidths are 23.33 and 13.27%. The insertion losses are −0.88 and −2.04 dB, respectively. Figure 16c shows the result when the switchable filter operates as a wideband BPF. At this time, the bias conditions are D₁ on, D₂ off, and D₃ off. The measured 3 dB fractional bandwidth is 73.36% (2.05–4.41 GHz) and the insertion loss is −0.69 dB. Figure 16d shows the result when operating as an APF. At this time, the bias conditions are D₁, D₂ don’t care, and D₃ on. The measurement results show that the APF exhibits an insertion loss of −2.34 dB in the entire operating frequency range of 0–7 GHz, confirming that signal transmission is efficient.

Table 2. Comparison with previous switchable filters.

Ref.	Function	Operating Frequency Range [GHz]	BW [%]	IL [dB]	Num. of Diodes	Structure	Circuit Size [λg × λg]
[6]	BSF/ASF/ BPF/DBSF	1.5–2.5/-/1.83–2/1.44–1.78 & 2.12–2.6	50.2/-/9.7/22.1 & 20	1.99	4	Parallel filter branches	0.21 × 0.22
[7]	BPF/BSF/ DBSF	1.41–2.43/1.63–2.21/1.55–1.61 & 2.2–2.26	53.1/30.2/3.8 & 2.7	0.9/2/2.2	6	SP3T-controlled filter branches	0.44 × 0.29
[8]	WBPF / BPF / BSF	1.51–4.38/2.55–3.29/1.41–4.59	95.7/25.3/106	0.94/0.8	6	SP3T-controlled filter branches	0.5 × 0.25
[9]	BPF/DBPF/BSF	1.19–2.93/1.25–1.87 & 2.25–2.89/ 1.73–2.33	84.5/39.7 & 24.9/29.6	0.9/1.13 & 1.5/2	6	SP3T-controlled filter branches	0.61 × 0.27
[10]	DBPF/ DBSF	1.7–2.2 & 2.2–2.7/1.7–2.3 & 2.3–2.9	25.6 & 20.4/30 & 23.1	4.6/5.6	7	Parallel filter branches	0.75 × 0.4
[11]	BPF/BSF	1.47–2.13/1.67–1.9	36.75/13	1.73/0.8	7	SP2T-controlled filter branches	0.74 × 0.6
[13]	BBPF/ DBPF/ TBPF	1.23–1.78/1.61–1.99 & 2.76–3.24/0.36–0.44 & 1.29–1.71 & 2.41–2.89	36.7/20.7 & 16.1 / 20 & 27.4 & 18.1	0.5/0.4/0.6	7	Switchable stub loaded line	0.75 × 0.4
[14]	BPF/DBPF	0.8–1/0.65–0.94 & 0.87–1.13	36.07/26.42 & 22.92	2.9–4.9	13	Switchable J-inverter loaded line	0.25 × 0.25
This work	DBPF/ DBPF/ WBPF/APF	–&–/–&–/2.05–4.41	23.5 & 14.42/23.33 & 13.27/73.36/-	1.33 & 1.3/0.88 & 2.04/0.69/2.34	5	50 Ω line embedded filter	0.4 × 0.05

presents the performance comparison between previous switchable filters and the filter proposed in this work.

5. Conclusions

In this paper, a compact switchable filter supporting four modes is designed and implemented. The proposed filter provides DB-BPF centered at 1.4 GHz/5.1 GHz and 2.4 GHz/4.2 GHz, a wideband BPF from 2.05 to 4.41 GHz, and an APF operating from 0 to 7 GHz. The bias circuit is integrated into the 50 Ω line to improve space efficiency, and the simulation and measurement results are generally in good agreement. In particular, the proposed filter satisfies the miniaturization and multifunctionality required for next-generation communication systems such as 5G and IoT and demonstrates excellent performance.

6. Patents

A patent application based on the work presented in this manuscript is currently under preparation but has not yet been submitted.