# Transimpedance Amplifier for Noise Measurements in Low-Resistance IR Photodetectors

^{1}

^{2}

^{*}

## Abstract

**:**

## 1. Introduction

_{B}; or measurements of the current noise through the DUT biased with a constant voltage V

_{A}[14]. The measurement of the Power Spectral Density (PSD) of the voltage noise S

_{VD}or the current noise S

_{ID}provides the very same information, as long as the current–voltage characteristic I

_{B}(V

_{B}) of the DUT and its small signal impedance Z

_{D}at the selected bias point and vs. frequency are known, since [15,16]:

^{2}/Hz and A

^{2}/Hz. However, as is common practice in the field of noise measurements and instrumentation, when referring to specific values in the text, we will often express these quantities in terms of their square roots (i.e., V/√Hz and A/√Hz).

_{ID}from S

_{VD}, however, requires detailed knowledge of the device impedance, which therefore needs to be accurately measured under exactly the same environmental and bias conditions as those under which the noise voltage spectrum was obtained. This procedure can become extremely time consuming and prone to error, since two different measurement steps and setups (for noise measurement and for impedance measurements) are involved.

## 2. Proposed Approach

_{D}) is biased at a constant voltage V

_{B}because of the virtual short circuit between the inputs of the operational amplifier OA

_{1}. The transimpedance gain between the DUT current source i

_{D}and the output voltage of OA

_{1}is set by the resistor R

_{R}. In choosing the value of R

_{R}, the linearity range of OA

_{1}must be taken into account, since the DC current flowing through the DUT also flows through the feedback resistance, causing a large DC component. Higher values of resistance R

_{R}will result in higher gain, but also in a reduced maximum bias current for the DUT. It will be assumed, however, that the gain of the first stage is sufficient to ensure a negligible influence of the noise contributions from the following stages to the overall noise of the system. The high-pass filter C

_{1}R

_{1}rejects the DC component at the output of OA

_{1}so that only the noise signal can be further amplified by the voltage amplifier. The value of A

_{v}ensures that the signal level at its output is compatible with the input range of the spectrum analyzer used for spectral estimation. The relevant noise sources that contribute to the background noise (BN) are also shown in Figure 1. For the amplifier to be effective, we need the noise contribution of these sources to be negligible with respect to the noise generated by the DUT. In the measurement bandwidth (determined by the corner frequency of the high-pass filter C

_{1}R

_{1}and the bandwidth limit of the transresistance and the voltage amplifier stages), the Power Spectral Density (PSD) S

_{OUT}recorded by the spectrum analyzer is:

_{in}, S

_{en}, and S

_{eB}are the PSDs of the noise sources i

_{n}, e

_{n}and e

_{B}, k is the Boltzmann constant, and T is the absolute temperature. To obtain the expression of the background noise S

_{iBN}, as shown in Equation (3), we assumed all noise sources to be uncorrelated.

_{en}and S

_{eB}can be tolerated, and this means that Junction Field-Effect Transistor (JFET) or Metal-Oxide-Semiconductor Field Effect Transistor (MOSFET) input operational amplifiers can be used to minimize the contribution from the current noise S

_{in}. Indeed, in many cases of interest, the main contribution to the background noise comes from the feedback resistance R

_{R}, which must be chosen to be as large as possible, in a manner compatible with the limitations mentioned above as well as the desired bandwidth of the system [20,21]. However, for DUT impedances on the order of 1 kΩ or less, the contribution from S

_{en}and S

_{eB}can become relevant and cannot be neglected any longer. To simplify this discussion, we limited our analyses to the low frequency range, in which the flicker noise component can be more easily detected. Moreover, at low frequencies, we assume the DUT impedance to behave as a resistance, that is Z

_{D}≈ R

_{D}. For this reason, the DUT model does not contain any reactive component.

_{D}and the background noise means minimizing the quantity Q

_{n}:

_{D}/R

_{R}<< 1 is a necessary condition to obtain S

_{iBN}/S

_{D}<< 1. If this condition is satisfied, we also have R

_{D}‖R

_{R}≈ R

_{D}and Equation (4) can be rewritten as:

_{eB}can be made negligible [22]. From Equation (5), if S

_{eB}≈ 0, the lowest value of Q

_{n}, for given values of S

_{in}and S

_{en}, is obtained when R

_{D}= R

_{Qmin}

_{Qmin}is frequency dependent, and that the value of R

_{D}is set by the DUT and cannot be easily modified.

_{D}= 1 kΩ with a thermal voltage noise of 4kTR

_{D}≈ 16.6 × 10

^{−18}V

^{2}/Hz $(\approx 4\mathrm{n}\mathrm{V}$/√Hz) at room temperature. Let us first explore the possibility of using a monolithic low-noise operational amplifier with BJT (Bipolar Junction Transistor) or FET input stage technology. With BJT input stages, lower levels of input voltage noise can be achieved. On the other hand, the current noise obtained when using BJTs is typically six orders of magnitude higher than that obtained using MOSFET input amplifiers. As an example, the voltage noise of the MOSFET input TLC2201 operational amplifier at 1 Hz is 3.6 × 10

^{−15}V

^{2}/Hz (60 nV/√Hz), with a specified current noise of less than 10

^{−30}A

^{2}/Hz (1 fA/√Hz). With our assumed reference R

_{D}, the contribution of S

_{in}to Q

_{n}is completely negligible (less than 10

^{−8}), but due to the value of S

_{en}, Q

_{n}is about 225. For the low-noise BJT input OP27, the voltage noise is 100 times lower (36 × 10

^{−18}V

^{2}/Hz) than that of the TLC2201, resulting in a contribution to Q

_{n}of about 2.25. On the other hand, the effect of the current noise is not negligible as before: at the same frequency (1 Hz), its value is about 22.4 × 10

^{−24}A

^{2}/Hz (≈ 5 pA/√Hz), and this results in a further contribution of about 1.6 to Q

_{n}, for a total value of Q

_{n}close to 4. Note, moreover, that the reference value for our chosen resistance is close to the value that minimizes Q

_{n}at 1 Hz for the OP27 (R

_{Qmin}= 1270 Ω). This means that for impedance values significantly below or above 1 kΩ, Q

_{n}becomes significantly greater than 4.

_{n}that is significantly below 1, we must therefore resort to a custom design for the operational amplifier OA1. In particular, we can take advantage of the noise characteristics of discrete component devices (BJTs or JFETs), allowing us to obtain significantly lower levels of equivalent voltage noise. The reduction in voltage noise, however, is accompanied by an increase in current noise. This means that unless we are dealing with very low impedances (well below 100 Ω), there is no advantage in using discrete BJTs as front-end devices. On the other hand, discrete low-noise JFET devices can make it possible to reach sufficiently low equivalent input voltage noise, with the contribution of the current noise to Q

_{n}remaining negligible.

## 3. Materials and Methods

_{1}, together with the resistance R

_{SS}, behaves as a current source for biasing the JFET pair (J

_{1}and J

_{2}). The collector current I

_{C}

_{1}of Q

_{1}, with V

_{DD}= V

_{SS}= 12 V, is approximately:

_{BEON}is the voltage drop between the base and the emitter of Q

_{1}in the active region (in the range from about 0.6 to 0.7 V). At rest and under ideal conditions, the gate voltages V

_{G}

_{1}and V

_{G}

_{2}of J

_{1}and J

_{2}are at zero potential, so that the two JFETs operate with the same current: I

_{D}

_{1}= I

_{D}

_{2}= I

_{C}

_{1}/2 = 3.5 mA. The voltage drop across the drain resistances R

_{D}

_{1}and R

_{D}

_{2}is, therefore, 7 V, so that the drain-to-gate voltages V

_{DG}

_{1}and V

_{DG}

_{2}are maintained at about 5 V, ensuring operation in the active region (the typical pinch-off voltage for the IF3602 is −350 mV). The values of the bias currents for J

_{1}and J

_{2}are the result of a compromise between the need for low noise (equivalent input noise decreases when increasing bias) and the need to limit the power dissipated by the active devices in order to limit the convective motion of the air close to the JFET, which can induce large fluctuations at low frequencies.

_{m}of each JFET can be expected to be about 70 mA/V [14], and, because R

_{D}

_{1}= R

_{D}

_{2}= R

_{D}= 2 kΩ, the differential voltage gain A

_{VDJ}of the JFET differential stage at low frequencies can be estimated to be:

_{1}and G

_{2}represent the non-inverting and inverting inputs, respectively. From Equation (8) and the fact that the DC gain of the OPA227 is about 160 dB, the DC gain of the SOA is above 200 dB, and therefore the internal compensation of the OPA227 is not sufficient to ensure stability for the entire amplifier. To address this issue, we resort to the compensation network made of R

_{C}

_{1}, R

_{C}

_{2}, C

_{C}

_{1}, and C

_{C}

_{2}in Figure 2. The OPA227 introduces a pole at about 3 Hz as part of its frequency response. The compensation network introduces two poles and two zeroes, which reduce the open-loop gain to zero dB before the high-frequency poles of the OPA227 are able to introduce a further phase shift that would cause instability. Because of the 90° phase shift introduced by the dominant pole of the OPA227, the compensation network is designed in such a way that its phase contribution is less than 45° at any frequency. This can be achieved if the frequency of the zero is no larger than 10 times that of the pole. With this constraint, the gain amplitude reduction that is obtained with a single zero-pole compensation network is insufficient to reach the desired goal. For this reason, in our design, we introduce two RC networks between the drain of the JFETs. With the values for R

_{C}

_{1}C

_{C}

_{1}and R

_{C}

_{2}C

_{C}

_{2}listed in Table 1, the pole and zero frequencies are:

_{R}set by the feedback resistance R

_{R}, that is:

_{O}

_{1}can be used to estimate the DC current through the DUT. It is for this reason that this voltage is carried by a buffer (OA

_{3}) to one of the outputs of the system (V

_{ODC}).

_{O}

_{1}is typically too low to be effectively detected using a spectrum analyzer, and therefore, a second stage is used to obtain high voltage gain (O

_{A}

_{2}) after rejecting the DC component using an AC coupling filter (C

_{A}

_{2}R

_{A}

_{2}).

_{B}across the DUT are often well below 1 V, and this means that the input offset of the SOA must be maintained low both for the voltage across the DUT to essentially coincide with the external bias voltage V

_{B}, and for the output V

_{ODC}to provide the correct value for the DC current through the DUT. With a discrete JFET input stage for the SOA, the offset can be relevant. This offset is essentially due to the mismatch between the two JFETs in the IF3602 device. Offsets as large as ±50 mV can easily be experienced in the case of the IF3602 [14], and these values are quite relevant for bias voltages on the order of a few hundred mV. Therefore, the schematic diagram in Figure 2 includes a system for adding a DC voltage at the non-inverting input of the SOA, which is obtained by exploiting a trimmer (R

_{T}) together with a voltage divider (R

_{O}, R

_{A}

_{1}) and a capacitor C

_{A}

_{1}in order to filter out, as much as possible, the thermal noise generated by the offset correction circuit itself.

_{C}

_{1}, R

_{C}

_{2}, C

_{C}

_{1}and C

_{C}

_{2}.

_{D}) and background noise will be estimated with reference to the input of the last voltage amplifier in the noise measurement chain (v

_{OB}, corresponding to the non-inverting input of OA

_{2}in Figure 2).

_{A}

_{2}; (c) the equivalent input current noise source at the input of OA

_{3}is omitted, since it is shorted by the very low impedance at the output of OA

_{2}. Because of the high open-loop gain of the SOA and the fact that we are mainly interested in the noise at low frequencies, we can perform noise estimation under the assumption of a virtual short circuit between the inverting G

_{2}(−) and non-inverting G

_{1}(+) inputs of the SOA. We will also assume that we are working above the cut-off frequency of the AC coupling filter, with the minimum frequency of interest being 1 Hz. Note, however, that the capacitors C

_{A}

_{1}and C

_{A}

_{2}are not replaced with short circuits, because, due to the large amount of thermal noise introduced by the resistances R

_{OE}and R

_{A}

_{2}, their finite impedance may result in a non-negligible contribution to the BN of the system even above the cut-in frequency [23].

_{OB}by adding the noise contribution from each single source.

_{OB}

_{_iD}to the PSD of the noise at v

_{OB}due to the DUT noise source i

_{D}with PSD S

_{iD}. We have:

_{J}

_{1}of e

_{j}

_{1}and S

_{j}

_{2}of e

_{j}

_{2}to be the same, and equal to S

_{J}, their total contribution S

_{OB_J}is given by:

_{OE}is essentially reduced to R

_{O}‖R

_{A}

_{1}, which, at the minimum frequency of interest (1 Hz), is much greater than the impedance X

_{CA}

_{1}of the capacitor C

_{A}

_{1}. Therefore, the contribution S

_{OB}

_{_OE}due to the thermal noise i

_{OE}of the resistor R

_{OE}is reduced to:

_{D}

_{1}and R

_{D}

_{2}and the equivalent input voltage (e

_{O}

_{1}) and current noise sources (i

_{1A}and i

_{1B}), assuming a virtual short circuit at the inputs of the SOA, there is no contribution to the output noise. This result, however, is only an approximation, and depends on the magnitude of the loop gain and the magnitude of the PSD associated with the noise sources. Proper calculations show that these contributions can be neglected at low frequencies as long as the loop gain of the SOA in the shunt–shunt configuration in Figure 2 is high and g

_{m}R

_{D}>> 1.

_{OB_RR}of the thermal noise generated by the feedback resistance R

_{R}, it is given by:

_{A}

_{2}C

_{A}

_{2}, the reactance of the capacitor C

_{A}

_{2}is much lower than the resistance R

_{A}

_{2}, and therefore, for the contributions S

_{OB}

_{_A2}of the resistance R

_{A}

_{2}and S

_{OB}

_{_OI2}of the noise source i

_{I}

_{2}, we have:

_{II}

_{2}is the PSD of the current noise source i

_{I}

_{2}.

_{OB_EI}

_{2}due to the equivalent noise source e

_{I}

_{2}, that is:

_{eI}

_{2}is the PSD of the current noise source e

_{I}

_{2}.

_{D}, and the condition R

_{D}/R

_{R}<< 1 is satisfied.

_{OB}of the overall noise at the output v

_{OB}in the form:

_{n}as before. We have:

_{n}is expressed as the sum of two contributions to stress the fact that the contribution Q

_{n}

_{2}can be made as small as desired by increasing the values of the coupling capacitors C

_{A}

_{1}and C

_{A}

_{2}, at the cost, however, of increasing the time constants τ

_{A}

_{1}and τ

_{A}

_{2}; the fact that these time constants increase means that the settling time of the circuit increases as well, and this, besides resulting in a waste of time when connecting a new DUT or setting a different bias voltage, can make offset correction a much more challenging task. When looking at the relative weight of the terms contributing to Q

_{n}

_{2}, we can start by evaluating the term that contains the PSD S

_{II}

_{2}. The ADA4625 is a JFET input operational amplifier. The PSD for the equivalent input current noise source reported in the datasheet is 4.5 fA/√Hz, and this means that the contribution of the fraction containing S

_{II}

_{2}in Equation (18) is much lower than 1 for values of R

_{A}

_{2}on the order of a MΩ or more (R

_{A}

_{2}= 1 MΩ in our prototype). This, together with the fact that the two time constants τ

_{A}

_{1}and τ

_{A}

_{2}have similar values, and the fact that:

_{n}

_{2}is essentially set by the first term in Equation (18), that is, by the thermal noise generated by the equivalent resistance R

_{OE}that is not completely filtered out by the capacitance C

_{A}

_{1}. At the minimum frequency of interest (1 Hz), assuming a typical value for R

_{D}of 1 kΩ, we have Q

_{n}

_{2}≈ 0.06. Note that, because of the proportionality to the inverse of the frequency squared, Q

_{n}

_{2}rapidly decreases with increasing frequency.

_{n}

_{1}, too, the highest value is obtained at the lowest frequency of interest because of the flicker noise component introduced by the JFETs and by the operational amplifier OA

_{2}. At the minimum frequency of interest (1 Hz), assuming a typical value of R

_{D}of 1 kΩ and a worst-case scenario in which R

_{R}is limited to 10 kΩ (R

_{R}/R

_{D}= 10), with S

_{J}= 1 × 10

^{−18}V

^{2}/Hz [14] and S

_{eI}

_{2}= 1 × 10

^{−16}V

^{2}/Hz, we obtain Q

_{n}

_{2}≈ 0.28. If the bias conditions are such that a feedback resistance of R

_{R}= 100 kΩ can be used, Q

_{n}

_{2}is reduced to about 0.12, a value essentially set by the noise contribution coming from the JFETs. Overall, therefore, under the same conditions explored in the introduction (DUT resistance on the order of 1 kΩ), a value of Q

_{n}is obtained that is a small fraction of 1 (from 0.18 to 0.24, depending on the value of the feedback resistance). This is to be regarded as an excellent result in consideration of the fact that, when investigating the flicker noise, the thermal noise generated by the DUT can be regarded as part of the background noise of the system (i.e., the flicker noise must be much larger than the thermal noise for a reliable characterization). Therefore, obtaining a value of Q

_{n}that is a small fraction of 1 means that we are operating quite close to the ideal conditions under which no excess noise will be introduced by the amplifier.

_{n}obtained above is relative to the minimum frequency of interest (1 Hz). As the frequency increases, the background noise decreases because of the reduction of both the impedance of the coupling capacitances and the flicker noise contribution introduced by the active devices.

_{D}decreases significantly, Q

_{n}increases and the background noise of the system becomes relevant with respect to the thermal noise introduced by the DUT.

## 4. Results

_{R}, i.e., 10 kΩ and 100 kΩ, as shown in Table 1. The test measurements were initially performed using known resistances as DUTs. In particular, we tested the system with both a 100 Ω resistor and a 1 kΩ resistor as the DUT. The 1 kΩ resistor was taken as representative of the typical impedance we expect with actual devices in noise measurements. The test using a 100 Ω resistor as a DUT was performed in order to more clearly evidence the noise contribution (background noise) introduced by the amplifier, since, as was shown in the previous section, its relative weight increases with decreasing DUT impedance.

_{R}= 10 kΩ, the background noise due to the amplifier had a noticeable effect on the measurement results, although it can be observed that even in this configuration, it should be possible to perform sensible flicker noise measurements within the frequency range in which the flicker noise is much greater than the thermal noise of the device. It can also be noticed that, as can be deduced from Equation (18), the background noise in this case is mostly due to the first term in Q

_{n}

_{1}, since R

_{D}/R

_{R}= 10

^{−2}. In other words, the background noise is set by the input JFETs, and increasing the value of the feedback resistance has a negligible effect on the background noise.

_{R}= 10 kΩ (blue curve) and R

_{R}= 100 kΩ (red curve) indicates, as should be expected, that employing a higher feedback resistance is beneficial as far as the BN is concerned. On the other hand, no significant difference can be expected when even a moderate level of flicker noise is present, and therefore, in general terms, and unless the DUT impedance is considerably below 1 kΩ, there is no significant advantage to employing feedback resistances above 10 kΩ, as this will result in a limitation of the bias level that can be applied to the DUT. It can be observed from Figure 4 that the measured noise increases above 10 kHz. This can be explained by the fact that, because of the compensation network, the gain in the first stage decreases as the frequency increases, and the equivalent input noise increases because the relative weights of the noise introduced by R

_{D}

_{1}, R

_{D}

_{2,}and OA

_{1}increase [14].

^{+}B

_{p}un

^{+})-based barrier backside illuminated device. It was grown on a GaAs substrate with GaAs and InAs Si-doped layers using molecular beam epitaxy. The architecture details of the investigated structure are shown in Figure 5. It consisted of four main layers, which were additionally supplemented by gradient layers. The absorber, which is the main layer on which the radiation is absorbed, was non-intentionally doped (n.i.d.) with n-type conductivity. The bandgap barrier for electrons was made using AlAsSb. The n

^{+}contact placed at the bottom was made of a highly Si-doped InAs

_{1-x}Sb

_{x}layer. To reduce tunneling currents and decrease the maximum electric field occurring on the junction the additional graded Si-doped InAs

_{1−x}Sb

_{x}layer was sandwiched between the n+ contact and the absorber. A Be-doped (p-type) AlSb barrier was used to cap the absorber layer. The Be-doped (8 × 10

^{18}cm

^{−3}InAs

_{1-x}Sb

_{x}contact layer was applied to the top of the structure. Thanks to this construction both dark current and noise were reduced. The ohmic contact to the structures was performed by etching followed by Au/Ti metallization. The overall structure with the contacts was closed inside a metal TO-8 package.

_{0}A) of commercially available InAs

_{x}Sb

_{1−x}diodes varies from about 4 to 60 Ω cm

^{2}(with 0 ≤ x ≤ 0.36, T = 300 K) [24]. The tested detector was optimized for the 5 µm wavelength and mounted on a thermoelectric cooler to make it possible to improve its overall performance by operating at lower temperatures. In our experiments, we used a dedicated PID thermoelectric cooler controller to set up and precisely stabilize the temperature during measurements. Moreover, to dissipate heat from the “hot” side of the TEC, it was placed on a large-area aluminum radiator using thermoconductive paste. At zero bias voltage and A = 0.01 mm

^{2}, the R

_{0}A product of our sample at 300 K was about 7 mΩ·cm

^{2}. Further electro-optical details about the detector can be found in [25].

**Figure 5.**The architecture of the investigated InAsSb-based barrier IR detector. Reprinted with permission from Ref. [25]. 2023, SPIE.

^{α}in situations in which it is dominant in the overall examined frequency range, where the α parameter has a value close to 1. The detailed behavior of the spectral noise in this type of photodetector has already been described in the literature [26], and it is outside the scope of this work.

_{sh}

_{,}I

_{g−r}, I

_{diff}and I

_{tun}are the shunt, generation-recombination, diffusion and tunneling dark current components, respectively, with the corresponding noise coefficients α

_{sh}, α

_{g−r,}α

_{diff}and α

_{tun}. Depending on the photodiode construction and its operating point (temperature and bias voltage), each of the above noise sources will affect the total noise differently (i.e., the noise coefficients will take different values). Based on the results described in [26,27,28], the inset region in Figure 9 shows that the diffusion and g − r current components predominate in the detector’s dark current in the low- and mid-voltage ranges (A,B). Meanwhile, at high voltage bias (C), the tunneling current components predominates. However, there is no correlation between the total dark current and the measured 1/f noise PSD. This suggests that there are some current components other than diffusion that could have a higher influence on 1/f noise at different levels of bias voltage.

## 5. Conclusions

## Author Contributions

## Funding

## Institutional Review Board Statement

## Informed Consent Statement

## Data Availability Statement

## Conflicts of Interest

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**Figure 2.**Proposed TIA amplifier. The component types and their values are listed in Table 1.

**Figure 4.**Test results when using resistances at room temperature as DUTs. Tests were performed on 100 Ω and 1 kΩ resistances using two different values for the feedback resistance R

_{R}. The continuous black lines represent the expected noise, that is, the thermal current noise generated when using the resistances as DUTs.

**Figure 9.**Current noise at 10 Hz vs. bias current. The I-V characteristics of the device at 280 K from Figure 5 are reported in the inset.

**Table 1.**Component list for the circuit in Figure 2.

Component | Name | Type/Value |
---|---|---|

J_{1}, J_{2} | Low-noise JFET differential pair | Interfet IF3602 |

OA_{1} | Low-noise operational amplifier | OPA227 |

OA_{2} | Low-noise operational amplifier | ADA4625 |

OA_{3} | Low-noise operational amplifier | OPA140 |

R_{1}, R_{2} | 0.1%, ¼ W, SMD 1206 | 10 Ω, 1 kΩ |

R_{D}_{1}, R_{D}_{2} | 0.1%, 1 W, metallic film | 2 kΩ |

R_{SS} | 1%, 1 W, metallic film | 1.65 kΩ |

R_{A}_{1}, R_{A}_{2} | 0.1%, ¼ W, SMD 1206 | 1 MΩ |

C_{A}_{1} | Polyester | 22 µF |

C_{A}_{2} | Polyester | 10 µF |

Q_{1} | Bipolar transistor | ZTX450 |

R_{C}_{1}, C_{C}_{1} | ---- | 470 Ω, 470 nF |

R_{C}_{2}, C_{C}_{2} | ---- | 100 Ω, 22 nF |

R_{T} | Multitrim potentiometer | 100 kΩ |

R_{o} | 0.1%, ¼ W, SMD 1206 | 10 MΩ |

R_{R} | 0.1%, ¼ W, SMD 1206 | 1 kΩ or 10 kΩ |

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## Share and Cite

**MDPI and ACS Style**

Achtenberg, K.; Scandurra, G.; Mikołajczyk, J.; Ciofi, C.; Bielecki, Z.
Transimpedance Amplifier for Noise Measurements in Low-Resistance IR Photodetectors. *Appl. Sci.* **2023**, *13*, 9964.
https://doi.org/10.3390/app13179964

**AMA Style**

Achtenberg K, Scandurra G, Mikołajczyk J, Ciofi C, Bielecki Z.
Transimpedance Amplifier for Noise Measurements in Low-Resistance IR Photodetectors. *Applied Sciences*. 2023; 13(17):9964.
https://doi.org/10.3390/app13179964

**Chicago/Turabian Style**

Achtenberg, Krzysztof, Graziella Scandurra, Janusz Mikołajczyk, Carmine Ciofi, and Zbigniew Bielecki.
2023. "Transimpedance Amplifier for Noise Measurements in Low-Resistance IR Photodetectors" *Applied Sciences* 13, no. 17: 9964.
https://doi.org/10.3390/app13179964