Novel High Step-Up DC–DC Converter with Three-Winding-Coupled-Inductors and Its Derivatives for a Distributed Generation System

Abstract

A novel step-up DC-DC converter with a three-winding-coupled-inductor which integrates a coupled-inductor and voltage-boost techniques for a distributed generation system is proposed in this paper. The two windings of the dotted terminal connection are charged by the input source; the proposed converter utilized smaller turn ratios, and can achieve higher gain when the active switch is turned on. The passive lossless clamped circuits not only can absorb the leakage energy, but also lower the switch voltage stresses; additionally, the reverse-recovery problem of diodes can be reduced to improve the system efficiency. Furthermore, the voltage stress of the output capacitor is reduced. The operating principle and corresponding theoretical analyses are discussed in detail. Finally, an experimental prototype with 50 kHz switching frequency, 40 V input voltage, 380 V output voltage and 400 W output power is set up to verify the validity of the proposed converter.

1. Introduction

In recent times, in order to reducing pollution caused by fossil fuels, new, clean, renewable energy systems have been rapidly developed. Multilevel inverters with higher performance and corresponding analysis methods are presented, and have been widely applied for many circumstances [1,2]. But new renewable energies can’t provide higher DC voltage; DC-DC converters with high step-up voltage gains and high efficiencies must be connected between lower input DC voltage and inverters [3,4,5,6,7].

In theory, the basic boost converter with unlimited voltage gain is a preferable candidate, but the higher voltage gain will use the higher duty cycle ratio, which will result in many serious problems such as the reverse-recovery of diodes, higher losses, and higher voltage stress across passive components [8,9]. Boost converters in series are presented to realize the higher voltage gain, but this results in complicated systems and influences reliability due to the presence of additional switches and control units [10,11,12]. Quadratic boost converters based on cascaded technology can improve voltage gain and are applied extensively, but two switches and control circuits in quadratic boost converters increase the complexity and cost of systems, and the voltage gain still needs to be improved to meet many application circumstances.

Some DC-DC converters with coupled-inductor cells are presented to realize higher voltage gains [13,14,15,16,17,18,19,20,21,22,23,24,25]. In [13], the converter gain is achieved via a switched capacitor, but this increases the conduction loss and reduces efficiency. In [14], the voltage gain of this converter was extend by switched inductor technology; however, the voltage stress on the switches will stay at a high level, and thus, the active switches induce serious conduction losses. In [15,16,17], a voltage lift technique has been proposed. By the voltage-double capacitor, the voltage-boost ability is increased; unfortunately, the capacitor will suffer high voltage and current stresses. In [18], the switched capacitor and switched inductor technique are used to realize high step-up voltage gains, but they lead to larger power losses. The improved coupled inductor DC-DC converters with two winding and three winding coupled-inductor cells are presented to realize higher voltage gains, but the switches are placed at the output diode to higher switch voltage stresses [19,20].

By adjusting the turn ratio of a coupled inductor, the converter can obtain a high step-up voltage gain in [21], but, on the down side, the leakage inductor will cause a high voltage spike on the active switches. To resolve this weakness, an active clamp circuit is presented in [22,23]. However, the active clamp circuit does not achieve the step-up voltage, reduces efficiency, and adds extra cost. In [24,25,26], the converter can recycle the energy of the leakage inductor and improve the performance by using a negative clamp circuit to increase efficiency.

In this paper, a novel high step-up converter with a smaller turn ratio is proposed, and can achieve higher gain through coupled inductor technology and voltage lift technology, which will be beneficial to connect the power grid and transmit renewable energy under broader application circumstances. The passive lossless clamped circuits not only can absorb the leakage energy, but also lower the switch voltage stresses; additionally, the reverse-recovery problem of diodes can be reduced to improve the system efficiency. In addition, the voltage stress of the output capacitor is reduced. The operating principle and corresponding theoretical analyses are discussed in detail. Finally, an experimental prototype with 50 kHz switching frequency, 40 V input voltage, 380 V output voltage and 400 W output power is set up to verify the validity of the proposed converter.

2. Proposed High Step-Up Converter and Corresponding Operating Principles

The integration processes of the proposed converter are shown in Figure 1. Figure 1a shows the conventional boost DC-DC converter. To improve the voltage gain, a two-winding coupled inductor boost converter is structured by replacing inductor L₁ with two winding coupled inductors, whose topology is shown in Figure 1b. Through the addition of auxiliary diodes (D₁ and D₂) and capacitors (C₁ and C₂), the voltage-double two-winding coupled inductor boost converter are presented to further produce higher voltage gain, as shown in Figure 1c. So, through adding one winding N₃ and the corresponding voltage-double capacitors and diodes, the proposed three-winding coupled inductor high step-up converter is constructed by a three-winding coupled inductors, a switch S, four diodes D₁–D₄, and four capacitors C₁–C₄, as shown in Figure 1d. Meanwhile, a passive clamped circuit which is made of C₁ and D₁ can suppress the voltage spike on the switch caused by inductor leakage, and recycle the inductor leakage energy. The capacitor C₂ is used to step-up and reduced the voltage stress on the diode D₃. The voltage stress of the capacitor C₃ can be reduced by connection with the capacitor C₄ in series.

Three-winding coupled-inductor cell can be modeled as an ideal transformer, and the following conditions are assumed:

(1)

The capacitors in this converter are large enough that the voltages of those capacitors can be considered constant in one period.

(2)

Switches and diodes are considered as ideal, and will not be discussed in transient states.

(3)

The coupled inductor’s coupling coefficient K is equal to L_M/(L_M + L_K).

Figure 2 shows waveforms in one switching period in continuous conduction mode (CCM). The corresponding operation modes are described as follows:

(1) Mode I [t₀, t₁]: at t = t₀, the switch S is turned on. D₁, D₂ and D₄ are turned on, and D₃ is off as shown in Figure 3a. In this interval, the power source charges the magnetizing inductor L_M and the leakage inductor L_K, and the currents $i_{L_M}$ and $i_{L_K}$ increase linearly. The current $i_{N_2}$ is increased too. Meanwhile, the capacitor C₂ is charged by the clamped capacitor C₁, the source, and the winding N₃. The capacitor C₄ is powered by the winding N₂. The capacitor C₃ and the secondary-side winding N₂ provide their energy to the load R. When $i_{D_2}$ is equal to zero at t = t₁, D₂ is naturally turned off, and the operating mode ends.

(2) Mode II [t₁, t₂]: during this time interval, the switch S is still turned on. The current-flow path is shown in Figure 3b. D₃ is still turned off. The currents $i_{L_M}$ and $i_{L_K}$ are still rising, and the winding N₃ of the coupled inductor is charged by the input source. The energy of the load is supplied by the secondary-side winding N₂ and the capacitor C₃. When the switch is turned off, this operating mode ends.

(3) Mode III [t₂, t₃]: at t = t₂, the switch S is turned off. Diodes D₁ and D₃ are turned on. Meanwhile, diodes D₂ and D₄ are turned off in Figure 3c. The energy of the clamped capacitor C₁ is provided by the primary side N₁ and secondary side winding N₃ of the coupled inductor; thus the voltage of the switch S is clamped by the input source and clamped capacitor C₁. The capacitor C₃ is charged by the source and primary-side winding N₁ and the secondary-side winding N₂ of the coupled inductor and the multiplier capacitor C₂. The energy of the load is supplied by the input source, the secondary-side winding N₁ and the primary-side winding N₂ of the coupled inductor and the multiplier capacitor C₂ the capacitor C₄. When the current $i_{D_1}$ is equal to zero at t = t₃, diode D₁ is turned off, ending operating mode.

(4) Mode IV [t₃, t₄]: during this time interval, the switch S is still turned off. The current-flow path is shown in Figure 3d.The input source connects in series with the primary-side winding N₁, the secondary-side winding N₂ of the coupled inductor, the capacitor C₂ and the capacitor C₄ to provide energy for the load through D₃. This mode ends at t = t₄ when the switch S is turned on at the beginning of the next switching period.

3. Steady-State Analysis

3.1. Analysis in Continuous Conduction Mode

In modes I and II based Figure 3a,b, the following equations can be obtained as follows:

$$V_{L_M}^{\mathrm{I}}=V_{L_M}^{\mathrm{II}}=\frac{N_{1}K{V}_{\mathrm{in}}}{N_1-N_{3}K}$$

(1)

$$V_{L_K}^{\mathrm{I}}=V_{L_K}^{\mathrm{II}}=\frac{N_1(1-K)V_{\mathrm{in}}}{N_1-N_{3}K}$$

(2)

$$V_{C_2}=\frac{N_1{V}_{\mathrm{in}}}{N_3-N_{3}K}+V_{C_1}$$

(3)

$$V_{C_4}=\frac{N_{2}K{V}_{\mathrm{in}}}{N_1-N_{3}K}$$

(4)

During the time of modes III and IV, the following equations can be expressed based on Figure 3c,d:

$$V_{C_1}=V_{L_M}^{\mathrm{III}}+V_{L_K}^{\mathrm{III}}+V_{C_1}=V_{L_M}^{\mathrm{IV}}+V_{L_K}^{\mathrm{IV}}+V_{C_1}$$

(5)

$$V_o=V_{\mathrm{in}}+(1+N_{3}K)(V_{L_M}^{\mathrm{III}}+V_{L_K}^{\mathrm{III}})+V_{C_2}+V_{C_4}=V_{\mathrm{in}}+(1+N_{3}K)(V_{L_M}^{\mathrm{IV}}+V_{L_K}^{\mathrm{IV}})+V_{C_2}+V_{C_4}$$

(6)

Using the volt–second balance principle on L_M and L_K yields:

$${\displaystyle \underset{t_0}{\overset{t_1}{\int }}{V}_{L_M}^{\mathrm{I}}{d}_t}+{\displaystyle \underset{t_1}{\overset{t_2}{\int }}{V}_{L_M}^{\mathrm{II}}{d}_t}+{\displaystyle \underset{t_2}{\overset{t_3}{\int }}{V}_{L_M}^{\mathrm{III}}{d}_t}+{\displaystyle \underset{t_3}{\overset{t_4}{\int }}{V}_{L_M}^{\mathrm{IV}}{d}_t}=0$$

(7)

$${\displaystyle \underset{t_0}{\overset{t_1}{\int }}{V}_{L_K}^{\mathrm{I}}{d}_t}+{\displaystyle \underset{t_1}{\overset{t_2}{\int }}{V}_{L_K}^{\mathrm{II}}{d}_t}+{\displaystyle \underset{t_2}{\overset{t_3}{\int }}{V}_{L_K}^{\mathrm{III}}{d}_t}+{\displaystyle \underset{t_3}{\overset{t_4}{\int }}{V}_{L_K}^{\mathrm{IV}}{d}_t}=0$$

(8)

From Equations (1)–(10), the voltage gain can be expressed as:

$$M_{CCM}=\frac{2N_1-N_{3}K+N_{2}K}{(1-D)(N_1-N_{3}K)}=\frac{2-n_{31}K+n_{21}K}{(1-D)(1-n_{31}K)}$$

(9)

where $n_{21}=\frac{N_2}{N_1}$ and $n_{31}=\frac{N_3}{N_1}$ are the turns ratios of the proposed converter.

The relationship between the voltage gain, the duty ratio, and the coupling coefficients of coupled inductor is shown in Figure 4a. It shows that the coupling coefficient results in a decline of voltage gain. It can be seen clearly that a smaller turns ratios can achieve a higher gain. If the coupled coefficient K is equal to 1 without considering the impact of the leakage inductance of the coupled inductor, then the ideal voltage gain can be expressed as:

$$M_{CCM}=\frac{2N_1-N_3+N_2}{(1-D)(N_1-N_3)}=\frac{2-n_{31}+n_{21}}{(1-D)(1-n_{31})}$$

(10)

In Figure 4b, the curve shows voltage gain comparisons among the proposed converter and the converters in [19,20] at CCM operation under N₃:N₂:N₁ = 2:1:4. As shown in Figure 4b, the proposed converter has the higher voltage gain compared to the converters in [19,20], when using the same turn ratios and duty ratios.

In the proposed converter, the following inequalities are satisfied:

$$N_1>N_3>0$$

(11)

When the winding turns N₂ and N₃ are changed, some converters are derived as follows.

In the proposed converter, the diode D₄ is turned on during one switching period when N₃ = 0. At the same time, the capacitor C₄ can be ignored because the across its voltage is equal to zero. Therefore Figure 5 can be obtained. The voltage gains of converters in Figure 5a–c under different winding turns are listed in

Table 1. Derived converter.

Condition	Voltage Gain	Converter
$N_1>0,N_2=0,N_3>0$	$\frac{2N_1-N_3}{(1-D)(N_1-N_3)}$	Figure 5a
$N_1>0,N_2>0,N_3=0$	$\frac{2N_1+N_2}{(1-D)N_1}$	Figure 5b
$N_1>0,N_2=0,N_3=0$	$\frac{2}{(1-D)}$	Figure 5c

.

3.2. Voltage Stress Analysis

The switch voltage stresses, voltages of capacitors C₁–C₄ and diodes D₁–D₄ are expressed as follows:

$$V_S=\frac{1}{1-D}{V}_{\mathrm{in}}$$

(12)

$$V_{D_1}=V_{C_1}=\frac{D}{1-D}{V}_{\mathrm{in}}$$

(13)

$$V_{D_2}=\frac{V_{\mathrm{in}}}{(1-D)(1-n_{31})}$$

(14)

$$V_{D_3}=V_{C_3}=\frac{(1+n_{21})V_{\mathrm{in}}}{(1-D)(1-n_{31})}$$

(15)

$$V_{D_4}=\frac{n_{21}{V}_{\mathrm{in}}}{(1-D)(1-n_{31})}$$

(16)

$$V_{C_2}=\frac{(1-n_{31}D)V_{\mathrm{in}}}{(1-D)(1-n_{31})}$$

(17)

$$V_{C_4}=\frac{n_{21}{V}_{\mathrm{in}}}{1-n_{31}}$$

(18)

4. Derivative Converters

According to the different connection position, the proposed circuits can be derived in Figure 6. When applying series-connected voltage-lift networks (VLN), the voltage gain is enhanced by adding some VLN modules in Figure 6a. The voltage gain equation is derived from:

$$M_{CCM}=\frac{2N_1-N_2+N_3+\cdots +N_n}{(1-D)(N_1-N_2)}$$

(19)

Likewise, the voltage gain can be directly enlarged by applying a multiple windings coupled-inductor and multiple diode-capacitor networks to the proposed converter, as shown in Figure 6b. The voltage gain equation can be expressed as follows:

$$M_{CCM}=\frac{2N_1-N_2+N_3+\cdots +N_n}{(1-D)(N_1-N_2)}$$

(20)

The converter with a high step-up voltage gain also can be obtained by an input-parallel-output-series technique. In this technique, the primary and second winding of the coupled-inductor are parallel charged when switch is turned on, and discharged in series when it is turned off. The improved topology is shown in Figure 6c. The detailed features of the aforementioned improved converters are listed in

Table 2. The component numbers and voltage gains of derived converters.

Diode Number	Capacitor Number	Winding Number	Voltage Gain	Converter
n + 1	n + 1	n	$M_{CCM}=\frac{2N_1-N_2+N_3+\cdots +N_n}{(1-D)(N_1-N_2)}$	Figure 6a
2n − 2	2n − 2	n	$M_{CCM}=\frac{2N_1-N_2+N_3+\cdots +N_n}{(1-D)(N_1-N_2)}$	Figure 6b
6	6	5	$M_{CCM}=\frac{4N_1-2N_3+N_2}{(1-D)(N_1-N_3)}$	Figure 6c

. The voltage gain equation can be expressed as follows:

$$M_{CCM}=\frac{4N_1-2N_3+N_2}{(1-D)(N_1-N_3)}$$

(21)

5. Experimental Verifications

An experimental prototype with 50 kHz switching frequency, 40 V input voltage, 380 V output voltage and 400 W output power is set up and corresponding parameters have been listed in

Table 3. System specifications of the proposed converter.

Specifications	Value
Input voltage Vin	40 V
Output voltage V0	380 V
Rated power P0	400 W
Switching frequency fs	50 kHz
MOSFET Switches S	IRFP4668
Diodes D0, D1, D2, D3	MUR460
Capacitor C3	470 μF, 0.1 μF
Capacitor C1, C2, C4	10 μF
Coupled inductors core	Core-EE55
Turn ratios	N1:N2:N3 = 30:16:16
Winding value	N1 = 542 μH, N2 = 149.5 μH, N3 = 149.5 μH

.

Duty cycle of the switch is about 0.58, and the switch voltage are about 87 V, as shown in Figure 7a. The switch voltage of active switch S and the voltage of primary winding N₁ of the coupled inductor are shown in Figure 7b; it can be seen that the winding N₁ are charged when the active switch S is turned on. The peak voltages of diodes (D₁, D₂, D₃ and D₄) are about 97 V, 193 V, 255 V, and 90 V, as shown in Figure 7c–f. From Figure 7c–f, it can be also seen that the current waveforms of diodes (D₁, D₂, D₃ and D₄) are consistent with theory analysis. It shows that the currents of diodes (D₁, D₂, D₃ and D₄) are decreased to zero quickly when the diodes are turned off, which means that losses can be reduced. The winding currents of the three-winding coupled inductor is shown in Figure 7a,g. As shown in Figure 7a,g, the operation states of three windings (N₁, N₂, N₃) are consistent with theory analysis through the switch sequence of active switch S.

Figure 8 shows the tested efficiency curve of the proposed converter. The power range is 40–400 W, and the efficiency is tested per 40 W. From Figure 8, it can be seen that the efficiency of the proposed converter is about 91.9% when output power is 400 W, while the maximum efficiency of proposed converter is about 95.6% when output power is 160 W.

6. Conclusions

In this paper, a novel three-winding coupled-inductor dc-dc converter is proposed with analysis of its operating modes. The proposed converter can produce a higher voltage gain. Also, a passive lossless clamping circuit which belongs to the part of the step-up is used to suppress voltage spikes across the switch. The proposed converter has the following advantages:

(1)

It utilized smaller turns ratios, and can therefore achieve a higher voltage conversion gain in a normal small duty cycle;

(2)

A reduced magnetic size can be applied in this converter to achieve high voltage gain. It is suitable for DG system based on distributed generation system;

(3)

Two diodes avoid the reverse-recovery problem by achieving turn-off naturally;

(4)

The presented converter can recycle the leakage inductance energy to improve performance.

(5)

The voltage stress of the output capacitor is reduced.