Mitigation of Switching Ringing of GaN HEMT Based on RC Snubbers

Abstract

Gallium nitride high electron mobility transistors (GaN HEMTs), characterized by their extremely high switching speeds and superior high-frequency performance, have demonstrated significant advantages, and gained extensive applications in fields such as aerospace and high-power-density power supplies. However, their unique internal architecture renders these devices highly sensitive to circuit parasitic parameters. Conventional circuit design methodologies often induce severe issues such as overshoot and high-frequency oscillations, which significantly constrain the realization of their high-frequency performance. To solve this problem, this paper investigates the nonlinear dynamic behavior of GaN HEMTs during switching transients by establishing an equivalent impedance model. Based on this model, a detailed analysis is implemented to elucidate the mechanism by which RC Snubber circuits influence the system’s resonance frequency and the amplitude at the resonant frequency. Through this analysis, an optimal RC Snubber circuit parameter is derived, enabling effective suppression of high-frequency oscillations during the switching transient of GaN HEMT. Experimental results demonstrate that the proposed design achieves a maximum reduction of 40% in voltage overshoot, shortens the ringing time to one-twentieth of the original value, and suppresses noise by 20 dB in the high-frequency range of 20 MHz to 30 MHz, thereby significantly enhancing the stability and reliability of circuit operation. Additionally, considering the heat dissipation requirements in high power density scenarios, this work optimizes the layout of devices, and heat sinks to maintain operational temperatures within safe limits, further mitigating the impact of parasitic parameters on overall system performance.

1. Introduction

Gallium nitride (GaN), as a wide bandgap semiconductor (WBG) material, possesses outstanding properties such as high electron mobility, high temperature resistance, and high voltage tolerance. Gallium nitride high electron mobility transistor (GaN HEMT), serving as a typical representative of third-generation WBG semiconductor devices, exhibits unique advantages in high-frequency and high-power-density applications such as aerospace, electric vehicles, fast charging, renewable energy, and data centers [1]. Compared with traditional silicon metal-oxide-semiconductor field-effect transistors (Si MOSFETs) of the same voltage rating, GaN HEMTs feature faster switching speeds. However, due to the effect of circuit parasitic parameters, high-speed switching operations may trigger high-frequency oscillation and voltage overshoot, posing threats to device performance and long-term reliability [2,3].

High-frequency oscillations and voltage overshoot limit the performance of GaN HEMT. Parasitic inductance and capacitance in the package induce oscillations at high $dv/dt$ and $di/dt$, which will accelerate the aging of WBG devices. High voltage overshoot may cause mechanical failure and the dielectric breakdown in WBG devices. High-speed switching operations will generate high-frequency oscillations, which can lead to electromagnetic interference (EMI) issues that can disrupt control and communication systems. High-frequency oscillations may also cause ground potential fluctuations, consequently making it difficult to comply with electromagnetic compatibility (EMC) standards. Additionally, the high switching speed of WBG devices exacerbates the negative effects of parasitic parameters, thereby imposing more stringent requirements on circuit design. Directly applying circuit design methodologies originally developed for Si MOSFETs may induce switching oscillations and voltage overshoot in GaN HEMTs, consequently jeopardizing system stability [4]. GaN HEMTs encounter more challenges than Silicon Carbide Metal-Oxide-Semiconductor Field-Effect Transistors (SiC MOSFETs) [5]. The coupling effects between parasitic parameters and high-speed switching in GaN HEMTs are significantly more pronounced, resulting in failure risks that substantially exceed those observed in SiC MOSFETs. Due to their compact dimensions and high power density characteristics, GaN HEMTs experience exacerbated electromigration and solder joint fatigue issues. Under voltage oscillations and overshoot conditions, these devices undergo accelerated performance degradation and significant lifetime reduction. In high-frequency and high-power application scenarios, GaN HEMTs impose extremely stringent requirements on parasitic parameters. Once the printed circuit board (PCB) trace inductance exceeds 5 nH, it readily induces oscillations [6,7,8,9,10,11]. Unlike SiC MOSFETs, GaN HEMTs lack intrinsic avalanche capability and are susceptible to irreversible breakdown when subjected to voltage overshoot. This inherent characteristic necessitates the incorporation of additional protective circuits to mitigate voltage overshoot in high-frequency and high-power scenarios, resulting in increased system costs and heightened design complexity. In practical applications, fully leveraging the performance advantages of GaN HEMTs requires many targeted measures, including device design, circuit optimization, and system layout to effectively address the challenges posed by voltage oscillations and overshoot [12,13,14].

Several methods are employed to suppress high-frequency oscillations, including advanced packaging technologies, optimized gate drive, RC Snubber, parasitic parameters, and layout optimization techniques. The advantages and disadvantages of these methods are summarized in

Table 1. High-frequency oscillation suppression methods in power electronic systems.

Method	Specific Technology	Principle	Advantages	Limitations	References
Device-Level Optimization	1. GaN HEMT (near-field scanning, Kelvin pins, surface-mount packaging)	Reduce packaging-induced parasitic parameters	Lowers parasitic inductance/capacitance; improves high-frequency switching performance	High cost	[12,13,14]
	2. Cascode GaN HEMT (stacked chips, compensation capacitors)	Eliminate oscillations and reduce EMI via parasitic compensation	Effective oscillation suppression; low EMI emissions	High manufacturing cost; complex process; strict high-precision requirements	[15]
Module-Level Optimization	1. Traditional layout optimization (widening/shortening patterns, reducing bond wire length)	Lower power loop parasitic inductance	Simple implementation; low cost; compatible with most module structures	Limited parasitic reduction; cannot fully optimize internal inductance	[16,17,18]
	2. New module technologies (decoupling capacitors, dual-source designs)	Further mitigate power loop parasitics beyond traditional layout	Enhances parasitic suppression; adaptable to high-power scenarios	High requirements for testing equipment	[16,17]
	3. Copper pillar connection optimization (layout reconstruction)	Reduce inductance via structural optimization of interconnections	Suppresses oscillations; reduces switching losses	Fails to fully optimize internal inductance; risks dynamic current sharing in multi-chip parallelism	[19]
Driver-Level Optimization	1. High-speed response and $dv/dt$ immune converter	Achieve slew rate acceleration, noise suppression, low delay design	No complex processing; low latency; suitable for GaN power ICs	Effectiveness for discrete devices unproven	[24]
	2. AOV and TBLS	Mitigate $dv/dt$ interference via active control and anti-interference shifters	Strong anti-interference capability; effective for GaN HEMT $dv/dt$ issues	Higher losses than gate resistance; needs dynamic driving optimization	[25]
Passive Snubber Circuits	1. RC Snubber	Absorb oscillation energy via external resistors/capacitors	No complex control; easy integration; low cost; minimal impact on switching speed	Increases current stress; lacks parasitic modeling	[26,27,28]
	2. RCD Snubber	Improve energy absorption speed via parallel diodes	Faster suppression than RC Snubber	Affected by diode reverse recovery; significant switching loss	[28]
	3. Clamped RCD Snubber	Restrain capacitor voltage to DC bus voltage; operate only during oscillations	Negligible impact on speed/losses; better suppression	Lacks parasitic modeling and systematic experimental analysis	[28]
Snubber Parameter Optimization	1. Phase margin maximization (third-order transfer function model)	Optimize Snubber resistance via closed-loop phase margin analysis	Reduces overshoot/losses; improves EMC; high power density	Fails to balance suppression and dynamic response; ignores non-PCB parasitics	[29]
	2. Root locus analysis (seventh-order equivalent model)	Suppress oscillations via dipole elimination (damping ratio > 0.4)	Effect oscillation suppression; high model accuracy	Complex high-order analysis; relies on designers’ expertise	[30]
PCB Layout Optimization	1. Horizontal/vertical layout and single-layer optimization	Reduce inductance via field cancellation/self-cancellation	Lowers inductance, overshoot, and losses	Ignores GaN nonlinear capacitances and coupling effects	[2]
	2. Balanced electrical-thermal layout	Co-optimize inductance and thermal impedance; reduce peak temperature	Adaptable to bidirectional conversion; solves heat dissipation issues	Relies on empirical formulas; lacks precise parasitic quantification	[31]
	3. Inner-layer power loop structure	Minimize loop area via multi-layer inner return paths	Optimal stray inductance; high calculation accuracy	Limited applicability; manufacturer data errors	[32]

. Optimizing the packaging technologies can reduce parasitic inductance in devices, thereby mitigating switching oscillations. Device-level parasitic parameters in GaN HEMTs can be mitigated through near-field scanning, Kelvin connections, and surface-mount packaging [15]. At the module level, reducing bond wire length decreases inductance. Emerging packaging technologies minimize parasitic inductance through decoupling capacitors, dual-side cooling designs, hybrid packaging, and mutual inductance utilization [16,17,18]. In [19], copper pillars are optimized to reduce parasitic inductance, but it introduces the risks related to current imbalance. For application engineers, improving GaN HEMT performance through packaging enhancements poses universal challenges. WBG devices with fast switching speeds can cause severe oscillations, which increase the risk of gate breakdown [20,21,22,23]. For Si MOSFETs, increasing gate resistance is commonly employed to suppress switching oscillations. However, directly applying this approach to GaN HEMTs degrades their switching performance. Reference [24] proposes a driver-level converter design with the dual capabilities of high-speed response and noise immunity, enabling the effective suppression of high-frequency oscillations. In [25], it introduces Active Overdrive Voltage (AOV) control and Triple-Branch Level Shifter (TBLS) techniques to address the reliability issues caused by dv/dt-induced effects during the high-speed switching of GaN HEMTs. Current driver optimizations often require extra active devices, increasing complexity, cost, and losses, and its parameter constraints also hinder the global optimization of losses, cost, and electromagnetic compatibility (EMC). For the above studies, gate drive optimization techniques often require additional active components, which not only increase circuit design complexity and hardware costs, but also introduce additional system losses and EMC issues.

Compared to driver optimization or packaging improvements, RC Snubbers require no complex control circuit, merely external resistors and capacitors, enabling easier integration into existing systems and making them suitable for cost-sensitive applications [26]. RC Snubbers with configurable parameters offer layout-specific adaptability, effectively suppressing oscillations, minimizing impact on switching speed, and reducing additional switching losses [27]. In [28], RC, RCD, and clamped RCD Snubbers are compared. The RC Snubber absorbs turn-off charge but increases turn-on current stress. The RCD Snubber enhances the absorption speed using a parallel diode, though it is affected by reverse recovery. The clamped RCD regulates capacitor voltage to the DC bus level, operating exclusively during oscillations without affecting switching characteristics. Its diode reverse recovery can be resolved using fast-recovery diodes. Reference [28] exhibits limited depth in parasitic parameter modeling, merely qualitatively attributing oscillations to parasitic inductance and capacitance based on experimental observations. Furthermore, it only briefly addresses Snubber resistor design in its theoretical section, without an experimental analysis of how resistor parameters affect switching performance. In [29], it establishes a third-order transfer function model incorporating PCB and power device parameters through Laplace transform and determines the optimal Snubber resistor value via phase margin optimization, achieving a significant reduction in voltage overshoot and circuit losses. Nevertheless, the study does not explicitly address how to balance voltage overshoot suppression with a dynamic response speed. Reference [30] developed a high-frequency equivalent model with parasitic parameters based on a double-pulse test circuit, deriving a seventh-order system expression for drain–source voltage. Through an analysis of this seventh-order system, it quantitatively identifies effective RC parameter regions, demonstrating that a damping ratio exceeding 0.4 achieves satisfactory suppression. However, the parameter optimization process for this 7th-order system remains computationally complex and heavily reliant on theoretical expertise. Furthermore, the generalizability of the proposed methodology to higher-order systems requires further validation, as its applicability beyond GaN half-bridge configurations, such as to other topological structures or device types, needs additional investigation.

PCB layout optimization is crucial for suppressing high-frequency oscillations and the voltage overshoot of GaN HEMTs. This method, when combined with an RC Snubber, achieves the enhanced suppression of oscillations and voltage overshoot. PCB parasitic parameters can significantly degrade RC Snubber effectiveness. Therefore, optimizing PCB layout to reduce parasitic inductance and shorten current paths is essential for enhancing Snubber performance. In high-frequency and high-power scenarios, PCB parasitic parameters become even more critical in RC Snubber design. In [2], the impacts of common-source inductance and power loop inductance are analyzed, with comparisons between horizontal and vertical layouts. Simultaneously, an optimized single-layer design is proposed to reduce parasitic inductance and voltage overshoot. Nevertheless, it neglects the nonlinear capacitances of GaN HEMT and the coupling interactions between LC and RC circuits. According to [31], the proposed novel layout achieves parasitic inductance in the power loop comparable to or lower than conventional designs, significantly reducing the peak temperature rise across operating modes. This approach optimizes both electrical and thermal performance by simultaneously minimizing power loop parasitic inductance and thermal impedance. Nevertheless, it does not quantify specific parasitic parameter values or their impact on switching losses through precise methods such as finite element analysis. Reference [32] analyzes three power loop structures and implements minimal current return paths through multilayer PCB design, thereby reducing parasitic inductance. Meanwhile, the study derives magnetic field and flux distributions based on the Biot–Savart law to propose an analytical method for calculating stray inductance. These methods have limited practical applicability and may suffer from potential errors due to packaging the data provided by manufacturers. Previous studies have not adequately considered the mutual effects between RC Snubber design and PCB layout optimization.

To solve the above problems, this paper investigates the coupling effects between RC Snubber design and PCB layout optimization to address the practical challenges in GaN-based half-bridge circuits. Experimental results demonstrate that the designed circuit effectively suppresses high-frequency oscillations, significantly reduces overshoot voltage, shortens the ringing time, and attenuates noise at high frequencies during the switching transients of GaN HEMTs. Compared to existing studies, the main contributions of this paper are summarized as follows:

• An equivalent impedance model is constructed for GaN HEMT-based half-bridge circuits during switching oscillations, incorporating parasitic parameters from packaging, PCB, and switching devices. The switching oscillations are decoupled into current and voltage oscillation subsystems, with transfer functions and key parameters derived to characterize oscillatory behavior. Based on this model, the mechanism of RC Snubbers is elucidated, and the influence of RC parameters on the circuit are quantified through simulation and experiments. A multi-level optimization strategy is proposed: calculating initial capacitance values based on target frequency bands, employing algorithms to balance losses and suppression effectiveness, dynamically adjusting parameters with experimental measurements, and incorporating robustness design. Post-optimization results demonstrate a maximum reduction of 40% in voltage overshoot, an oscillation duration shortened to 1/20 of the original value, and high-frequency noise (above 20 MHz) attenuated by 20 dB, significantly enhancing circuit stability.
• A co-optimization methodology for device and heatsink layout is proposed. Q3D simulations are employed to compute electromagnetic field distributions generated by device arrangements, where parasitic inductance and capacitance coupling paths are disrupted through optimized component sequencing and routing strategies. By holistically addressing thermal management and EMI suppression requirements, this approach ensures thermal stability under high-power-density conditions while effectively reducing the negative impacts of parasitic inductance and capacitance on system performance.

The subsequent sections are structured as follows: Section 2 constructs the equivalent impedance model through theoretical derivation and characteristic analysis; Section 3 deeply analyzes the mechanism of RC Snubber and parameter optimization design; Section 4 focuses on the influence of loop inductance on switching ringing through Q3D simulation; Section 5 systematically evaluates the effects of RC Snubber and loop inductance on circuit performance through experimental verification; Section 6 summarizes the research achievements and outlines prospects.

2. Analysis of Switching Process and Oscillation Mechanism

This section elaborates on the switching process of GaN HEMTs, focusing on analyzing the circuit states during turn-on and turn-off oscillations. The GaN HEMT-based half-bridge circuit, as shown in Figure 1, incorporates parasitic parameters from both the devices and the power loop [33]. The power loop comprises the AC and DC loops. Parasitic parameters consist of drain parasitic inductances ($L_{d\_H}$, $L_{d\_L}$), source parasitic inductances ($L_{s\_H}$, $L_{s\_L}$), gate–drain capacitances ($C_{gd\_H}$, $C_{gd\_L}$), gate–source capacitances ($C_{gs\_H}$, $C_{gs\_L}$), drain–source capacitances ($C_{ds\_H}$, $C_{ds\_L}$), loop inductance ($L_{loop}$), and loop resistance ($R_c$). This circuit configuration is more aligned with the operating conditions found in practical applications.

2.1. Turn-On Process

The turn-on process can be divided into five phases: the turn-on delay phase ($t_0-t_1$), the current rising phase ($t_1-t_2$), the first voltage falling phase ($t_2-t_3$), the second voltage falling phase ($t_3-t_4$), and the turn-on oscillation phase ($t_4-t_5$). The variations in key voltages and currents during each phase are shown in Figure 2. During the turn-on oscillation phase, various parasitic parameters have a significant impact on the circuit, causing intense fluctuations in voltage and current across GaN HEMTs.

During the turn-on oscillation phase, the circuit can be represented by the model shown in Figure 3. The upper switch, acting as the active device, is fully turned on and can be approximated as the resistance ($R_{ds}$). The oscillation frequency during the turn-on oscillation phase exceeds 10 MHz, at which point the skin effect becomes significant. As a result, the impedance of conductors within the power loop increases at high frequencies, which is the primary cause of the rapid attenuation of oscillation energy during this phase. Then, an equivalent AC resistance ($R_{ac}$) can be introduced into the AC loop, as shown in Figure 3. The lower switch, acting as the passive device, remains off when the upper switch is operating. At this moment, it can be replaced by an LC network, which includes the output capacitance ($C_{oss\_L}$), $L_{d\_L}$, and $L_{s\_L}$. During the turn-on oscillation phase, the voltage oscillation on $V_{ds\_L}$ of the GaN HEMT is given by (1).

$$\begin{array}{c}{V}_{dc}=L_p{C}_{oss}{\displaystyle \frac{d^2{V}_{ds\_L}}{d{t}^2}}+\left(R_c+2R_{ac}\right)C_{oss}{\displaystyle \frac{d{V}_{ds\_L}}{dt}}+I_L\left(R_c+R_{ac}\right)+V_{ds\_L}\hfill \\ L_P=L_{d\_H}+L_{s\_H}+L_{d\_L}+L_{s\_L}+L_{loop}\hfill \\ C_{oss\_L}=C_{gd\_L}+C_{ds\_L}\hfill \end{array}$$

(1)

2.2. Turn-Off Process

The turn-off process is also divided into five phases: the turn-off delay phase ($t_0-t_1$), voltage rising phase ($t_1-t_2$), voltage rising phase ($t_2-t_3$), current falling phase ($t_3-t_4$), and turn-off oscillation phase ($t_4-t_5$). Similarly, the variations in key voltages and currents during each phase are illustrated in Figure 4. During the turn-off oscillation phase, the mutual coupling effects among parasitic parameters (including parasitic capacitance, junction capacitance, and stray inductance) become more pronounced. The dynamic nonlinear characteristics of these parasitic parameters cause a strong frequency modulation effect on the circuit, leading to irregular oscillations in both voltage and current waveforms. Meanwhile, sharp high-frequency spikes are superimposed on the voltage waveform, which oscillates rapidly with high-frequency noise. Both instantaneous voltage and current values experience multiple significant jumps within a very short time, resulting in severe waveform distortion and energy oscillation.

For the turn-off oscillation phase, the equivalent circuit is shown in Figure 5. The upper switch, acting as the main control device, is fully turned off. In this case, the upper switch can be replaced by an LC network consisting of the output capacitance $C_{oss\_H}$, $L_{d\_H}$ and $L_{s\_H}$. Due to the resonant interaction between parasitic inductance and input–output capacitance of GaN HEMTs, distinct oscillation waveforms emerge during the turn-off phase. Similarly to the turn-on oscillation phase, the turn-off oscillation frequency exceeds 10 MHz, requiring $R_{ac}$ to account for the damping effect as well. However, the AC loop during the turn-off oscillation phase exhibits distinct characteristics compared to the turn-on oscillation phase. The position of $R_{ac}$ also differs, as shown in Figure 5. During the turn-off oscillation phase, in addition to the parasitic capacitance and inductance, the reverse conduction path inherent to the lower switch also participates in the power loop. At this stage, the voltage oscillation on $V_{ds\_H}$ is given by (2).

$$\begin{array}{c}{V}_{dc}=L_p{C}_{oss}{\displaystyle \frac{d^2{V}_{ds\_H}}{d{t}^2}}+\left(R_c+R_{ac}\right)C_{oss}{\displaystyle \frac{d{V}_{ds\_H}}{dt}}+V_{ds\_H}\hfill \\ L_P=L_{d\_H}+L_{s\_H}+L_{d\_L}+L_{s\_L}+L_{loop}\hfill \\ C_{oss\_H}=C_{gd\_H}+C_{ds\_H}\hfill \\ C_{oss\_L}=C_{gd\_L}+C_{ds\_L}\hfill \end{array}$$

(2)

3. RC Snubber Design

3.1. Analysis of the RC Snubber

The design objective of the Snubber circuit is to mitigate the adverse effects of the switching ringing on various circuit components by reducing the voltage and current rise/fall times during switching operations. The Snubber circuit, consisting of a resistor and a capacitor, strikes an optimal balance between cost and efficiency while aiming for a superior switching ringing suppression performance. The selection of the Snubber capacitor ($C_{snub}$) and Snubber resistor ($R_{snub}$) directly determines the effectiveness of the Snubber in optimizing circuit performance. $C_{snub}$ is selected based on the required amplitude of the voltage spikes to be suppressed and the allowable charging time. A larger $C_{snub}$ results in a more significant suppression effect on voltage spikes. However, it prolongs the charging time, which may degrade the circuit’s response speed. $R_{snub}$ should be chosen by considering both the capacitor charging time and the allowable power loss. A smaller $R_{snub}$ shortens the charging time of $C_{snub}$ but increases the power loss. The RC Snubber in this study is connected in parallel with the drain–source terminals of the power device, as shown in Figure 6. The operating cycle of the Snubber circuit, which directly corresponds to that of the half-bridge circuit, is divided into two phases: the turn-on and turn-off stages.

The working principle during the turn-on stage is illustrated in Figure 7a. When the switching device is turned on, the current begins to flow through it. At this point, the capacitor is discharged through the resistor and the discharging current gradually increases. This capacitor limits the rise rate of the current during the turn-on of the switching device, thereby reducing turn-on losses and EMI. Due to the presence of the capacitor, the voltage rise rate across the switching device is slowed down, effectively mitigating the voltage spike. During the discharging process, the resistor limits the discharging current to prevent excessive current from damaging the switching device or other components.

The working principle of the turn-off stage is illustrated in Figure 7b. When the switching device is turned off, the current flowing to this device rapidly drops to zero. The load inductor ($L_{load}$) provides freewheeling energy to the half-bridge, generating a reverse voltage spike across the switching device. At this point, $C_{snub}$ begins to charge, absorbing the energy of the circuit to suppress the voltage spike. During the charging process, the resistor dissipates part of the energy, further reducing the voltage spike. It also controls the charging speed of the capacitor to prevent excessive charging current from damaging the circuit components.

3.2. Acquisition of RC Snubber Parameters

The appropriate values of the key parameters $R_{snub}$ and $C_{snub}$ of the Snubber circuit need to be determined from actual circuit experiments. As shown in Figure 8, in the actual circuit, the RC Snubbers are connected in parallel across both the active and passive switches to reduce interference.

This paper establishes an impedance analysis model to determine accurate values for $R_{snub}$ and $C_{snub}$. Based on the circuit structure shown in Figure 8, this model incorporates various parasitic parameters in the power loop, ultimately producing the equivalent circuit model for the turn-on oscillation stage (Figure 9a) and the turn-off oscillation stage (Figure 10a). During the switching oscillation stage, the upper switching device can be modeled as a series combination of the parasitic inductor $L_{sd}$ ($L_{s\_H}$ + $L_{d\_H}$) and the device’s on-resistance ($R_{ds}$), while the lower switching device can be modeled as $L_{sd}$ and $C_{oss}$. Since the upper and lower switching devices in the circuit structure of this paper are identical, parameters with the same values are not differentiated. Additionally, the effects of the parameters $R_c$, $R_{ac}$, $R_{snub}$, and $C_{snub}$ must be considered. During the turn-on oscillation stage, high-frequency oscillation occurs in the loop of the lower switching device. In this case, $R_{ac}$ is in series with the lower switching device, and the RC Snubber for the upper switch is shorted and can be neglected. The equivalent circuit at this point is shown in Figure 9b. Since the value of $C_{bulk}$ is significantly larger than that of the parasitic capacitance, its influence can be neglected. The final equivalent impedance circuit for the switching device during the turn-on oscillation stage is presented in Figure 9c, where $L_P=L_{sd}+L_{loop}$ and $R_P=R_c+R_{ac}+R_{ds}$. During the turn-off oscillation stage, the upper switching device can be modeled as a series combination of $L_{sd}$ and $C_{oss}$, while the lower switching device can be modeled as a series combination of $L_{sd}$ and $R_{ds}$. The parameter $R_c$ also needs to be considered. During the turn-off oscillation period, high-frequency oscillation occurs in the loop of the upper switching device. In this case, $R_{ac}$ is in series with the upper switching device, and the RC Snubber for the lower switch is shorted and can be neglected. The equivalent circuit at this point is shown in Figure 10b. The final equivalent impedance circuit for the switching device during the turn-off oscillation stage is presented in Figure 10c, where the values of $L_P$ and $R_P$ are the same as those in the final turn-on model. By combining Figure 9 and Figure 10, it can be observed that both turn-on and turn-off oscillations share the same equivalent impedance model, with their impedance expressions obtained by (3).

After adding the RC Snubber, the impedance equation corresponding to the switching device is shown in (3). The oscillation behavior of the switching device is directly related to the impedance. The smaller the impedance peak, the more stable the oscillation performance. Based on this characteristic, when selecting the RC parameters, it is necessary to minimize the impedance as much as possible to effectively suppress oscillations.

$$Z_{snub}\left(j2\pi f\right)=\left({\displaystyle \frac{1}{j2\pi f{C}_{oss}}}\right)//\left[j2\pi f{L}_{sd}+\left(R_P+j2\pi f{L}_P\right)//\left(R_{snub}+{\displaystyle \frac{1}{j2\pi f{C}_{snub}}}\right)\right]$$

(3)

$C_{snub}$ significantly impacts the resonant characteristics and impedance frequency response of the circuit, as shown in Figure 11. The simulation results indicate that, when $C_{snub}$ is small, the Snubber exhibits weak resonance suppression, resulting in sharp resonance peaks. The system operates close to its natural resonant state with minimal distortion in frequency characteristics. When $C_{snub}$ is larger, the Snubber’s suppression capability is enhanced, leading to flattened peak shapes and improved stability. Significant changes in frequency characteristics occur, affecting the high-frequency response speed of the circuit. As the capacitance increases from 1 nF to 25 nF, the resonant frequency shifts toward lower frequencies and the low-frequency impedance peak decreases continuously. This demonstrates that $C_{snub}$ plays a significant role in suppressing resonance and adjusting frequency. However, the high-frequency impedance peak remains relatively unchanged, which is around 10 $\Omega$. When $C_{snub}=5$ nF, the low-frequency impedance peak is already lower than the high-frequency impedance peak. Further increasing $C_{snub}$ not only fails to reduce the highest overall impedance peak but also increases additional losses in the RC Snubber. Therefore, selecting $C_{snub}=5$ nF is sufficient to meet the design requirements for this circuit.

$R_{snub}$ also significantly impacts the circuit impedance. The variation of the resonance characteristic curve is shown in Figure 12. When $R_{snub}$ < 5 $\Omega$, the system is in an underdamped state. As the Snubber resistor value increases, both the high-frequency and low-frequency resonance peaks gradually attenuate. Essentially, the damping resistor suppresses the resonance intensity by dissipating energy as Joule heat, reducing the system’s overstress. As the Snubber resistor value increases, the bandwidth of the resonance peak widens significantly, and its frequency response range expands accordingly. When $R_{snub}$ is around 5 $\Omega$, the system reaches a critically damped state. Meanwhile, the resonance peak becomes flatter, and the frequency coverage range widens further. When $R_{snub}$ > 5 $\Omega$, the system enters an overdamped state, and no low-frequency or high-frequency impedance spikes appear. As $R_{snub}$ increases, the resonant frequency remains the same as without the Snubber, and the impedance peak value rises accordingly.

The selection of parameters for $R_{snub}$ and $C_{snub}$ should follow the principle of minimizing the impedance amplitude in order to effectively suppress switching oscillations. Under this premise, efforts should be made to minimize the values of $R_{snub}$ and $C_{snub}$ to reduce losses. The novel RC Snubber design proposed in this paper can significantly reduce the impedance amplitude, which exceeds 100 $\Omega$ without the RC Snubber. As shown in Figure 13, when $R_{snub}$ is set to 5 $\Omega$, the high-frequency impedance amplitude is suppressed to below 10 $\Omega$. It can be observed that, as $C_{snub}$ gradually increases, when $C_{snub}$ reaches 5 nF, the low-frequency impedance amplitude is reduced to less than 10 $\Omega$, thus satisfying the principle of minimizing impedance amplitude. Further increasing $C_{snub}$ results in only a limited reduction in low-frequency impedance amplitude, and the overall impedance amplitude does not decrease further. At this moment, the high-frequency impedance amplitude remains unchanged. To avoid an additional losses, the optimal RC parameter combination in this paper is $R_{snub}$ = 5 $\Omega$ and $C_{snub}$ = 5 nF, which ensures that the impedance across the entire frequency range remains below 10 $\Omega$.

4. Power Loop Inductance Optimization

4.1. Critical Influence Mechanisms of Loop Inductance

In the GaN HEMT-based half-bridge circuit, loop inductance, as a key parasitic parameter, has a decisive impact on the overall system performance. Optimizing loop inductance can enhance the switching performance, reduce electromagnetic interference (EMI), improve system stability, decrease energy loss, and improve load response.

Loop inductance and parasitic capacitance form a resonant circuit during switching transients. Because of a high $dv/dt$ and $di/dt$, sustained high-frequency oscillations are easily induced in the resonant circuit. These oscillations exhibit a strong correlation with loop inductance. Larger inductance values result in lower resonant frequencies and longer oscillation decay times. Additionally, an increase in parasitic inductance leads to a high voltage overshoot of the switching device.

The high-frequency oscillations can interfere with gate drive signals through coupling paths, causing abnormal fluctuations in the drive voltage. When the fluctuation amplitude approaches the turn-on threshold voltage of the device, the risk of false triggering increases significantly, potentially leading to faults such as bridge arm shoot-through. Moreover, EMI from high-frequency oscillations disrupts surrounding devices, deteriorating the system’s EMC performance. The presence of loop inductance also prolongs the switching times, directly increasing the switching losses. Under high-frequency operating conditions, frequent energy exchange between inductance and capacitance introduces additional reactive power losses, resulting in a significant reduction in system efficiency. Importantly, excessive loop inductance amplifies overshoot, subjecting devices to excessive current stress, accelerating device aging, and shortening their lifetime.

Without loop inductance optimization, the RC Snubber has limitations in improving the switching performance. When loop inductance exceeds a critical value, it not only diminishes the effectiveness of the RC Snubber in suppressing oscillations and mitigating overshoot, but also causes a surge in energy losses within the Snubber circuit, thereby negatively impacting the overall efficiency of the system.

4.2. Layout Optimization Design for Loop Inductance

To mitigate the issues caused by loop inductance, scientific layout design and structural optimization methods are employed to minimize parasitic inductance [2]. During the PCB layout phase, parasitic inductance is reduced by shortening the connection paths between different devices, minimizing the number of pathways, and avoiding narrow traces.

Experimental comparisons show that the layout in Figure 14a exists a high loop inductance due to excessively long grounding paths, large power loop areas and an asymmetric arrangement of the busbar. In contrast, the layout in Figure 14b significantly reduces loop inductance by optimizing the length of grounding wires, reducing the loop area, and achieving a symmetric configuration of the busbar, thus verifying the effectiveness of this layout strategy.

The calculation equation for $L_{loop}$ is expressed in (4) [7]. The resonant frequency before optimization is $f_{ring1}$ = 18.54 MHz. After optimization, it is $f_{ring2}$ = 20.84 MHz. It can be derived that $L_{loop}$ reaches 57.6 nH before optimization and is reduced to 45.3 nH after optimization. All the values of parasitic inductances ($L_{loop}$) used in this paper were obtained through Q3D simulation.

$$L_{loop}\approx {\displaystyle \frac{1}{4{\pi }^2{f_{ring}}^2{C}_{oss}}}$$

(4)

A symmetric power loop configuration enables the uniform distribution of current paths, effectively suppressing loop inductance. Furthermore, adopting a multi-layer board vertical layout architecture reduces loop parasitic inductance through the magnetic field cancellation effect. This approach has been validated in relevant literature [2]. GaN power devices exhibit significant heat generation during long-term operation. To mitigate the risk of thermal failure, heat dissipation structures must be integrated on the backside of the devices. Such bottom-side cooling schemes limit the applicability of the vertical architecture design. Therefore, to ensure efficient heat dissipation, a symmetric design of the power loop should be employed to minimize the adverse impact of loop inductance on system performance. Reasonable layout optimization can reduce loop inductance, bringing the RC Snubber closer to its ideal operating state and fully maximizing its damping suppression effect.

5. Experimental Verification

5.1. Test Setup

Among the various experimental methods for parameter acquisition, the Double-Pulse Test (DPT) is commonly used. This method is convenient and reliable, enabling the accurate capture of the circuit’s response during a single transient moment. However, due to its structural limitations, DPT cannot withstand the energy generated by continuous operation, making it incapable of capturing operational parameters during prolonged circuit operation. In practical applications, circuits often need to undergo long-term operation, requiring a shift to alternative testing methods. The Multi-Pulse Test (MPT) simulates the long-term operating state of devices through continuous pulses, allowing the acquisition of operational parameters such as overshoot and oscillation decay under sustained circuit operation. According to [34], MPT can obtain circuit operating parameters that approximate real working conditions, thus meeting the experimental requirements.

The MPT is applied for the analysis of the dynamic characteristics of switching devices, as shown in Figure 15. In this circuit, the upper switch ($S_H$) functions as the active switching device. The lower switch ($S_L$) acts as the passive switching device. $L_{load}$, as an inductive load, maintains current continuity during the turn-off phase of switching devices, serving as a freewheeling current source to provide an energy freewheeling path. The load resistor ($R_L$) directly affects the current amplitude and di/dt during switching transients. The value of $R_L$ must be selected within a reasonable range to ensure stable circuit operation.

Similarly to the DPT circuit, the current path exhibits variations across distinct operating phases. During the turn-on phase, the upper switch ($S_H$) is enabled, while the lower switch ($S_L$) remains off. The current path is depicted in Figure 16a. In this case, the power supply provides energy, and $L_{load}$ stores energy. During the freewheeling phase, $S_H$ is turned off, and $S_L$ is turned on. Since $L_{load}$ acts as an inductive load, it maintains current continuity. As shown in Figure 16b, $L_{load}$ provides the energy to form a freewheeling path. In this phase, $L_{load}$ releases the stored energy to sustain the load current, preventing abrupt voltage changes, and ensuring the stable operation of the circuit.

This paper employs the GaN chip from Innoscience (INNO040FQ015A, 40 V/50 A) as the power device in the half-bridge circuit. The driver uses the LM5113 half-bridge driver chip developed by Texas Instruments (TI), which provides two channels of control voltage signals. This chip can respond promptly to high-frequency signals, and its output driving voltage range meets the requirements for turning on and off the GaN HEMT. The control signals are provided by a control module with the TMS320F28335 (150 MHz) from the TI DSP chip as the main controller.

The experimental setup is shown in Figure 17. The PWM signal is generated by a signal generator and used to control the switches. The DC power supply provides energy for the power loop, while the auxiliary power supply is applied for driving GaN HEMTs. Waveforms of $V_{ds\_H}$, $V_{ds\_L}$, $V_{gs\_H}$, and $V_{gs\_L}$ are displayed in real-time via an oscilloscope. Power resistors and air-core inductors form the load circuit simulate the load characteristics during operation. In the PCB design, RC Snubbers must be placed in close proximity to the switching devices to minimize any negative impact of the Snubber circuit on the half-bridge system. The circuit parasitic parameters involved are shown in

Table 2. Values of the critical parameters in the experimental setup.

Parameter	Value
$V_{dc}$	12 V
$C_{bulk}$	1500 $\mathsf{\mu F}$
$C_{oss}$	1.28 nF
$L_P$ ($L_{sd}$ + $L_{loop}$)	60.3 nF
$L_{sd}$ ($L_{d\_H}$ + $L_{s\_H}$)/($L_{d\_L}$ + $L_{s\_L}$)	3 nF
$R_P$ ($R_c$+$R_{ac}$ + $R_{ds}$)	1.2 $\Omega$
$R_c$	1 $\Omega$
$R_{ac}$	0.1 $\Omega$
$R_{ds}$	1.5 $\mathrm{m\Omega }$

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5.2. Time-Domain Analysis of Experimental Results

In this paper, the Multi-Pulse Test (MPT) platform is used to conduct experimental analyses on the tested circuit board. The overall experimental diagram for the full cycle of the MPT experiment is shown in Figure 18 and the specific analytical waveforms for switching process analysis are presented in Figure 19, Figure 20, Figure 21, Figure 22, Figure 23, Figure 24 and Figure 25. When the current exceeds 15 A, switching oscillations become extremely severe, which can cause irreversible damage to the GaN HEMT. To ensure that the GaN HEMT operates under normal conditions, the half-bridge circuit must be equipped with the protection of RC Snubbers.

When only the RC Snubber is incorporated, the maximum voltage of the upper switch ($V_{maxH}$) decreases from 25 V to 20 V at a load current ($I_L$) of 5 A, representing a 20% reduction. Under the conditions of $I_L$ = 10 A, $V_{maxH}$ drops from 42 V to 31 V, achieving a 26% decrease. Additionally, the RC Snubber shortens the ringing time ($T_{ring}$) to approximately 10% of that in the original circuit. By damping resonance, the RC Snubber significantly suppressed both the voltage spikes and ringing time. However, even with the RC Snubber installed, circuits with a high $L_{loop}$ still exhibited noticeable overshoot, which was more pronounced at $I_L$ = 10 A.

When only $L_{loop}$ is reduced and $I_L$ = 5 A, $V_{maxH}$ decreased from 25 V to 24 V, a 4% reduction. When $I_L$ = 10 A, $V_{maxH}$ dropped from 42 V to 35 V, achieving a 16% decrease. In this scenario, $T_{ring}$ is reduced from 1.53 $\mathrm{\mu s}$ to 1.39 $\mathrm{\mu s}$, a 9% shortening, indicating that the effect is weaker than that of the RC Snubber. Although the layout optimization can reduce parasitic inductance, its standalone effect is limited, and it needs to be combined with the RC Snubber to achieve better performance.

When the RC Snubber and PCB layout optimization are simultaneously applied, under the conditions of $I_L$ = 5 A, $V_{maxH}$ is reduced from 25 V to 14 V, achieving a 44% reduction, while the ringing time of the upper switch ($T_{ringH}$) is shortened from 1.4 $\mathrm{\mu s}$ to 0.13 $\mathrm{\mu s}$, a 91% decrease. When $I_L$ = 10 A, $V_{maxH}$ is reduced from 42 V to 25 V (40% reduction), and the maximum voltage of the lower switch ($V_{maxL}$) is reduced from 24 V to 17 V (29% reduction), both of which are below the maximum voltage of the device (40 V). Under these cases, $T_{ringH}$ is shortened from 1.53 $\mathrm{\mu s}$ to 0.09 $\mathrm{\mu s}$, and the ringing time of the lower switch ($T_{ringL}$) is reduced from 1.26 $\mathrm{\mu s}$ to 0.07 $\mathrm{\mu s}$, with both achieving a 94% reduction. When $I_L$ = 15 A, reducing $L_{loop}$ from 57.6 nH to 45.3 nH results in the mean of $V_{maxH}$ decreasing from 35 V to 25 V, and $T_{ringH}$ dropping sharply from 0.12 $\mathrm{\mu s}$ to 0.08 $\mathrm{\mu s}$. Similarly, $V_{maxL}$ decreases from 24 V to 21 V, and $T_{ringL}$ shortens from 0.11 $\mathrm{\mu s}$ to 0.07 $\mathrm{\mu s}$. When $I_L$ increases to 20 A, with $L_{loop}$ = 45.3 nH and the RC Snubber, $V_{maxH}$ rises to 46 V, approaching the tolerance limit of the switch. However, $V_{maxL}$ further decreases to 16 V, and the average values of $T_{ringH}$ and $T_{ringL}$ can be maintained at extremely low levels of 0.07 $\mathrm{\mu s}$ and 0.06 $\mathrm{\mu s}$, respectively. The synergistic scheme of the layout optimization and RC Snubber ensures that the overshoot of the GaN HEMT remains below the voltage withstand threshold, providing a sufficient voltage safety margin. The reduction in parasitic inductance brings the RC Snubber closer to its ideal operating condition.

$V_{gs}$ is significantly affected by the coupling of $V_{ds}$ oscillations. Without any suppression measures, severe noise and fluctuations in $V_{gs}$ lead to extended switching times and the impaired integrity of the drive signal. When $I_L$ = 5 A, obvious oscillations of $V_{gs}$ are observed, posing a risk of false activation, which becomes more intense when $I_L$ = 10 A. When only the RC Snubber is activated are the noise and fluctuations in $V_{gs}$ alleviated. The interference noise of $V_{gs}$ caused by oscillations is reduced, and the risk of false triggering is mitigated. In contrast, PCB layout optimization alone yields limited improvement in $V_{gs}$ disturbance. When employing the PCB layout optimization alone, the effectiveness of noise suppression remains limited. Although shortening parasitic coupling paths can marginally reduce the amplitude of $V_{gs}$ fluctuations, improvements in $V_{gs}$ disturbances are constrained, with oscillations persisting near the threshold level. When the RC Snubber and PCB layout optimization are simultaneously applied, the stability of $V_{gs}$ is significantly enhanced. When $I_L$ = 10 A, $V_{gs}$ fluctuations are within an acceptable range. When $I_L$ = 5 A, $V_{gs}$ stabilizes near zero with notably reduced noise, completely eliminating the hidden danger of false triggering. Detailed experimental data are presented in

Table 3. Parameters of switching waveform in GaN HEMT half-bridge circuit.

Parameter	Value
${\mathit{I}}_{\mathit{L}}$/A	5				10				15		20
${\mathit{L}}_{\mathit{loop}}$/nH	57.6		45.3		57.6		45.3		57.6	45.3	45.3
Snubber	wo	w	wo	w	wo	w	wo	w	w	w	w
Mean of $V_{maxH}$/V	25	20	24	14	42	31	35	25	43	36	46
Mean of $V_{maxL}$/V	24	21	23	20	23	17	24	21	20	18	16
Mean of $T_{ringH}$/$\mathrm{\mu s}$	1.40	0.14	1.29	0.13	1.53	0.14	1.39	0.09	0.12	0.08	0.07
Mean of $T_{ringL}$/$\mathrm{\mu s}$	1.41	0.19	1.36	0.14	1.26	0.12	1.02	0.08	0.11	0.07	0.06

“wo” denotes the circuit without RC Snubbers and “w” denotes circuit with RC Snubbers.

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5.3. Spectrum Analysis of Experimental Results

By utilizing frequency-domain analysis, the suppression effect of the RC Snubber and layout optimization on oscillations in GaN HEMT can be more intuitively quantified, as illustrated in Figure 26, Figure 27, Figure 28 and Figure 29. The spectrum analysis reveals that the RC Snubber exhibits a significant suppression effect on high-frequency noise, while PCB layout optimization further enhances this effect. Without the RC Snubber, the noise fluctuates drastically in specific frequency bands, with an spike of up to 30 $\mathrm{dB\mu V}$, accompanied by voltage spikes exceeding the system tolerance threshold. After installing the RC Snubber, the amplitude of noise is suppressed by 30 dB. Therefore, the EMC performance of the circuit is substantially improved.

In the frequency band around 20 MHz, when $I_L$ = 5 A, the amplitude of noise is reduced from 99 $\mathrm{dB\mu V}$ to 75 $\mathrm{dB\mu V}$, representing a 24.2% decrease, as shown in Figure 26. At a load current of 10 A, the amplitude of noise drops from 101 $\mathrm{dB\mu V}$ (without the Snubber) to 75 $\mathrm{dB\mu V}$ when the layout optimization and RC Snubber act synergistically, achieving a 25.7% reduction, as illustrated in Figure 27. The main difference between the two scenarios lies in a slight reduction in the noise floor under low load current conditions, while the effectiveness of layout optimization in suppressing high-frequency noise remains stable. This is consistent with the characteristic that high-frequency components are primarily dominated by parasitic parameters. Figure 28 demonstrates that when the load current is 15 A, reducing the loop inductance from 57.6 nH to 45.3 nH decreases the amplitude of noise from approximately 93 $\mathrm{dB\mu V}$ to below 90 $\mathrm{dB\mu V}$. Figure 29 shows that, if the loop inductance is fixed at 45.3 nH, an increase in the load current to 20 A causes the overall harmonic amplitude to rise significantly, with the rise being particularly significant in the medium and high-frequency ranges. However, the amplitude of noise can still be controlled around 90 $\mathrm{dB\mu V}$ through the optimization of the RC Snubber and loop inductance.

From the perspective of spectral features, without the RC Snubber circuit, the amplitude of noise above 20 MHz reaches 101 $\mathrm{dB\mu V}$, with sharp resonant spikes. These high-frequency components are prone to interfering with surrounding circuits through coupling paths, increasing the risk of circuit failure. In contrast, with the synergistic effect of layout optimization and the RC Snubber, the amplitude of noise at 20 MHz is reduced to 75 $\mathrm{dB\mu V}$ and the resonant spikes become flattened, achieving a noise reduction of 25.7%. The mechanism lies in the RC damping effect, which weakens the resonant intensity. Additionally, layout optimization shortens the current paths, mitigating the resonant effects induced by parasitic parameters. The relevant mechanisms can be validated by (3) and (4).

5.4. Influence of $C_{snub}$ and $R_{snub}$

Figure 30 and Figure 31 illustrate the effects of different $C_{snub}$ and $R_{snub}$ parameter values on switching characteristics. The impedance characteristics revealed in Figure 11 and Figure 12 provide theoretical support for analyzing the experimental phenomena in Figure 30 and Figure 31, collectively establishing a correlation mechanism between RC Snubber parameters and switching characteristics.

Figure 30 shows that varying values of $C_{snub}$ significantly affect the waveform of $V_{ds\_H}$. When $C_{snub}$ = 1 nF, the voltage peak reaches the maximum with a large oscillation amplitude. This is due to the limited energy storage capacity of the capacitor with a small value, which results in weak buffering against switching transients. Consequently, it fails to suppress the intrinsic resonance of the LC loop, leading to intense oscillations with slow attenuation and an overshoot of 33.2 V. As $C_{snub}$ increases, the oscillation amplitude of the drain–source voltage gradually decreases, and the convergence time is progressively shortened. When $C_{snub}$ = 5 nF, the oscillation suppression effect meets engineering requirements, with the voltage overshoot reduced to 26.8 V. Meanwhile, the convergence speed significantly improved. This aligns with the optimized impedance characteristics in Figure 30, where the resonant frequency shifts to the low-frequency range and the impedance peak is notably reduced. When $C_{snub}$ = 25 nF, the voltage peak is the lowest, with the smallest oscillation frequency and amplitude. Due to the increased energy storage in the capacitor of the RC Snubber, the mitigation effect on high-frequency oscillations in the drain–source voltage is enhanced, resulting in the effective suppression of the voltage peak. In Figure 11, as the capacitance value of the Snubber capacitor increases, high-frequency oscillations are effectively suppressed. It should be noted that an appropriate capacitance value must be selected to balance power losses and avoid introducing additional parasitic parameters. Additionally, an excessively large $C_{snub}$ may increase the parasitic inductance of $C_{snub}$ and power losses. Thus, a 5 nF capacitor for $C_{snub}$ is determined as the optimal choice, with detailed data presented in

Table 4. Impacts of $C_{snub}$ on voltage oscillation and overshoot.

Parameter	Value
$C_{snub}$/nF	1	2.5	5	10	25
Mean of overshoot/V	33.2	28.4	26.8	24.2	22.6
Mean of $f_{ring}$/MHz	21.3	19.1	17.7	15.8	15.2

$f_{ring}$ denotes ringing frequency.

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Figure 31 demonstrates that $R_{snub}$ has a critical influence on the damping characteristics and energy conversion process of the circuit, and its operational mechanism can be systematically explained by combining the impedance characteristic laws revealed in Figure 12. When $R_{snub}$ < 5 $\Omega$, the circuit operates in an underdamped state. As the resistance increases, the oscillation amplitude gradually decreases. This aligns with the characteristic of sharp resonant peaks in Figure 31, where slow oscillation attenuation is attributed to insufficient energy dissipation. When $R_{snub}$ = 5 $\Omega$, the circuit reaches a critically damped state, achieving the optimal oscillation convergence effect. The overshoot is reduced to 26.8 V. This corresponds to the optimal damping matching of impedance characteristics in Figure 31, fully complying with the critical damping design criteria. When $R_{snub}$ > 5 $\Omega$, the circuit enters an overdamped state. As the resistance increases, the output overshoot continues to rise (reaching 32.6 V), with higher voltage spikes, more significant oscillations, and slow attenuation. This is directly associated with the characteristic in Figure 31, where the excessive broadening of the resonant peak leads to the failure of impedance regulation. Experimental results confirm that $R_{snub}$ should not be excessively large. Otherwise, it will not only cause voltage spikes to damage the switching transistor due to excessive overshoot but also weaken the convergence effect of the Snubber circuit. This fully reflects the trade-off mechanism of resistance values between energy dissipation and resonance suppression, with detailed data presented in

Table 5. Impacts of $R_{snub}$ on voltage oscillation and overshoot.

Parameter	Value
$R_{snub}$/$\Omega$	1	2.5	5	10	25
Mean of overshoot/V	19.8	22.8	26.8	29	32.6
Mean of $f_{ring}$/MHz	10.1	10.3	17.4	19.2	19.4

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The impedance characteristics depicted in Figure 11 and Figure 12 provide a quantitative explanation for the experimental phenomena observed in Figure 30 and Figure 31. The Snubber capacitor mitigates the overshoot by regulating the resonant frequency and peak impedance, while the Snubber resistor controls the oscillation decay rate by adjusting the damping ratio. The optimal parameter combination derived from experimental optimization ($R_{snub}$ = 5 $\Omega$, $C_{snub}$ = 5 nF) precisely corresponds to the optimal matching point of impedance characteristics in Figure 11 and Figure 12. At this point, the circuit not only suppresses resonance via the capacitor but also achieves damping balance through the resistor, ultimately resulting in a 40% reduction in voltage overshoot and reducing the number of ringings to 1.

6. Conclusions

This paper focuses on the suppression of switching oscillations in GaN HEMTs using RC Snubbers. Through the theoretical investigation of this paper, the following conclusions are derived:

Through multiple sets of comparative experiments, the synergistic oscillation suppression strategy combining RC Snubbers and loop inductance optimization has been quantitatively validated for its effectiveness in suppressing oscillations for GaN HEMTs. This strategy achieves a balance between high efficiency and reliability. Under $I_L$ = 10 A, the maximum overshoot reaches 42 V without any mitigation measures. Employing only the RC Snubber reduces the overshoot by 26%, while solely optimizing the layout results in a 16% reduction. In contrast, the combination of RC Snubber and layout optimization lowers the overshoot to below 25 V, achieving a 40% reduction, significantly outperforming either strategy individually.

With the synergistic strategy of RC Snubber and layout optimization, the gate drive voltage exhibits no oscillations, completely eliminating the risk of false triggering. The ringing time is reduced below 0.1 $\mathrm{\mu s}$ (a 95% reduction from the original). In the high-frequency band above 20 MHz, the noise is reduced from 101 $\mathrm{dB\mu V}$ to 75 $\mathrm{dB\mu V}$ (a 25.7% decrease), significantly improving the electromagnetic environment. Additionally, a high-frequency equivalent model reveals the coupling relationship between $R_{snub}$, $C_{snub}$, and $L_{loop}$, preventing the impedance mismatch between the Snubber circuit and parasitic parameters. The parameter combination of $R_{snub}$ = 5 $\Omega$ and $C_{snub}$ = 5 nF achieves critical damping while maintaining lower power losses and a higher oscillation suppression performance.

In summary, the RC Snubber studied in this paper enhances the tolerance of GaN HEMTs under high load conditions, suppresses voltage oscillations and overshoot in the power loop, and prevents high-frequency oscillations from interfering with the drive loop. Meanwhile, loop inductance exacerbates high-frequency oscillations. Optimizing the layout to reduce loop inductance not only suppresses high-frequency oscillations and enhances device tolerance, but also improves the Snubber performance. By combining the RC Snubber and layout optimization, it can achieve the maximum suppression effectiveness of oscillations in GaN HEMTs. This collaborative strategy can significantly improve circuit stability and reliability, offering a quantifiable approach for designing high-power-density power systems.

Future work will focus on adopting novel top-side cooling packaging to investigate the impact of vertical layout on the performance of GaN HEMT-based circuits. By converting the current path from horizontal to vertical, the vertical layout can significantly shorten the physical length of the loop, which is a key principle for optimizing circuit performance. Additionally, combined with a magnetic field cancellation design, the layout can further reduce parasitic inductance. Then, a dual-improvement mechanism with path optimization and parasitic suppression should be established. Ultimately, the above design is expected to significantly reduce the switching oscillation and voltage overshoot, reaching a comprehensive enhancement in the performance of GaN HEMTs.