Low Profile Sinuous Slot Antenna for UWB Sensor Networks

Abstract

This article describes the design and implementation of a low-profile sinuous slot antenna, intended for ultra-wideband (UWB) sensor networks, which can be produced on one conductive layer. The article explains the design and optimization of the sinuous slot antenna and its modifications, including its sinusoidal curve shape. Other modifications were aimed at optimizing the antenna feeding. Desirable properties of the designed and implemented antenna modifications were verified both by simulation and empirically. Experimental measurements of the antenna’s properties were carried out using a vector network analyzer in an anechoic chamber and also by a pulsed UWB radar in the frequency range from 0.1 to 6 GHz. The low-profile antennas were implemented on a Rogers RO3206 substrate.

1. Introduction

At present, ultra-wideband (UWB) radar systems appear to be an appropriate tool in many applications [1,2]. They are capable of non-invasive motion detection, while their broadband properties provide them with sufficient precision for their application as contactless heart sensing or breathing sensors [3]. UWB radars can also be used for many other applications, such as medicine, civil engineering, impedance spectroscopy [4], ground penetrating radar (GPR) applications [5], through-wall security applications [6], and many others [7]. Additionally, UWB technology is currently widely used in sensor networks, providing high robustness to interference as well as a low complexity of transmitters and receivers while decreasing the energy consumption. The IEEE 802.15.4a standard defines UWB-based sensor networks as having a high degree of flexibility, including their modulation, coding, and multiple access schemes. Antennas and their properties are the key elements in the field of sensors [8]. In some applications, in addition to their large bandwidth and electrical characteristics, their dimensions are also critical. Examples of such applications are wearable devices [9] or energy harvesting [10,11].

This article presents the design of a UWB antenna for through-wall measurements in security applications. The requirements for such an antenna include minimum ringing and a bandwidth greater than 500 MHz, or 0.2 times the central frequency in a relative form [12]. Other requirements include linear polarization, the width of the main lobes being more than 90˚, a low-profile structure, easy implementation combined with low production costs, and minimum dimensions for the use of frequencies lower than 0.8 GHz.

The shape of the proposed antenna is based on sinuous curves drawn in the polar coordinate system. The curve of this antenna was originally designed in the 1980s by R. H. Duhamel [13]. Its main advantage is that it is an analytical curve and it allows the creation of logarithmically periodic structures and shapes. Just the log-periodic structure of the antenna allows it to achieve a large frequency bandwidth in a compact and planar shape.

A number of recent publications describe similar UWB antennas [14,15,16,17,18,19], but these solutions do not work at such low frequencies and do not achieve such a large relative bandwidth and compact dimensions as our solution presented below.

This form of antenna has been known for several decades, but in a non-planar form [20,21]. The main purpose of the new design was to modify the UWB sinuous antenna for radar systems with linear polarization. A large relative width of the frequency band and frequencies below 0.8 GHz allow its electromagnetic waves better penetration and transmission through walls. The production of the antenna itself and its excitation in planar form at a low production cost, using inexpensive technological processes and dielectric materials, favor this antenna in the construction of light and portable systems.

The antenna was implemented in a slotted form. This means that the antenna is etched or milled into the conductive layer of a printed circuit board (PCB) substrate. In the same layer in which the antenna is produced, the excitation of the antenna is included as a coplanar waveguide (CPW). It should be noted that the resulting slot antennas have a polarization vector oriented perpendicular to the axis of the main arms of the slot.

2. Design of the Antenna Geometry

2.1. Basic Shape

The sinuous shape introduced by Duhamel [13] provides a logarithmically periodic structure of an antenna pattern that is theoretically frequency-independent. Thanks to this feature, the sinuous form (analytical curve) is suitable for tuning and producing an antenna suitable for UWB radar systems.

In general, the antenna consists of N arms consisting of P incrementally growing cells. The size of the p-th cell grows (scaling) from the first cell (inner cell—the smallest and closest to the center) to the last cell (outer cell—the largest and closest to the outer edge). The symbols r_p and r_p+1 denote the inner and outer radii of the p-th cell. The sinusoidal curve (in the polar coordinate system) that is the basis for the shape of the antenna can be described by the following equation [13,19]:

$$\begin{gathered} {\Phi }_p\left(r\right)={\left(-1\right)}^{p-1}{\alpha }_p\sin \left(\pi \frac{ln\left(\frac{r}{r_p}\right)}{ln\left({\tau }_p\right)}\right), \\ {r}_p\le r\le r_{p+1} \end{gathered}$$

(1)

where α_p and τ_p are the key design parameters. Parameter α_p is a positive number that defines the maximum value of Φ in a cell via the index p, in which half of the sinusoidal wave curve in the polar system is present. Parameter τ_p defines the so-called cell growth. Such a sinusoidal curve fulfills the logarithmic periodicity condition, thereby achieving the theoretical frequency independence of the antenna’s bandwidth. Parameter τ_p is defined as the ratio of the outer and inner radii [13]:

$${\tau }_p=r_{p+1}/r_p$$

(2)

Parameters α_p and τ_p are usually unchanged for the entire antenna.

The slot contour curve of one antenna arm is then formed from two sinuous curves by rotating them with a defined angle (±δ). In Figure 1, it is possible to see a basic curve with one cell highlighted (bold curve).

The total sinusoidal curve is created by joining individual cells (gray curve in Figure 1). By adding additional curves and by their mutual rotation by the angles of +δ and −δ, an arm of the antenna is created. It is terminated at both ends by the arc (dashed curves). The radius of the inner circle is r₁ and the length of the arc is 2δ. The radius of the outer circle is r₁τ ^P and the angular length of the arc is 2δ as well. The second arm of the antenna arises by simply rotating the first arm by an angle of π rad.

For localizing moving objects behind a barrier, an electromagnetic field with linear polarization is often used [6,12,22,23,24]. For this reason, the resulting antenna consists of only two arms (N = 2), as shown in Figure 2. When N is greater than 2, the sinuous antenna is able to provide two patterns with orthogonal linear polarizations [13,19,25,26].

The analytical equation of the curve, Equation (1) above, was transformed from a polar to a Cartesian form to support its use in the CST MicroWave Studio (CST MWS) simulation environment. The resulting curve equation can thus be written as:

$$X_p\left(t\right)=t\cos \left({\Phi }_p\left(t\right)\right),$$

(3)

$$Y_p\left(t\right)=t\sin \left({\Phi }_p\left(t\right)\right)$$

(4)

where t is in the range <r₁τ ^p-1, r₁τ ^p > in each cell, with an index of p, α is the cell’s angular width, r₁ is the smallest antenna radius, and τ is the cell growth index.

2.2. Antenna Ripple

The main idea of modifying the antenna by forming ripples on its cells is to increase the antenna’s bandwidth towards lower frequencies while maintaining the same outer dimensions [26]. The antenna cells’ ripples cause extensions of surface current pathways. Figure 3 compares the contour curves of an antenna cell with and without a ripple.

To achieve the ripple of the antenna curves, it is necessary to modify Equations (3) and (4) into the following forms:

$$X\left(t\right)=t+A_1\frac{t}{r_1{\tau }^{p-1}}\cos \left(pv\frac{\raisebox{1ex}{$\pi $}\!\left/ \!\raisebox{-1ex}{$2$}\right.}{\alpha }{\Phi }_p\left(t\right)\pm \delta \right)\cos \left({\Phi }_p\left(t\right)\pm \delta \right),$$

(5)

$$Y\left(t\right)=t+A_1\frac{t}{r_1{\tau }^{p-1}}\cos \left(pv\frac{\raisebox{1ex}{$\pi $}\!\left/ \!\raisebox{-1ex}{$2$}\right.}{\alpha }{\Phi }_p\left(t\right)\pm \delta \right)\sin \left({\Phi }_p\left(t\right)\pm \delta \right)$$

(6)

where A₁ represents the relative ripple amplitude of the curves relative to the radius r in cell p. Another parameter, pv, represents the number of waves in one cell. In the antenna tuning process, several antenna modifications were created, differing in the pv parameter, which ranged from 0 to 50. The value of pv = 0 means that the curve in the cell is not curled but smooth (basic shape). Examples of the effect of the choice of parameter values α, δ, A₁, and pv on the resulting shape of the contour curve are illustrated in Figure 3.

2.3. Antenna Feeding

Structures based on a micro-strip terminated with a UWB “fan” open stub [23] are generally used for feeding low-profile slot antennas. In order to minimize the reflection due to a mismatch between the feeding line and a sinuous slot, the radius and angular dimensions of the “fan” and the width of the strip must be optimized. For a lower cut-off frequency of the antenna, the radius of the “fan” increases. On the other hand, radius r₁ defines the upper frequency limit of the antenna. For frequencies that are well below 1 GHz, the “fan” radius is greater than the radius r₁, which determines the upper frequency limit and thus also the maximum bandwidth. Therefore, a CPW was used for the excitation of the proposed antenna. The CPW is formed on the same conductive layer as the sinuous slot antenna. In the case of the CPW, the signal wire is separated from the ground by two gaps and terminated by a connection to the ground plane. The internal conductor of the CPW has a trapezoidal form. Such transmission lines behave as impedance transformers. The resulting configuration is shown in Figure 4.

The point of connection to the connector is the trapezoid’s widest point and its connection with the slot width, m, is the narrowest. The widest dimension of the inner conductor is denoted by CPWfeed50 and the narrowest by CPWfeed100. The outer edges of the conductive surface have the same distance along the entire excitation length. This changes the gap between the inner and outer conductors. These gaps are similarly labeled as CPWgap50 and CPWgap100.

The second excitation modification was the use of a coplanar waveguide with ground (CPWG) feeding. In this form of excitation, the grounding of the CPWG conductor on the other side of the substrate was connected to the base surface of the slot antenna by via holes (Vias). The width of the grounding strip on the other side of the dielectric and its length are denoted by wkp and lkp, respectively. The horizontal and vertical spacings are marked as vzpx and vzpy, respectively. The radius of the through-holes is denoted by rv. Figure 4 shows the final antenna model, including both the CPWG and feeding parameters.

3. Results

To support the experiments, antennas were made in both of the forms of excitation. During the simulations, many combinations of parameters (described above) were tuned, leading to a great deal of data. Of these, the most favorable ones were chosen, in terms of the bandwidth and minimum frequency which achieves a satisfactory parameter value, s₁₁. Figure 5 illustrates the effect of the values of the parameters α and δ as an example. From the simulation process in the frequency range of 0.1–6 GHz, it was found that the antenna design produced the best results with the selected parameters of δ ≈ 38˚ and α ≈ 26˚. The resulting parameters of the manufactured CPW antennas can be seen in

Table 1. The resulting parameters of the manufactured antenna with coplanar waveguide (CPW) feeding.

Parameter	Value	Parameter	Value	Parameter	Value
W	170 mm	H	160 mm	CPWfeed50	2 mm
α	26.4˚	δ	38.3˚	CPWgap50	0.4 mm
τ	1.151	r1	10 mm	CPWfeed100	0.6 mm
m	4.8 mm	pv	22.1	CPWgap100	1.1 mm
A1	1.2

.

Both variants of the slotted antennas with the coplanar excitation (CPW and CPWG) were made using a subtraction method on a Rogers RO3206 substrate. Verification of the simulated characteristics of the proposed antennas was performed through standard measurement, using a vector network analyzer in an anechoic chamber (with dimensions of 5.3 m × 3.8 m × 3.1 m, including the RF absorber cones and shielding walls), but also using the impulse UWB radar [27]. The comparison of the measured and simulated results of the parameter s₁₁ (which represents how much power is reflected from the antenna, and hence is known as the reflection coefficient or return loss) and radiation characteristics can be seen in Figure 6, Figure 7, Figure 8, Figure 9 and Figure 10.

The radiating diagrams of the antenna for the frequencies of 1–4 GHz were also simulated and measured by two methods and in two perpendicular planes, as shown in Figure 7.

Table 2. The resulting parameters of the manufactured antenna with coplanar waveguide with ground (CPWG) feeding.

Parameter	Value	Parameter	Value	Parameter	Value
W	170 mm	H	160 mm	wkp	4 mm
α	26.5˚	δ	38.3˚	lkp	77.65 mm
τ	1.152	r1	10 mm	vzpx	3.5 mm
m	4.7 mm	pv	20	vzpy	3 mm
A1	1.2	CPWfeed50	1.3 mm	CPWfeed100	0.4 mm
rv	0.2 mm	CPWgap50	0.55 mm	CPWgap100	1.0 mm

shows the dimensions and other parameters of the second version of the implemented antenna with the CPWG feeding.

Both sides of the implemented antenna design can be seen in Figure 8.

Figure 9 compares the simulated and measured s₁₁ parameters of the second antenna version (with CPWG feeding).

The measured antenna radiation patterns in the range from 1 GHz to 4 GHz, with a step of 1 GHz, are shown in Figure 10.

In our experiments, we measured the impulse response for our antennas and compared it to the available commercial antennas. The commercial antennas were QRH400 (Quad Ridged Horn Antenna) for the 0.4–6 GHz frequency band and DRH10 (Double Ridged Waveguide Horn Antenna) for the 0.74–10.5 GHz band, supplied by RFspin. Figure 11 shows the measured results for the CPWG antenna. The measurements were performed using an 8-Channel Digital Sampling Converter SD-10806 in the 0.1–6 GHz frequency range, supplied by Geozondas Ltd.

4. Discussion and Conclusions

This article presented the design of a UWB sinuous slot antenna intended for UWB radar applications. Several variants of the antenna were simulated, and two versions achieved sufficiently promising results to be produced as working prototypes.

By combining the coplanar waveguide and the sinuous slot antenna, a new structure was formed as an analytical curve. Choosing the parameters which defined the shape involved a wide set of tools to optimize and tune it.

The manufactured versions of the antennas differ in the way they are powered. The antennas were made in both grounded and ungrounded coplanar waveguides.

The change of the ripple amplitude was very significant in changing the s₁₁ parameter (the reflection coefficient or return loss). On the one hand, in some cells, the number of ripples had a significant effect on the s₁₁ result, while in other cells, this parameter had no effect. Therefore, in the embodiment shown in Figure 8, some cells are curled and others appear without a ripple.

From the simulated results for the antenna with CPW feeding (the first version), it can be seen that the frequency range of 0.53–4.86 GHz corresponds to an s₁₁ value below –10 dB. Thus, it can be said that the absolute bandwidth is 4.33 GHz or the relative bandwidth is 9.1:1.

The values of parameter s₁₁ obtained by measurements using a circuit analyzer showed that the working frequency band of the antenna at the –10 dB s₁₁ value started at 0.459 GHz and ended at 4.5 GHz, which means the bandwidth is 4.041 GHz or, in relative terms, it is up to 9.8:1. Measurements using the UWB radar generally confirm the measured results.

The radiating patterns show that the main lobe width is about 70° and the antenna gain is within the range of 2.5–6 dBi for the listed range of frequencies.

The measured results of the second version of the antenna with CPWG excitation were also compared with the simulation results. Simulated s₁₁ values showed that the antenna is suitable for the 0.53–4.5 GHz frequency band, and so the absolute bandwidth is 3.97 GHz or the relative one is 8.49:1. However, the measured s₁₁ parameter determined for the antenna was in the band range of 0.459–4.29 GHz, resulting in a bandwidth of 3.831 GHz or 9.3:1. The antenna gain ranged from 2.9 to 5.7 dBi. From the antenna radiation diagrams, the width of the main lobe can be determined to be in the vicinity of 70°. However, modification of the antenna (with CPWG feeding) contributed to improving the radiation pattern’s symmetry.

The measured results in both cases are better than the simulated ones. Both the resulting measured characteristics and experiments showed that these antennas are well-suited for a wide range of UWB radar applications.

From the perspective of further development, it may be said that the monolayer structure of the antenna could be suitable for development on dielectric foils or fabrics. For textile fabrics, it would be necessary to choose a technological process that would ensure a sufficient accuracy of geometric shapes because the shape has a great effect on the antenna’s impedance. In the case of dielectric films, we have a Jet Printing process in the preparation stage. Both approaches are particularly suited, for example, for wearable devices that could be used as patient monitors (heart rate, breathing) or for car drivers. In the case of drivers, such systems could be built into the seats.