Design of Shared-Aperture Base Station Antenna with a Conformal Radiation Pattern

Abstract

Aiming at solving the problem of radiation pattern distortion caused by coupling between antennas in different frequency bands in traditional shared-aperture base station array antennas, a new shared-aperture array antenna integrating high-frequency filtering units and medium-frequency electromagnetic transparent antenna units is proposed. Without adding additional decoupling structures, it is possible to effectively reduce the coupling of different frequencies, while weakening common-mode and scattering interferences, making the radiation pattern conformal. The array consists of an electromagnetic transparent antenna unit in the medium-frequency (1.71–2.70 GHz) band and four filtering antenna units in the high-frequency (3.30–3.70 GHz) band. The four high-frequency antenna units form two 2 × 1 linear arrays arranged on both sides of the medium-frequency antenna unit and share a reflector. The simulation and measurement results show that the voltage standing wave ratio (VSWR) in the working frequency band is less than 1.50, the average gain in the medium-frequency band is 8.80 dBi, the average gain in the high-frequency band is 12.20 dBi, and the radiation pattern is normal. It is suitable for the field of shared-aperture base station antennas.

1. Introduction

The development of mobile communication technology has entered the fifth generation, but the old communication standards will not be withdrawn from the network any time soon. It is foreseeable that the coexistence of 2G, 3G, 4G, and 5G will last for a long time [1,2]. However, to reduce the space occupied by base station antenna equipment and alleviate the shortage of base station site resources, operators hope to integrate 5G antenna arrays with 2G/3G/4G. Therefore, shared-aperture base station antennas have been increasingly widely used as a technical means. The shared-aperture base station antenna [3,4,5,6,7,8,9,10] places antenna arrays of multiple frequency bands in a radome with antenna units that share a reflector. To reduce the size of the antenna and reduce operating costs, antennas of different frequency bands are often placed very closely, resulting in strong coupling [11,12,13,14,15,16,17] and distortion of the radiation pattern. As shown in Figure 1, the size of the low-frequency band antenna is often large and confined in a limited space, which can inevitably block the high-frequency band antenna. When the high-frequency antenna is working, the low-frequency antenna will cause scattering interference to the high-frequency antenna, causing the high-frequency antenna radiation pattern to be distorted. When the low-frequency antenna works, the radiation surface of the high-frequency antenna and the coaxial cable will couple to the low-frequency current, called the common-mode current. The radiation field generated by the low-frequency current coupled to the high-frequency antenna causes common-mode interference to the low-frequency antenna, causing the low-frequency antenna radiation pattern to deform.

Domestic and foreign scholars have conducted extensive research on different frequency decoupling methods. Common methods include loading decoupling networks [18,19,20,21,22], loading stealth cloaks [23,24,25], loading frequency selective surfaces [26,27,28,29,30,31,32,33,34,35], and etching gaps [36,37,38,39,40] to reduce the coupling of different frequencies. In [18], a T-type decoupling network is proposed, which is expanded into a three-element decoupling network by combining it with a thin transmission line and further generalized into a multi-element linear decoupling network. The antenna impedance matching is significantly improved. In [22], an antenna decoupling network is proposed that uses two unequal Wilkinson power dividers, significantly improving antenna isolation. Although decoupling networks can reduce coupling, the presence of the decoupling network will reduce the radiation efficiency of the antenna. Moreover, they can only reduce the mutual coupling between the antenna feed ports, but cannot reduce the coupling between antenna radiators in different frequency bands. In [23], an elliptical stealth cloak is proposed that is put on the low-frequency antenna dipole arm. The induced current generated on the stealth cloak is opposite to the current on the antenna dipole arm, suppressing the scattering interference of the low-frequency antenna on the high-frequency antenna, thereby achieving a stealth effect. However, the cloak has a limited scope of application and can only be used in vertical dipole or monopole antennas. In [29], an antenna array based on FSS stacking is proposed, where the high-frequency antenna is stacked on top of the low-frequency antenna. The FSS serves as the floor of the high-frequency antenna, showing reflection characteristics for the high-frequency antenna, and as the phase-inducing structure of the low-frequency antenna, showing electromagnetic transparent characteristics for the low-frequency antenna. Due to the stacking structure, the height of the entire antenna will be increased. In [34], a new type of non-uniform metasurface is proposed and applied to the low-frequency antenna unit, which can restore the radiation pattern of the high-frequency unit. However, the influence of the high-frequency unit on the low-frequency unit is not considered, resulting in a lower gain of the low-frequency unit. In [38], a shared-aperture base station antenna with a U-shaped slot is proposed, which operates in two frequency bands: 1.70–2.20 GHz and 3.30–3.60 GHz. Etching a U-shaped slot on the low-frequency unit reduces the influence of the low-frequency unit on the high-frequency unit. However, the proposed low-frequency antenna has a narrow bandwidth and does not consider the effect of common-mode current on the low-frequency antenna. In [40], a low-frequency electromagnetically transparent antenna is proposed. By etching grooves on the dipole, the electromagnetically transparent characteristic is achieved to reduce coupling. However, the antenna bandwidth is very narrow and the radiation surface is very large, which is not conducive to the integration of shared-aperture array antennas.

In summary, the decoupling of different frequencies of shared-aperture array antennas can mostly solve only one problem, either common-mode interference or scattered interference, and only a few studies have simultaneously eliminated common-mode interference and scattering interference. Moreover, most decoupling methods will introduce certain losses and occupy additional space, which is not conducive to integrating and miniaturizing multi-frequency shared-aperture array antennas. To be able to effectively decouple different frequencies and suppress common-mode interference and scattered interference to keep the radiation pattern conformal while not affecting the performance and size of the antenna is an urgent problem.

Therefore, this paper proposes a new type of dual-frequency shared-aperture array antenna, which integrates four high-frequency filtering antenna units and one medium-frequency electromagnetic transparent antenna unit. It can effectively reduce coupling without adding additional decoupling structures, weakening common-mode and scattering interference and thus making the radiation pattern conformal. Additionally, it has a higher gain within the working frequency band. It has important research value and broad application prospects in the field of shared-aperture base station antennas.

2. High-Frequency Filter Unit Design

2.1. Antenna Structure Design

Figure 2a shows the overall structure of the antenna, which consists of a radiating surface, a feeding plate, a reflector, a coaxial cable, and a copper column. Two copper columns and 50 $\mathsf{\Omega }$ coaxial cable connect the radiating surface and the feed board. The copper columns ground the radiating surface and also support and fix it. The size of the reflector is 85 × 85 $\mathrm{m}{\mathrm{m}}^2$. The radiation surface structure is shown in Figure 2b. Two dipole radiation patches are vertically placed and printed on the front of the dielectric plate. The size of the dielectric plate is 32 × 32 $\mathrm{m}{\mathrm{m}}^2$, with a thickness of 0.762 mm, a dielectric constant of 3.0, and a loss tangent of 0.008. A metal ring and a rectangular patch with an L-shaped groove are printed on the back of the dielectric board to increase the effective path of the current and play a role in impedance matching. Four circular through-holes are reserved on the heat dissipation surface for installing nylon columns to facilitate the fixing and welding of the heat dissipation surface.

2.2. Antenna Design Process

Figure 3 shows the antenna’s design process. First, two pairs of elliptical radiation patches are printed on the front of the dielectric plate to obtain antenna 1, which has good radiation characteristics within the working frequency band. A rectangular patch with an L-shaped groove is loaded on the back of the dielectric plate of antenna 1 to obtain antenna 2. Through a simulation analysis, it is known that loading rectangular patches of different sizes on the back of the dielectric plate will affect the filtering effect of antenna 2 and the radiation characteristics within the working frequency band. The filtering effect of antenna 2 is shown in Figure 4. After adding rectangular patches on the back of the dielectric plate, the frequency selectivity becomes better and the filtering effect is significantly enhanced. As the size of the rectangular patch decreases, the edge of the gain curve becomes steeper, the bandwidth of the filtering band also increases, and the filtering effect also becomes better. However, loading rectangular patches will cause resonance within the working frequency band of antenna 2, making antenna 2 unable to radiate normally. The smaller the size of the rectangular patch, the more the in-band gain of antenna 2 drops sharply, so the length of the rectangular patch is finally selected to be 31 mm, sacrificing a part of the filtering effect to ensure the stability of its radiation characteristics.

Add a circular ring based on antenna 2 to obtain antenna 3. The circular ring is used to adjust the impedance matching of antenna 3. Figure 5 shows the impedance values of the two ports when circular rings of different radii are added to antenna 3. It can be seen that the radius of the circular ring is 7 mm, the impedance value of the two ports of antenna 3 is about 50 $\mathsf{\Omega }$, and the impedance matching is good. As the radius of the circular ring increases, the impedance is completely mismatched, so it is finally decided to add a circular ring with a radius of 7 mm. Adding this circular ring will not affect its filtering effect. Antenna 4 is obtained by digging grooves on the radiation patch on the front side of the dielectric plate of antenna 3. The grooves are used to adjust the VSWR and isolation of the antenna. The VSWR and isolation comparison of antenna 3 and antenna 4 is shown in Figure 6. The VSWR of antenna 4 in the 3.6–3.7 GHz frequency band is reduced from 2 to below 1.5, and the isolation is also reduced at the two frequency points of 3.3 GHz and 3.7 GHz. Compared with antenna 3, the VSWR and isolation of antenna 4 are significantly improved. When one antenna polarization works, the other polarization will couple to a part of the energy, resulting in poor VSWR and isolation. Therefore, the current distribution on the surface of the radiation patch is observed, grooves at the position with high current intensity are dug, the current path is extended, and the currents on both sides of the groove are equal in magnitude and opposite in direction, which cancel each other out and can thus effectively improve the VSWR and isolation of the antenna.

The antenna filtering characteristics are generated by the special design of the antenna structure itself. To study its filtering principle, the current distribution of the antenna at the center frequency of the filtering band, 2.2 GHz, can be analyzed. As shown in Figure 7, the direction of the surface current vector synthesis of the dipole radiation patch on the front of the dielectric plate and the rectangular patch on the back of the dielectric plate is shown by a black arrow. The overall direction of the current on the surface of the dipole radiation patch and the rectangular patch is opposite, and the radiation fields generated cancel each other out, thus realizing the filtering function. Therefore, when the high-frequency filtering antenna forms an array with the medium-frequency antenna unit, it can make the coupled intermediate-frequency currents cancel each other out and reduce common-mode interference.

2.3. Simulation and Measure

To study the actual performance of this design, a physical antenna is processed for verification. The physical antenna and the environment of microwave anechoic chamber measure are shown in Figure 8.

Figure 9 is a comparison of the VSWR and isolation results of the high-frequency antenna measured and simulated. It can be seen that the VSWR simulation result is less than 1.45, and the isolation simulation result is less than −23 dB. There is a slight difference between the measured and simulated results, which is due to the influence of the amount of solder in the coaxial cable.

Figure 10 is a comparison of simulated and measured gain. The measured gain in the 3.30–3.70 GHz operating frequency band slightly differs from the simulated gain, but the average gain is 9.0 dBi. There is a big difference between the simulated and measured antenna filtering effects. This is because the simulation is performed under ideal environmental conditions, and no environmental factor (electromagnetic waves in space, temperature, humidity, microwave equipment, metal conductors, etc.) affects the antenna radiation. From the measured gain curve in Figure 10, it can be seen that in the 1.71–2.70 GHz frequency band, the antenna gain is less than 0 dBi, and the maximum suppression can reach below −20 dB, which can effectively filter out the medium-frequency band.

3. Medium-Frequency Electromagnetic Transparent Unit Design

3.1. Antenna Structure Design

The overall structure of the medium-frequency antenna unit is shown in Figure 11a. The antenna consists of a radiation surface, a balun, a feeding plate, and a reflector. The radiation surface consists of two pairs of symmetrical window-shaped metal radiation patches printed on the upper layer of the dielectric plate. The size of the dielectric plate is 55 × 55 $\mathrm{m}{\mathrm{m}}^2$, the thickness is 0.762 mm, and the dielectric constant is 3.0. The two baluns are placed perpendicular to each other. The balun consists of a $\mathsf{\Gamma }$-shaped microstrip transmission line and a metal patch printed on the front and back surfaces of the dielectric board. The excitation at the port is coupled to the radiation surface through the $\mathsf{\Gamma }$-shaped microstrip transmission line to complete the antenna feeding. The metal patch printed on the back of the balun dielectric board serves as a grounding. A matching circuit is printed on the front of the feed plate to adjust the antenna’s impedance matching. The size of the reflector is 150 × 150$\mathrm{m}{\mathrm{m}}^2$. The sizes of the antenna radiation surface, balun, and matching circuit are shown in Figure 11b–e.

3.2. Antenna Design Process

First, a conventional medium-frequency antenna unit is designed, as shown in antenna 1 in Figure 12, which consists of two pairs of vertically placed rectangular patches. The initial length W of the rectangular patch can be calculated by Equations (1) and (2):

$$L_1={\displaystyle \frac{c}{4f_0}}$$

(1)

$$W=\sqrt{{\displaystyle \frac{{\left(L_1\right)}^2}{2}}}$$

(2)

where c is the speed of light, $L_1$ is the length of the quarter-wavelength dipole arm, and $f_0$ is the center frequency of the antenna working frequency band. Through simulation analysis and taking into account the cable loss in actual assembly, W = 25 mm and L = 55 mm are finally determined. In the 1.71–2.70 GHz frequency band, the gain is $9.0\pm 0.5$ dBi, and the H-plane wave width is $60\pm 5^{\circ }$, which meets the market index requirements.

Radar cross section (RCS) is a physical quantity that quantitatively describes the scattering strength of an object. Therefore, the electromagnetic transparency characteristics of an antenna can be analyzed by RCS [41,42,43,44,45,46,47]. The analysis and design process of the medium-frequency unit to achieve electromagnetic transparency is shown in Figure 13, Figure 14 and Figure 15. Figure 14 shows the RCS values of antenna 2, antenna 3, and antenna 4 when the linearly polarized plane wave is vertically incident. Figure 15 shows the electric field distribution of the high-frequency unit at 3.5 GHz. As shown in Figure 15a, antenna 2, antenna 3, and antenna 4 are placed directly above the high-frequency unit. Figure 15b–d show the electric field distribution of the high-frequency unit below antenna 2, antenna 3, and antenna 4, respectively. Figure 15e is the electric field distribution when the high-frequency unit radiates alone.

Antenna 2 is obtained by loading four rectangles inside antenna 1. The RCS value of antenna 2 is shown in Figure 14. Resonance occurs at the 3.4 GHz frequency point, and the RCS value can reach below −20 dB, but the RCS value is only about −15 dB in the 3.5–3.7 GHz frequency band. The electromagnetic transparent effect can be seen in Figure 15b. Only a few electromagnetic waves of the high-frequency antenna unit can penetrate antenna 2, and the electromagnetic transparent effect of antenna 2 is very poor. Antenna 3 adds four open resonant rings based on antenna 2. From Figure 14 and Figure 15c, the RCS value of antenna 3 is lower than that of antenna 2. Some energy of the high-frequency antenna unit can be radiated through antenna 3, but compared with Figure 15e, it can be found that the radiated energy is weak, and the electric field strength is far less than the electric field strength when the high-frequency antenna unit radiates alone. Antenna 4 adds multiple open branches based on antenna 3. From Figure 14, the RCS value of antenna 4 is significantly reduced, and the RCS value is less than −25 dB in the entire high-frequency band. As shown in Figure 15d,e, the electric field strength of the high-frequency antenna unit is the same, indicating that most of the energy of the high-frequency antenna unit can be radiated normally through antenna 4. Antenna 4 has a good electromagnetic transparent effect and can well realize the “stealth” function.

The principle of the medium-frequency electromagnetic transparent antenna unit, which achieves the electromagnetic transparent function and suppresses scattering interference, can be analyzed by the current distribution on the antenna surface. Figure 16 shows the current distribution of the medium-frequency antenna at 3.5 GHz. The black arrow shows the current direction on the radiation arm of the medium-frequency antenna unit, and the surface current direction of each pair of dipoles is opposite, canceling each other out. Therefore, when forming an array with the high-frequency antenna unit, the high-frequency current coupled to the surface of the medium-frequency antenna unit will not generate additional radiation fields to affect the radiation of the high-frequency antenna unit itself and can reduce the scattering interference to the high-frequency antenna unit.

3.3. Simulation and Measure

The medium-frequency antenna unit is processed and measured. The physical antenna and environment of microwave anechoic chamber measure are shown in Figure 17.

The comparison between the VSWR and isolation simulation and measurement results is shown in Figure 18. The measured and simulated VSWR of the antenna’s two ports are below 1.5, and the isolation simulation result is −25 dB. The measured result is slightly worse, but also less than −20 dB.

Figure 19 compares the measured and simulated gain of the medium-frequency antenna unit. The average gain of the antenna simulation is 9.0 dBi, and the measured average gain is lower, at 8.5 dBi. The reason for this deviation is the actual processing error and cable loss.

4. Design of Dual-Frequency Shared-Aperture Array Antenna

Designing a dual-frequency shared-aperture antenna array that integrates a filtering high-frequency antenna unit and electromagnetic transparent medium-frequency antenna unit can effectively reduce the coupling of different frequencies and make the radiation pattern conformal. Figure 20a shows the structure of the array, which consists of a medium-frequency unit and four high-frequency units. The high-frequency antenna unit is 13 mm high and the medium-frequency antenna unit is 34 mm high, placed on a 150 × 150 $\mathrm{m}{\mathrm{m}}^2$ reflector. The four high-frequency antenna units are 70 mm apart and form two 2 × 1 linear arrays, which are arranged on the left and right sides of the medium-frequency antenna unit. As shown in Figure 20b,c, the physical antenna is assembled and the radiation pattern is measured in a microwave anechoic chamber to verify the decoupling effect.

To better illustrate the principle of the conformal radiation pattern of the shared-aperture array antenna, the current distribution at the center frequency of the medium-frequency band and high-frequency band of the shared-aperture array antenna in Figure 21 can be analyzed. As shown in Figure 21, the black arrows indicate the direction of the current. When the two high-frequency units on the left side of the medium-frequency unit are excited at the same time, the shared-aperture array antenna works in the high-frequency band. The high-frequency units radiate normally, the surface of the medium-frequency unit is coupled to a part of the high-frequency current, and the high-frequency current flows in opposite directions on the arm of the medium-frequency unit, canceling each other out, and no additional radiation field will be generated to interfere with the normal radiation of the high-frequency units. It shows that the medium-frequency antenna unit has good electromagnetic transparent characteristics, which can effectively suppress scattering interference and make the high-frequency unit radiation pattern conformal. When the medium-frequency unit is excited, the shared-aperture array antenna works in the medium-frequency band. The medium-frequency unit radiates normally; the surface of the high-frequency unit is coupled to a part of the medium-frequency current and the medium-frequency current flows in opposite directions on the arm of the high-frequency units, canceling each other out. It will not affect the radiation of the medium-frequency unit. This shows that the high-frequency units have a good filtering effect, which can effectively weaken the common-mode coupling and make the medium-frequency unit radiation pattern conformal. Therefore, the dual-frequency shared-aperture array antenna that integrates high-frequency filtering units and medium-frequency electromagnetic transparent unit has excellent different frequency decoupling capabilities, which can solve the problem of radiation pattern distortion caused by the close arrangement of different frequency band units of the shared-aperture base station antenna.

Simulation and Measure

The dual-frequency shared-aperture array antenna operates in two frequency bands: the medium-frequency (1.71–2.70 GHz) and the high-frequency (3.30–3.70 GHz). The VSWR and isolation are measured using a Rohde & Schwarz vector network analyzer. Since the results of the two high-frequency 2 × 1 linear arrays are similar, the high-frequency measure results of the shared-aperture antenna in this paper are all the test results of the high-frequency units’ linear array on the left side of the medium-frequency antenna unit. As shown in Figure 22, the simulation and VSWR measure of the shared-aperture array antenna at the medium-frequency are both less than 1.5, and the isolation is less than −25 dB. The simulation and VSWR measure of the high-frequency are both less than 1.5, and the isolation is less than −22 dB. The gains of the two frequency bands of the shared-aperture antenna are shown in Figure 23. The average simulation gain of the medium-frequency is 8.8 dBi, and the measure average gain is 8.5 dBi. The average simulation gain of the high-frequency is 12.2 dBi, and the measure average gain is 11.7 dBi. The low measured gain is caused by the loss of the cable. In all working frequency bands, the radiation efficiency of the shared-aperture array antenna is above 85%.

As shown in Figure 24, the radiation pattern of the shared-aperture array antenna in the medium-frequency band is compared with the radiation pattern of a single medium-frequency antenna unit. It can be seen that the shapes of the radiation patterns are the same, indicating that the high-frequency antenna units of the shared-aperture array antenna can effectively weaken the common-mode interference and keep the medium-frequency radiation pattern conformal. Figure 25 compares the radiation pattern of the shared-aperture array antenna in the high-frequency band and the radiation pattern of the high-frequency array without the medium-frequency antenna unit. It can be seen that the radiation patterns overlap, indicating that the medium-frequency antenna unit of the shared-aperture array antenna can effectively suppress scattering interference and keep the high-frequency radiation pattern conformal. In addition, the medium-frequency and high-frequency axial cross-polarization ratios of the shared-aperture array antenna are both greater than 15 dB, and the front-to-back ratios are both greater than 20 dB. It meets the market indicator requirements of communication base station antennas.

Table 1. Antenna performance comparison.

Reference	Frequency Bands (GHz)	S11 (dB)	Gain (dBi)	Array Spacing ($\mathit{\lambda }$)
[48]	1.60–2.30 & 3.30–3.80	≤−10	7.20 & 10.30	1.50 × 1.50 × 0.38
[49]	1.70–2.36 & 3.30–3.80	≤−14	7.60 & 9.10	1.16 × 1.16 × 0.39
[50]	2.26–2.73 & 3.00–4.30	≤−14	8.90 & 10.00	1.64 × 1.42 × 0.26
[51]	1.70–2.26 & 3.30–3.70	≤−14	8.00 & 11.00	1.80 × 1.80 × 0.51
[52]	1.71–2.17 & 3.30–3.80	≤−12	8.50 & 11.20	1.44 × 1.44 × 0.43
This work	1.71–2.70 & 3.30–3.70	≤−14	8.80 & 12.20	1.50 × 1.50 × 0.34

compares this work with other similar dual-frequency shared-aperture antennas, where $\lambda$ represents the guided wavelength at 3.0 GHz frequency. Compared with other references, this design has a wider working bandwidth in the medium-frequency band and higher gain in the high-frequency band. The average gain in the high-frequency band is 12.2 dBi, which is at least 1.0 dBi higher than that of other references in the high-frequency band.

5. Conclusions

This paper proposes a dual-frequency shared-aperture base station antenna that integrates a high-frequency filtering antenna unit and a medium-frequency electromagnetic transparent antenna unit, and processes and verifies the proposed shared-aperture antenna. The results show that the shared-aperture antenna can effectively reduce the coupling of different frequencies without adding additional decoupling structures, making the radiation pattern conformal, and performing well within the working frequency band. This design is suitable for the field of shared-aperture base station antennas. It can make the different-frequency antenna units more compact and more flexible in layout, greatly reducing the size of the base station antennas. This paper solved the coupling problem between the medium-frequency band and the high-frequency band of the base station antenna. However, some base station antennas include more working frequency bands. In the future, we can try to solve the coupling problem of three or more frequency bands at the same time based on this paper.