Low-Backward Radiation Circular Polarization RFID Reader Antenna Design for Sports-Event Applications

Abstract

This paper presents the design of a circularly polarized RFID ground mat antenna for UHF-band sports-event applications. Considering a practical sports-event timing system, the ground-based mat antenna with characteristics of a low-backward radiation and circular polarization is proposed. A multilayer square patch antenna using an acrylic dielectric substrate with a wideband branch-line coupler feeding network is employed to improve overall radiation efficiency, which, in turn, provides two excitation port with a phase difference of 90°. Thus, right-hand circular polarization can be obtained. Instead of a conventional FR4–air–FR4 structure, the proposed FR4–acrylic–FR4 composite configuration is adopted to substantially increase the antenna’s mechanical strength and durability against external pressure from runners. The antenna’s performance is attributed to the use of an effective composite dielectric constant and an optimized design of its parameters. Additionally, the patch antenna’s low-backward radiation characteristic helps reduce multipath interference in real-world applications. The measured results are in good agreement with the simulated data, validating the proposed antenna design. In order to further assess the practical performance of the antenna, outdoor measurements are carried out to validate the estimated reading distances derived from controlled anechoic chamber tests. The measured return loss remained below −10 dB across the frequency range of 755–990 MHz, exhibiting a slight discrepancy compared to the simulated bandwidth of 800–1030 MHz. For the characteristic of the circular polarization, the measured axial ratio is below 3 dB within the range of 860–920 MHz. While a more relaxed criterion of an axial ratio below 6 dB is considered, the operating frequency range extends from 560 MHz to 985 MHz, which falls within the frequency band relevant for RFID reader applications.

1. Introduction

In recent years, the outbreak of the COVID-19 pandemic significantly disrupted the organization of numerous sporting events. However, in the post-pandemic era, sports activities have gradually regained momentum, highlighting their enduring importance across all age groups. The sports industry, particularly sectors that integrate advanced technologies, has become a vital component of future economic development. Among these technologies, Radio Frequency Identification (RFID) systems have played a crucial role in various sporting events [1,2,3,4,5,6]. RFID has been widely adopted in timing and data analytics for short-, medium-, and long-distance marathons, cycling races, motor sports, football matches, pigeon racing, and so on. These systems enable real-time tracking and monitoring of individual athletic performance. By providing accurate and impartial timing through wireless and contactless data transmission, RFID not only reduces the need for manual labor but also enhances the fairness and efficiency of competitions.

RFID floor mat antennas are commonly deployed at the start and finish lines of marathon events to record the athletes’ departure and arrival times, thereby determining the total duration of the race. However, polarization mismatch is a critical issue because most antennas and RFID tags are linearly polarized. When a linearly polarized antenna encounters an RFID tag with an orthogonal linear polarization, the polarization loss factor (PLF) can approach zero [7,8], resulting in the failure of the reader antenna to effectively excite the tag for time recording. To mitigate this issue and reduce the likelihood of missed tag reads caused by polarization loss, circularly polarized (CP) antenna designs are employed [9,10,11,12]. Reference [11] proposed a circularly polarized slot-type UHF RFID antenna that achieves broader impedance and axial ratio bandwidths by incorporating slot techniques. This antenna also features low-backward radiation. The feeding structure on the ground plane adopts an L-shaped metal strip, and by tuning the length and width of this strip, two orthogonal modes (with equal phase and a 90° phase difference) can be excited within the slot antenna. Moreover, as documented in [5,9,12], slot antennas generally exhibit broader reflection coefficient bandwidths, wider axial ratio bandwidths, and more compact sizes compared to other types of antennas.

Consequently, this study adopts a square slot antenna as the fundamental design. Since the reader antenna developed here is intended to be embedded within a floor mat, undesired back radiation could introduce additional interference, which may compromise the reliable activation of RFID tags [7]. To mitigate this, the slotted ground choke technique is implemented on the ground plane. By adjusting the dimensions and layout of these slots, the surface current distribution can be modified effectively. Additionally, instead of employing the L-shaped strip feeding network described in [11], the proposed design adopts a branch-line coupler as its feeding mechanism. Leveraging the inherent characteristics of the branch-line coupler, the antenna can easily generate two input signals with equal amplitude and a 90° phase difference. This configuration enables the antenna to operate in the TM₁₁ mode, thereby achieving circularly polarized radiation characteristics. The measured axial ratio below 3 dB indicates that excellent circular polarization is achieved over the 860–920 MHz range. By adopting a more relaxed criterion of an axial ratio below 6 dB, the operating bandwidth can be further extended to 560–985 MHz. Furthermore, to enhance the mechanical robustness of the RFID ground antenna embedded within the floor mat, the design intentionally avoids adopting an air-filled slot antenna structure [9], which could be easily damaged under repeated impact from marathon runners. This paper proposed a novel architecture with physical multilayer configuration (FR4–acrylic–FR4). The issue of runners stepping on the mat has been resolved, while the antenna efficiency has also been significantly improved due to a physical intermediate layer. Thus, a compound structure is introduced to enhance mechanical robustness against runner impact while simultaneously improving antenna efficiency. In addition, the feeding network is located at the bottom layer of the antenna structure. In addition, the UHF RFID frequency allocation varies across regions due to local regulatory standards. In Europe, RFID systems typically operate within 865–868 MHz, while in the United States, the standard operating band is 902–928 MHz. In China, UHF RFID is allocated to the 920–925 MHz range, and in Japan, the frequency range extends from approximately 916.8 MHz to 923.4 MHz. Considering these regional differences and to ensure compatibility and broader applicability, this study adopts a UHF operating bandwidth of 860–960 MHz for antenna design and evaluation.

The architecture of the paper is organized as follows: Section 2 presents the detailed antenna design and its corresponding theoretical analyses and simulated verifications. Section 3 provides details on the experimental results and their discussion. Section 4 concludes the paper. Finally, Appendix A compares the measured read distance with the commercial antenna reader.

2. Slotted Ground Choke Reader Antenna with Circular Polarization

The RFID reader antenna with low-backward radiation is fabricated in a 3-layer physical stacked configuration (FR4–acrylic–FR4) to form a multilayer square patch antenna, which has the dielectric constants of 4.4, 2.8, and 4.4, respectively. The antenna architecture is illustrated in Figure 1, with a branch-line coupler feeding network at the bottom layer to excite equal amplitude and different phase of 90° for circular polarization. Instead of an all FR4-layer structure, the material of acrylic is utilized in the intermediate layer to lower the tangent loss and then improve the overall radiation efficiency [13]. According to Poynting’s theorem, power dissipation can be expressed as:

$$ϵ={ϵ}_r{ϵ}_0(1-j\mathit{tan}\delta ),$$

(1)

$$P_{loss}={\displaystyle \frac{\sigma }{2}}{\int }_V{\left|\overline{E}\right|}^{2}dv+{\displaystyle \frac{\omega }{2}}{\int }_V\left({ϵ}^{″}{\left|\overline{E}\right|}^2+{\mu }^{″}{\left|\overline{H}\right|}^2\right)dv,$$

(2)

where ${ϵ}_r \mathrm{a}\mathrm{n}\mathrm{d} {ϵ}_0$ are the permittivity of the material and free space, respectively, while μ is the permeability of the material.

Equation (2) shows that a minimized $\mathit{tan}\delta$ can improve power loss, which, in turn, increases overall radiation efficiency. The loss tangent of acrylic material is approximately 0.017, which is slightly lower than that of FR4. Therefore, the FR4–acrylic–FR4 structure is adopted. The effective permittivity of the composite multilayer structure can be obtained by [14]:

$${\epsilon }_{rc}={\displaystyle \frac{{\sum }_{i=1}^n{h}_i}{{\sum }_{i=1}^n\frac{h_i}{{\epsilon }_{ri}}}},$$

(3)

Based on Equation (3), the compound permittivity is around 2.898, which is lower than that of all FR4 layers. Replacing the intermediate layer of FR4 in acrylic reduces the loss tangent within the dielectric layer of the patch antenna in the cavity model, thereby enhancing its efficiency.

Reference [13] introduces the concept of the effective dielectric constant (${\epsilon }_{eff}$). Since the electric field lines from the patch antenna propagate to the ground plane through both air and the dielectric substrate, the resulting discontinuous field lines can be treated as a strip line operating within a material characterized by an effective dielectric constant. The electric field is thus approximated as a continuous field. For the case where W/h ≥ 1, the effective dielectric constant ${\epsilon }_{eff}$ can be approximately calculated as follows:

$${\epsilon }_{eff}={\displaystyle \frac{{\epsilon }_r+1}{2}}+{\displaystyle \frac{{\epsilon }_r-1}{2}}\sqrt{{\displaystyle \frac{1}{1+\frac{12h}{W}}}}$$

(4)

According to Equation (4), the physical length L′ required for the patch antenna to radiate can be calculated as:

$$L^{\prime }\approx {\displaystyle \frac{{\lambda }_g}{2}}={\displaystyle \frac{1}{2f_r\sqrt{{\epsilon }_r}\sqrt{{{\epsilon }_0\mu }_0}}}$$

(5)

where ${\epsilon }_r$ is the relative dielectric constant of the substrate, and ${\epsilon }_0 \& {\mu }_0$ are the permittivity and permeability of free space, respectively. However, due to the edge effects arising from the propagation of the electric field through both air and the substrate, the electric length of the field lines within the substrate becomes longer than the actual physical length. Therefore, it is necessary to apply corrections to the physical length using the effective dielectric constant and corresponding formulas. The change in length caused by the edge effects is given by the following expression:

$$L^{\prime }\approx {\displaystyle \frac{{\lambda }_g}{2}}={\displaystyle \frac{1}{2f_r\sqrt{{\epsilon }_r}\sqrt{{{\epsilon }_0\mu }_0}}}$$

$$\mathrm{L}={\mathrm{L}}^{\prime }-2\Delta L$$

(6)

The values of the compound substrate dielectric constant ${\epsilon }_{rc}$ (2.898) calculated above are then substituted into Equations (4)–(6) as the initial parameters for the antenna, followed by parameter investigation and optimization.

In the implementation, the acrylic middle layer was realized by potting the material between the upper and lower FR4 layers. During the process, a mold was used to maintain the desired dimensions, and the assembly was allowed to rest and cure, ensuring that the final dimensions closely matched those parameters used in the simulations. To prevent the antenna’s overall thickness from causing injury to runners due to excessive height when stepping on the mat, the total antenna thickness is limited to a maximum of 10 mm. For manufacturability, the substrate thickness h_p is set to 0.4 mm. To reduce the volume of the FR4 substrate, which exhibits a relatively high loss tangent, within the cavity structure, acrylic material with a lower loss tangent is employed to fill the remaining height h_m, set at 6 mm. This approach effectively reduces the overall internal loss of the cavity. Including the ground plane, the total antenna thickness is 8 mm. Detailed geometrical parameters are given in

Table 1. The design parameters of the proposed antenna structure.

Parameter	Wg	Lg	hg	Ws	Ls	Wp	Lp	hp	hm
Length (mm)	145	145	1.6	3	17	91	91	0.4	6

to the illustration in Figure 1, which were obtained from the complete antenna design for circular polarization.

2.1. Branch-Line Feeding Network

To achieve circular polarization characteristics in a square patch antenna, two excitation sources with equal magnitude and a 90° phase difference can be used. This feeding method enables the antenna to support two orthogonal modes of equal amplitude, thereby generating circular polarization. A branch-line coupler is utilized to generate two excitation signals with equal amplitude and a 90° phase difference through a single input port. This coupler serves as the input feeding network for the square patch antenna, as illustrated in Figure 2. The signal is fed into Port 1, and through impedance matching of the microstrip lines and the phase shift introduced by quarter-wavelength transmission lines, the input signal is equally divided in power and outputs two signals with a 90° phase difference at Port 2 and Port 3. Port 4 functions as the isolated port, which is terminated by a 50 ohm resistor. Figure 3 shows the simulated S parameters and phase difference at output ports. The branch-line coupler is designed using FR4 as the substrate (ε_r = 4.4, tan δ = 0.02, h = 1.6 mm). Simulation results show that within the operating band, output ports 2 and 3 exhibit equal power division of approximately −3 dB and a 90-degree phase difference, while the isolation port (port 4) maintains isolation better than −30 dB. This indicates that the branch-line coupler can function properly within the required 860–960 MHz band. At the center frequency of 910 MHz, the design achieves good impedance matching and nearly equal power splitting, which ensures its suitability for exciting the patch antenna to operate in the TM10 and TM01 modes.

The difference in frequency response between S21 and S31 of the branch-line coupler is primarily due to the fact that the quarter-wavelength transmission lines are designed for a specific center frequency. As the frequency deviates from this design point, variations occur at different output ports. Moreover, in composite structures, impedance mismatches are more likely to arise at frequencies outside the intended operating band, thereby reducing the effective bandwidth. Nevertheless, the resulting bandwidth still exceeds the requirements of the RFID frequency band.

To further verify the circular polarization characteristics, a branch-line coupler is employed as the feeding circuit to generate two orthogonal modes, enabling the square pate antenna to exhibit circular polarization. This configuration allows the antenna to achieve a wider circular polarization bandwidth in the direction of maximum radiation (+z direction). Figure 4 shows the time-varying current density distribution of the square patch antenna, illustrating the evolution of surface currents over time, which indicates a right-hand circular polarization (RHCP) pattern.

2.2. Low-Backward Radiation Verification

In contrast to conventional patch antennas, the second-layer metallic ground plane, as shown in Figure 1b, incorporates the slotted ground choke technique to achieve low-backward radiation for RFID floor mat antenna applications [5,6,10,11,12]. The adoption of slot-based structures in antennas effectively reduces back radiation, which is particularly advantageous for floor-embedded reader antennas. This reduction in back radiation primarily stems from the inherent radiation mechanism of slot antennas, where the electromagnetic energy is radiated through openings in the ground plane rather than directly from protruding conductive elements. As a result, the ground plane itself serves as a shielding layer, significantly attenuating undesired radiation toward the back side. Furthermore, by carefully designing the shape, dimensions, and placement of the slots, the surface current distribution on the ground plane can be precisely controlled. This optimization confines the currents near the slots and away from the edges of the ground plane, thereby suppressing edge diffraction and further lowering back radiation. Techniques such as the slotted ground choke can be employed to enhance this effect by disrupting parasitic surface currents that would otherwise contribute to unwanted radiation. Therefore, by adjusting the dimensions of the slots on the ground plane, the surface current distribution can be effectively modified. This technique enhances the antenna’s overall performance in terms of back radiation suppression. Therefore, by varying the slot width W_s on the ground plane, it is observed that changes to this parameter have minimal impact on the S-parameters, directivity, and circular polarization bandwidth. However, as the W_s decreases, the magnitude of back radiation in the −z direction also reduces, as shown in Figure 5a. This leads to an improvement in the antenna’s front-to-back ratio, which helps minimize back radiation and thereby reduces the likelihood of read failures caused by multipath interference. Nonetheless, this improvement comes at the cost of a slight decrease in radiation efficiency, illustrated in Figure 5b.

The variation of the parameter (L_s) exhibits negligible influence on the antenna’s S-parameters; however, it contributes to a slight upward shift in the center frequency of the circular polarization bandwidth. As L_s increases, a moderate degradation in directivity is observed, accompanied by an improvement in radiation efficiency, which results in a net increase in overall gain, as illustrated in Figure 6a. Furthermore, under certain conditions, the gain response across the operating frequency band becomes more uniform, effectively mitigating the gain roll-off observed from 860 MHz and 960 MHz. This flattening of the gain curve enhances frequency stability. Nevertheless, an increase in L_s also leads to elevated back radiation levels, as depicted in Figure 6b, c, and d at 860 MHz, 910 MHz, and 960 MHz, respectively, thereby reducing the front-to-back ratio. From the simulation results, it can be observed that when L_s is reduced below 30 mm, the improvement in backward radiation suppression becomes limited, while the antenna gain slightly decreases. Therefore, in the antenna design, a moderate value of L_s is selected to achieve substantial backward radiation suppression while maintaining a reasonable antenna gain. This trade-off highlights the need for an optimization between gain enhancement and the suppression of undesired back radiation.

2.3. Efficiency Consideration

In addition to dual excitation and slot ground choke for the proposed RFID antenna with circular polarization and low-backward radiation, respectively, efficiency is another issue to extend reading distance of the ground mat reader antenna. To address the issue of low efficiency, a multi-layered architecture is proposed as an effective solution to enhance overall performance. The radiation efficiency ($\eta$) of an antenna is related to its radiated power ($P_{rad}$), which can be expressed as:

$$P_{rad}={\mathsf{\eta }}_{cd}\ast P_{acc},$$

(7)

$$\eta ={\eta }_r\ast {\eta }_{cd},$$

(8)

where $P_{rad}$ represents the radiated power of the antenna, $P_{acc}$ denotes the power delivered to the antenna after accounting for matching losses, ${\eta }_{cd}$ is defined as the radiation efficiency, and ${\eta }_r$ is defined as the reflection efficiency of the antenna, representing the efficiency loss due to impedance mismatch. According to (7) and (8), enhancing the radiated power is proposed as a solution to improve both radiation efficiency and antenna gain. Thus, to reduce the loss tangent (tan δ) of the intermediate dielectric layer, acrylic material was adopted as the filler for the intermediate layer, as shown in Figure 1b. The overall substrate can be significantly reduced by a lower equivalent dielectric constant of 2.898 in this the multilayer structure.

To verify that the physical multilayer antenna (FR4–acrylic–FR4) offers advantages in efficiency and gain compared to a fully solid FR4 structure, we first matched the input of both configurations to the desired frequency band in order to minimize efficiency loss caused by impedance mismatch (decrease ${\eta }_r$). As shown in Figure 7, after replacing the intermediate dielectric layer with acrylic material, which has a lower loss tangent, the efficiency improved significantly. The entire efficiency is boosted from an initial value of 0.5 to approximately 0.8–0.9. Furthermore, assuming that the antenna’s directivity remains largely unchanged, it can be inferred that the antenna gain is likely to improve. It is worth mentioning that the simulated efficiency increases significantly at 1.35 GHz. However, the gain improvement is not as prominent. The frequency point at 1.35 GHz is outside the main operating frequency band, resulting in poor input impedance matching, which in turn limits the gain. Although the efficiency is higher at 1.35 GHz, the gain remains relatively low. Figure 8 depicts the comparison of gains between the two structures. The corresponding antenna gain also shows a significant improvement. Therefore, by adopting an FR4–acrylic–FR4 configuration, the overall effective dielectric constant is reduced, and with appropriate parameter design, the radiation efficiency of the antenna can be improved.

3. Measurement Validation

3.1. RFID Floor Mat Antenna Performance

A compound multilayer square patch antenna with slotted ground choke technique was implemented for measurement validation. The top and bottom views of an antenna prototype are shown by photographs in Figure 9a,b, where the input feeding network was constructed with a branch-line coupler to form a dual excitation for circular polarization, measured in the anechoic chamber, as shown in Figure 9c.

Th input return loss was first examined to evaluate the input matching behavior with a maximized reflection efficiency ${\eta }_r$, as shown in Figure 10. The measured results exhibited a frequency shift toward the lower end compared to the simulated results in the operating band. The measured return loss remained below −10 dB in the frequency range from 755 MHz to 990 MHz, which shows a slight deviation from the simulated range of 800 MHz to 1030 MHz. The input mismatch issue mainly arises from small air gaps between the bonded layers in the three-layer structure. Additionally, the SMA connector at the input side also contributes to the observed frequency shift. However, due to the wide bandwidth characteristic of the proposed antenna design, the return loss within the operating band still meets the design specifications. To further demonstrate the antenna’s radiation behavior, Figure 11a and b illustrate the simulated and measured radiation patterns at frequency of 910 MHz in x-z cut and y-z cut, respectively. The results consistently demonstrate its low-backward radiation characteristics, confirming the intended performance. Furthermore, the measured results indicate that the half-power beamwidth (HPBW) of the antenna is approximately 100 degrees. A wide HPBW contributes to an extended effective reading range. In addition, the front-to-back ratio is approximately −14.5 dB. Figure 12 illustrates the measured antenna gain performance across the full frequency band. The results show a discrepancy of approximately 1.7 dB between the measured and simulated gains. The peak gain within the operating band is approximately 3.85 dBi, which is slightly lower than the simulated value of 5.5 dBi. The discrepancy between the simulated and measured gain may be attributed to factors such as differences between the actual compound permittivity of the packaging configuration and the calculated value, as well as discontinuities introduced by the soldering surfaces. Additionally, the multi-layer (FR4–acrylic–FR4) structure implementation may contribute some losses.

In addition, a branch-line coupler is utilized as the feeding network to excite two mutually orthogonal modes (RHCP). Figure 13 depicts the measured antenna’s axial ratio to examine the property of circular polarization. The discrepancies between the simulated and measured axial ratios are primarily caused by errors introduced in the measured accuracy. These errors, when processed through the calculation formulas, lead to the observed differences. Based on the calculated data, it was found that the bandwidth over which the antenna maintains an axial ratio below 3 dB is limited to the range of 860 MHz to 920 MHz. When considering a broader criterion of an axial ratio below 6 dB, the frequency range extends from 560 MHz to 985 MHz, which falls within the frequency band applicable to RFID reader applications.

To further evaluate the circular polarization characteristics and corresponding gain of the RFID reader antenna,

Table 2. Comparison of the proposed antenna and other RFID.

Reference	Dimensions (mm3)	Frequency Range & BW (MHz)	Peak Gain (dBic)	Front-to-Back Ratio (dB)	Radiation Direction
ICMMT 2018 [5]	120 × 120 × 1.6	706–1007 (301)	4.8/CP	n/a	$\pm z$
IEEE TAP 2015 [6]	130 × 120 × 0.8	618–998 (380)	3.41/CP	n/a	$\pm z$
IEEE TAP 2017 [15]	150 × 110 × 1	780–1070 (290)	3.4/CP	n/a	$\pm z$
IEEE JRFID 2022 [16]	68 × 68 × 1.6	903–927 (24)	2.9/CP	n/a	$\pm z$
IEEE RFID-TA 2019 [17]	200 × 200 × 57	618–1105 (487)	7.97/CP	15 *	+z
IEEE TAP 2020 [18]	156 × 140 × 41	837–1400 (563)	7.2/CP	13 *	+z
	250 × 250 × 33	881–947 (66)	8/CP	12 *	+z
This work	145 × 145 × 8	755–900 (235)	6.7/CP	14.5	+z

* Estimated from measured antenna pattern.

presents a comparison with previously reported reader antennas in the literature. The circularly polarized antenna gain (dBic) is calculated as:

$$G_{CP}\left(dBic\right)=20 {\mathit{log}}_{10}\left(a\right)+20 {\mathit{log}}_{10}\left({\displaystyle \frac{1+{10}^{(-{AR}_{\left(dB\right)})/20}}{\sqrt{2}}}\right),$$

(9)

where a represents the semi-major axis of the electric field ellipse and AR denotes the axial ratio.

Although the antennas reported in [5,6,15,16] feature more compact overall dimensions, their circularly polarized gains are relatively lower, and their radiation is primarily directed along the ±z-axis, lacking the characteristic of low-backward radiation. Furthermore, while the antennas in [17,18] exhibit higher circularly polarized gain compared to the proposed design, their overall dimensions are also significantly larger.

3.2. Estimated Effective Reading Range

The antenna designed in this study is intended for use in sports-event timing applications. Figure 14 illustrates the application scenario. The proposed antenna is placed at the finish line of a running track. When an RFID tag worn by a runner passes through the reading zone of the antenna, the finish time is transmitted to the application system, thereby completing the timing process. As shown in the figure, the antenna faces the runners along the x-axis, which corresponds to the track depth (D). The y-axis, defined in the antenna coordinate system, lies across the width of the track (track width W), indicating how far a tag can be laterally offset from the center of the antenna while still being successfully read. Lastly, the z-axis represents the vertical direction above the ground, referred to in this context as the reading height (H). Since most runners wear RFID tags either on the waist or on the surface of their shoes, the performance at reading heights of 10 cm, 50 cm, and 100 cm is evaluated for verification.

During antenna measurements conducted inside an anechoic chamber, the center of the transmitting antenna is always aligned with the center of the antenna under test (AUT). As a result, the measured gain corresponds to the θ and φ components. However, this is not the case in practical application scenarios. Assuming a horizontally polarized RFID tag with an electric field component only in the y-direction, the electric field strength sensed by the tag should be expressed as:

$$\left|E_y\right|=\left|E_{\phi }cos\phi +E_{\theta }cos\theta sin\phi \right|,$$

(10)

Similarly, for a vertically polarized tag passing through the reader antenna’s field, the received electric field strength should be corrected as:

$$\left|E_z\right|=\left|E_{\theta }sin\theta \right|,$$

(11)

Based on the relationship between antenna gain and electric field strength, the correction factors for the gains of the horizontally polarized CF_H and vertically polarized CF_V can be expressed as follows:

$${CF}_H=[{cos}^2\phi +{\left({\displaystyle \frac{E_{\theta }}{E_{\phi }}}\right)}^2{cos}^2\theta {sin}^2\phi ],$$

(12)

$${CF}_V={sin}^2\theta ,$$

(13)

As discussed earlier, the target application scenario focuses on the x–z plane (ϕ = 0°) of the AUT, as well as at azimuth angles ϕ = 30°, 45°, and 60°, for estimating and measuring the corresponding reading distances. Furthermore, the gain values measured in the anechoic chamber are first normalized. These normalized gain values, when combined with the required horizontal or vertical correction factors corresponding to the tag antenna’s polarization in different planes, represent the required received power (P_min) at various spatial positions. This approach enables the estimation of the reading distance under practical deployment conditions. Consequently, when the RFID tag is located at a spatial position (R′, θ, ϕ), the P_min can be used to determine the value of effective reading range (R′), expressed as:

$$R^{\prime }=R_0\sqrt{{\displaystyle \frac{P_t}{P_{min}}}},$$

(14)

where R₀ is a fixed reading range in the chamber. Then, by applying trigonometric relationships, the reading height (H′) with respect to the finish line can be derived for any given position in space. Based on this, the estimated reading depth (D′) and reading width (W′) are calculated for three different reading heights: 10 cm, 50 cm, and 100 cm.

Figure 15a shows the commercial RFID reader module (CF-RU6401) (Shenzhen Chafon Technology Co., Ltd., Shenzhen, China), used in this measurement, while the tag (AZ-9654) (Alien Technology, New Taipei city, Taiwan) is adopted in this paper, as shown in Figure 15b. During the measurement, the distance R₀ between the tag antenna and the proposed reader antenna is fixed at 2 m. By adjusting the transmit power of the reader module via the application software on a connected computer, the required P_min is obtained, which can be used to estimate the corresponding reading distance.

3.3. Outdoor Measurement and Verification of Estimated Reading Distance

To validate the estimated reading distances derived from the anechoic chamber measurements, outdoor circumstance is also measured to emulate a practical application. Therefore, the entire antenna is encapsulated with a protective potting compound. Thin acrylic layers are placed on both the top and bottom surfaces to facilitate integration within the ground mat structure. This design not only provides antenna protection during outdoor events such as marathons, particularly under rainy conditions, but also enhances the mechanical robustness of the antenna against repeated stepping by runners.

The actual outdoor measurement setup is illustrated in Figure 16. An RFID tag was attached to a PVC pipe to ensure that no metallic objects would interfere with the reading range. The AUT was placed horizontally on the ground, and the same reader module (CF-RU6401) used in the chamber tests was employed as the power source. The transmit power of the reader module was set to 30 dBm as a reference. The proposed ground mat antenna was performed at 910 MHz, and the goal was to evaluate the reading distance for RFID tags with different polarization orientations. The performance criterion was defined as the ability of the reader to successfully read at least 100 tags per second.

For measurement on planes with various azimuth angles (ϕ), the tag was moved along the corresponding direction toward the origin (i.e., toward the antenna), and the reading depth D′ and reading width W′ were recorded once the tag was successfully read.

Table 3. Estimation and measurement of reading depth and width on the ϕ = 0° plane (Unit: meter).

	Horizontal Polarization				Vertical Polarization
	D′		W′		D′		W′
H′	E	M	E	M	E	M	E	M
0.1 m	1.01	0.44	0	0	1.31	0.8	0	0
0.5 m	1.3	0.82	0	0	1.57	1.27	0	0
1 m	1.31	1.1	0	0	1.13	1.04	0	0

E: Evaluation; M: Measurement.

,

Table 4. Estimation and measurement of reading depth and width on the ϕ = 30° plane (Unit: meter).

	Horizontal Polarization				Vertical Polarization
	D′		W′		D′		W′
H′	E	M	E	M	E	M	E	M
0.1 m	0.77	0.21	0.44	0.14	0.81	0.58	0.47	0.32
0.5 m	1.32	0.63	0.77	0.33	1.3	0.99	0.75	0.49
1 m	1.1	0.91	0.63	0.49	0.74	0.64	0.43	0.35

E: Evaluation; M: Measurement.

,

Table 5. Estimation and measurement of reading depth and width on the ϕ = 45° plane (Unit: meter).

	Horizontal Polarization				Vertical Polarization
	D′		W′		D′		W′
H′	E	M	E	M	E	M	E	M
0.1 m	0.57	0.14	0.57	0.14	0.59	0.47	0.59	0.47
0.5 m	1	0.43	1	0.43	0.9	0.69	0.9	0.69
1 m	0.83	0.55	0.83	0.55	0.43	0.57	0.43	0.57

E: Evaluation; M: Measurement.

and

Table 6. Estimation and measurement of reading depth and width on the ϕ = 60° plane (Unit: meter).

	Horizontal Polarization				Vertical Polarization
	D′		W′		D′		W′
H′	E	M	E	M	E	M	E	M
0.1 m	0.35	0.1	0.61	0.23	0.41	0.21	0.69	0.5
0.5 m	0.48	0.21	0.84	0.5	0.54	0.42	0.94	0.83
1 m	0.49	0.37	0.85	0.73	0.43	0.44	0.74	0.94

E: Evaluation; M: Measurement.

list comparison of estimated and experimental reading depth and width at ϕ = 0°, 30°, 45°, and 60°, respectively. Based on the measurement results, the maximum sensing distance reached 1.27 m, and the maximum reading width reached 0.94 m. This performance is attributed to the high measured gain of the antenna and its favorable half-power beamwidth characteristics. Furthermore, the results indicate that under identical test conditions, the antenna exhibits superior performance when configured in vertical polarization. These findings also demonstrate that, compared to conventional patch antennas, the reduced back-lobe radiation of the proposed antenna effectively enhances the sensing range. Additionally, the antenna maintains a consistent sensing distance in both horizontal and vertical polarizations, indicating that it possesses circular polarization characteristics.

4. Conclusions

This paper presents the design of a ground mat RFID reader antenna for sports timing systems, with a focus on achieving high front-to-back ratio and circular polarization to effectively reduce missed reads caused by multipath interference. Unlike previous studies that utilized air as the dielectric medium, the proposed design employs a multilayer square patch structure using acrylic dielectric material. This approach enhances mechanical robustness against foot traffic while minimizing the risk of injury due to uneven height differences. Overall, the proposed antenna design demonstrates a balanced improvement over existing solutions by simultaneously achieving compact dimensions, enhanced circularly polarized gain, and reduced back radiation. Additionally, in order to improve the alignment between theoretical predictions and practical measurements, a correction factor is applied to the estimated read range model. Outdoor measurements were conducted in an outdoor environment to emulate practical application scenarios. The performance of the proposed antenna was comprehensively evaluated via theoretical analysis, full-wave electromagnetic simulations, prototype implementation, and field testing under actual marathon event conditions. The measured results demonstrate that the design meets the expected performance and is well suited for deployment in practical sports timing applications.