# Free Space Optic Receiver with Strongly Overlapped Photodetectors’ Field of View

^{*}

## Abstract

**:**

## 1. Introduction

#### 1.1. State-of-the-Art

#### 1.2. Receiver Concept

_{1}to PD

_{6}, which are attached to the walls of the truncated pyramid (Figure 1). The pyramid ground plan is hexagon and its walls and base are at a 45° angle. The photodetectors are mounted on small printed circuit boards (PCBs) and are connected with the following amplifiers, which are located on a separate board below the base of the pyramid. The connections between photodetectors and the following amplifiers had to be as short as possible. For this reason, the photodetectors were finally moved from the centers of the segments to their edges.

#### 1.3. Parallel Amplifier Topology

_{in}in the photodetector’s optical part is given by the total captured useful signal power p

_{s}to the total captured noise power p

_{n}in an optical channel:

_{nA}. The signal-to-noise ratio (SNR) in the photodetector’s electrical part is given by:

_{s}/p

_{n}ratio is given by the photodetector’s optical part and there is no possibility of increasing the ratio in the photodetector’s electrical part. However, if sufficient power amplification K

_{21}

^{2}is not available, the final signal to noise ratio SNR

_{out}is much lower due to p

_{nA}.

_{in}since the spatial angle, along with unwanted spurious signals, is reduced. However, complexity and low reliability come as trade-offs. Note, no easy way of recognizing the strongest signal photodetector exists. Many interference optical signals are due to, for example, artificial illumination, sunlight or lightning, which might be stronger than the optical signal used and can confuse even the strongest signal photodetector selection circuit.

_{s}, together with optical noise power p

_{n}, enter the receiver in the same way. The total useful signal and noise currents must be s times higher than those for one photodetector, and similarly, the total useful signal and noise powers must be s

^{2}higher than those for one photodetector. Each photodetector and its follow-up amplifier contribute their own noise power p

_{nA}. Since all photodetectors are standalone components, their noise powers are uncorrelated and expressed as p

_{nA1}, p

_{nA2}, …, p

_{nAs}. To prove the merits of a parallel amplifier, its uncorrelated noise powers must be summed. The procedure is as follows [27]:

_{n}, n = 1,2,…,N and Y

_{m}, m = 1,2,…,N have the same number of realizations N such that each member of the first sequence X

_{n}makes N possible sums with the members of the second sequence Y

_{m}. This is mathematically expressed as X

_{n}+ Y

_{m}, where m = 1,2,…,N. Since the sequences are the same size, N

^{2}sums with different indexes nm exist as a result.

_{n}sequence member and the all Y

_{m}sequence members can be written as:

_{m}sequence mean value is zero, then the last part of Equation (3) must also be zero. Equation (3) can be shortened to:

_{nA}

_{1}, u

_{nA}

_{2}, …, u

_{nAs}sequences, which are voltage realizations of noise powers p

_{nA}

_{1}, p

_{nA}

_{2}, …, p

_{nAs}. For p

_{nA}

_{1}= p

_{nA}

_{2}= … = p

_{nAs}conformity, Equation (5) can be rewritten as:

^{2}component, p

_{nA}may be eliminated more easily. As result, SNR

_{out}will be almost equal to SNR

_{in}in Equation (1), which is desirable.

#### 1.4. Conditions Preserving Parallel Amplifier Advantages

_{out}in Equation (7) also becomes smaller because the shadowed photodetectors are only delivering noise. Therefore, the aim was to find some marginal conditions that ensure that the SNR

_{out}of all photodetector assemblies (PDAs) are equal to the SNR

_{out}of only one PDA in use. These conditions might be expressed, for example, by a minimum angle between the pyramid base and the direction of falling rays. Unfortunately, there is no universal minimum angle, although many minimum angles are available that vary according to the pyramid’s shape and the number of walls. The minimum angle may also be calculated for any particular case. Analytic geometry offers many powerful tools, some of which are further described. For more details, see Sedlacek, M. et al. [28], Dostal, Z. [29].

_{1}a

_{2}a

_{3}]

^{T}, B = [b

_{1}b

_{2}b

_{3}]

^{T}, C = [c

_{1}c

_{2}c

_{3}]

^{T}, and D = [d

_{1}d

_{2}d

_{3}]

^{T}. These might represent the photodiode active area corners. To rotate the square on the x-axis or z-axis, the vectors must be multiplied by the matrix Arotx or Arotz, respectively, as follows:

_{p}from the P (A), P (B), P (C), and P (D) corners:

_{z}angle (Figure 4). The illumination point Z Cartesian coordinates are calculated separately corresponding to the run along the circle, as shown in Figure 4, and they are not listed here. The only variable listed is the φ

_{z}angle, which is the illumination point Z Polar coordinate (we assumed that midpoint S holds the initial position).

_{z}is shown in Figure 5. All calculations were performed in Excel software (Microsoft, Redmond, WA, USA, 2007).

_{out}= 1. The parallel amplifier sums the useful signal currents and the noise currents of all photodetectors. The total useful signal current will be a six-multiple of one photodetector’s useful signal current. The total noise current will be a $\sqrt{6}$-multiple of one photodetector’s noise current. Assuming that all the sensitive areas of the photodetector are perpendicular to the direction of falling rays, SNR

_{out}will be enhanced ${\left(6/\sqrt{6}\right)}^{2}$ times, which is six times. Since the useful signal current and the sensitive area are proportional [1], the currents and surfaces may be swapped, which matches the curve analysis shown in Figure 5.

_{z}= 90°. If φ

_{z}decreases, the total surface screening also decreases and reaches its minimum at φ

_{z}= 0°, being slightly over 1. However, the noise level surface equivalent stays the same, being equal to $\sqrt{6}=2.449$. To preserve the parallel amplifier’s advantage, the total surface screening must be higher than 2.449. This is fulfilled for angles greater than the minimum angle φ

_{z}= 32°. In other words, at angle φ

_{z}= 32°, the SNR

_{out}value of all PDAs is equal to the SNR

_{out}of only one PDA in use. The best SNR

_{out}is achieved for the angle φ

_{z}= 90°, though the maximum value is only ${\left(3\sqrt{2}/\sqrt{6}\right)}^{2}=3$ (or 4.8 dB). The curve breaks in Figure 5 correspond to the angles when the photodetectors in shadow become illuminated.

_{nA}(the PDA noise power) and enhances the signal to noise ratio SNR

_{out}.

_{n}from an optical channel determines the situation. If it is much lower than the PDA noise power, p

_{n}<< p

_{nA}, Equation (16) becomes:

_{out}may be expected because the numerator grows by the power of two of the illuminated photodetector’s number s’. Enlargement of the photodetector is again beneficial. The only problem is the non-zero noise power p

_{n}from the optical channel. With the relationship p

_{n}> p

_{nA}, Equation (16) becomes:

_{out}. Similarly, any photodetector enlargement will deteriorate the situation.

_{n}.

## 2. Free Space Optic (FSO) Receiver Design

#### 2.1. Receiver and Transmitter Parameters

^{-1}and a total radiant power of 935 mW. Both these values are achievable with a forward current of 1 A. Its beam allows switching between 0 A and 1 A currents with 15 ns and 18 ns rise and fall times, respectively. The LED driver was built as a high current, high frequency source. The tandem driver and the power LED can deliver a pulse train with a data speed as high as 10 Mbit·s

^{−1}.

#### 2.2. FSO Receiver Bandwidth

^{−1}transfer speed but maintains functionality even for a 2 Mbit·s

^{−1}transfer speed, which is an advantage because it broadens the testing capability and includes slower equipment.

^{−1}, some space remained, and the maximum lowered to a transfer speed of 2 Mbit·s

^{−1}. Based on this, the receiver’s low and high cut-off frequencies were 100 kHz and 1.5 MHz, respectively. Please note that the high cut-off frequency follows the rule of optimal filtering. This means that the PE spectral density high frequency decline starting at ¾ f

_{bit}may be approximated by the first order low pass filter roll-off, expressed as the number f

_{h}= ¾ f

_{bit}= ¾ 2 × 10

^{6}= 1.5 MHz. The low cut-off frequency of f

_{l}= 100 kHz was determined from BER measurements on the 2 Mbit·s

^{-1}data stream.

#### 2.3. Selecting a Suitable Photodetector

_{in}in the optical part, as in Equation (1). The input useful signal and input noise are both amplified in the same manner. Internal amplification is afflicted with its own noise, which increases with the magnitude of amplification. Internal amplification is therefore kept low and set only to a necessary magnitude that ensures that the photodetector noise current i

_{nPD}is triple the follow-up amplifier noise current i

_{nA}. Mathematically written:

_{nPD}depends not only on the photodetector amplification magnitude but also on the photodetector background light level. Second, the internal amplification photodetector is unnecessary if the ordinary photodetector noise current i

_{nPD}exceeds the noise current i

_{nA}of the follow-up amplifier. It is obvious that the photodetector environmental conditions, particularly background light level, are important. The discussed FSO receiver’s high background light levels means that Equation (19) is fulfilled even with ordinary PIN photodiodes.

_{nPD}optimization. It is expressed as follows:

_{h}− f

_{l}is the useful signal bandwidth, and the photodiode noise power current spectral density is given by the well-known Schottky equation:

_{0}is the dark current of the photodiode. From Equation (21), only quantity I

_{0}remains which may be tuned by the user. The dark current I

_{0}has three components: I

_{dg}(darkness generation current induced by phonons), I

_{db}(darkness background current induced by far infrared /FIR/ photons), and finally I

_{pb}(photocurrent background induced by near infrared /NIR/ and visible light photons). Written mathematically [1]:

_{pb}current greater than the others. In an NIR photodetector and NIR application, I

_{db}is almost zero (I

_{db}= 72.97 × 10

^{−21}A for BPW34 and I

_{db}= 47.09 × 10

^{−21}A for BP104F), and I

_{dg}is negligible (I

_{dg}= 2 nA, according to the datasheets of both photodiodes). The result is that the dark current I

_{0}is determined only by I

_{pb}. The total I

_{pb}current can be found from the relative spectral sensitivity curve integration. Please note that the narrower spectral bandwidth range has a lower current I

_{pb}and lower current noise i

_{nPD}. Comparing the BPW34 and BP104F photodiodes, only the BP104F photodiode ensures lower I

_{pb}currents and lower current noise i

_{nPD}. Lower BP104F currents were confirmed by measurements for several types of illumination sources, shown in Table 1. The true I

_{dg}currents are also provided, confirming there is no better choice than the BP104F photodiode.

_{PD}as permitted.

#### 2.4. Selecting a Suitable Follow-Up Amplifier

_{ps}is the power supply voltage, C

_{ps}is power supply decoupling capacitance, and R

_{L}is load resistance. For variant (a), the amplifier input impedance Z

_{a}is not expressed and is covered by resistance R

_{L}. Similarly, for variant (b), the resistance R

_{L}is not expressed and is covered by feedback impedance Z

_{fb}. For direct topology, the K

_{21}box is a bipolar or FET transistor; for feedback topology, it is a bipolar or FET operational amplifier. Due to the low noise requirements, any simple design is suitable. The direct amplifier topology (a) should be the best option, but it has many limitations [1]. The feedback amplifier topology, shown in Figure 6b, creates stability questions due to back feeding.

_{p}is the useful signal photocurrent, 2qI

_{0}is the photodiode noise current density, and R

_{p}and C

_{p}are internal photodiode resistance and capacitance, respectively. In the amplifier, R

_{a}and C

_{a}are amplifier input resistance and capacitance, respectively (previously included in Z

_{a}); 2qI

_{a}is the amplifier input noise current density; 4kT (Ψ/G) ω

^{2}C

_{in}

^{2}is the amplifier output noise current density calculated for the amplifier input; Gu

_{1}is the amplifier useful signal output current; and R

_{fb}and C

_{fb}are feedback (load) resistance and capacitance, respectively (previously included in Z

_{fb}). The stray capacitances (e.g., the PCB capacitance) are not introduced explicitly and may be included in C

_{fb}. Since all components are in parallel, the input impedance Z

_{in}may be expressed as:

_{0}and 2qI

_{a}are the shot noises of the photodiode and amplifier, respectively; 4kT/R

_{in}is the thermal noise of the total input resistance R

_{in}according to Equation (23) and 4kT (Ψ/G) ω

^{2}C

_{in}

^{2}is the amplifier output noise current density calculated for the amplifier input. This current density can also be expressed in shorter form as e

_{n}

^{2}ω

^{2}C

_{in}

^{2}. Since the shot and thermal noises are independent of frequency, they are white noise and may be substituted by element $2q{I}_{n}=2q{I}_{0}+2q{I}_{a}+4kT/{R}_{in}$. It is useful to unify Equation (24):

_{h}= 1.5 MHz), only the latter optimization is discussed.

_{n}

^{2}ω

^{2}C

_{in}

^{2}element, which becomes dominant. An amplifier with the lowest input voltage noise e

_{n}and the lowest input capacitance C

_{a}(determining C

_{in}) is necessary. The other conditions are as follows:

_{in}in Equation (28) is calculated from Equation (27). Equation (29) serves only to verify that the high cut-off frequency of PDA f

_{h}is fulfilled. If not, high frequency amplification loss should be compensated by a separate corrective circuit. An operational amplifier whose gain is big enough can cancel this loss and maintain the stability of the amplification up to frequency f

_{x}(see below).

_{n}and the lowest input capacitance C

_{a}in a discrete FET or the input FET operational amplifier. However, it has been proven that FET voltage noise e

_{n}and FET input capacitance C

_{a}are reciprocal [1]: the lower the voltage noise e

_{n}, the higher the input capacitance C

_{a}and vice versa. In other words, capacitive matching is needed:

_{21}high frequency roll-off. System stability is determined by the intersection angle between the blue and red lines. If the angle is equal to the red line and the horizontal axis angle, the system is stable; if it is greater, the system is unstable.

_{p}of the amplifier passive network is defined as:

_{T}is determined by the amplifier roll-off and the 0 dB level intersection, which is also known as the gain-bandwidth product of an operational amplifier (GBW). Finally, the amplifier roll-off and noise gain also intersect, whose frequency f

_{x}is:

_{x}should equal f

_{h}, which is the high cut-off frequency of the PDA. For the FSO receiver discussed, this is f

_{h}= f

_{x}= 1.5 MHz.

_{n}and capacitance C

_{a}operational amplifiers are listed in Table 2.

_{a}, input voltage noise e

_{n}, and transit frequency f

_{T}are typical values seen in operational amplifier datasheets. Total input capacitance C

_{in}is the sum of the BP104F photodiode typical capacitance C

_{p}= 16 pF (10 V reverse biased), stray capacitance C

_{s}= 3 pF and the operational amplifier input capacitance C

_{a}. The frequency f

_{x}was matched to the PDA’s high cut-off frequency f

_{h}(f

_{x}= f

_{h}= 1.5 MHz). Due to the high frequency PDA, high cut-off frequency f

_{h}is the same as the zero frequency f

_{z}, shown in Figure 7c, f

_{z}= f

_{h}= 1.5 MHz. The amplifier output noise current density e

_{n}

^{2}ω

_{x}

^{2}C

_{in}

^{2}was determined by calculation. The R

_{in}and I

_{a}conditions are derived from Equations (27) and (28), respectively. Similarly, pool frequency f

_{p}, feedback resistance R

_{fb}(which determines R

_{in}), and feedback capacitance C

_{fb}are derived from Equations (33), (31) and (32), respectively. Based on Table 2 and our calculations, we concluded the following: the OPA602BP (Texas Instruments, Dallas, TX, USA, 2002) device does not fulfil the capacitance matching conditions (C

_{a}= 1 pF, C

_{p}= 16 pF, C

_{a}<< C

_{p}); input voltage noise e

_{n}is the highest, with the small input capacitance C

_{a}being eliminated by the enormous C

_{p}capacitance. This was confirmed by the highest e

_{n}

^{2}ω

_{x}

^{2}C

_{in}

^{2}element. Therefore, the OPA602BP device is not suitable for the FSO receiver design. The OP27G (Analog Devices, Norwood, MA, USA, 2015) and the OPA627BP (Texas Instruments, Dallas, TX, USA, 2015) devices have much lower e

_{n}

^{2}ω

_{x}

^{2}C

_{in}

^{2}values. OP27G appears to be the best for the application. For a stable feedback loop, the resistance R

_{fb}must be lower than the required minimum resistance R

_{in}, R

_{fb}< R

_{in}, which destroys the good noise parameters of an operational amplifier. Neither OPA602BP nor OP27G meet with the PDA’s requirements. The opposite relation R

_{fb}> R

_{in}is only provided by the OPA627BP device, in which the e

_{n}

^{2}ω

_{x}

^{2}C

_{in}

^{2}value is three times higher and a high resistance R

_{fb}keeps thermal noise low, leading to high stage transimpedance. Therefore, the OPA627BP is the only device suitable for the low noise PDA requirements.

_{fb}keeps the amplifier feedback stable, bending the noise gain NG at the zero frequency f

_{z}(Figure 7). It is also responsible for the shape of the PDA closed loop amplitude frequency characteristic around the high cut-off frequency point f

_{h}. No overshoot should exist in the frequency characteristic for a well-compensated PDA. If this is the case, the frequency characteristic is closer to the first order declining asymptote from the left, and high frequency noises are kept to their minimum. All other cases are undesirable. Even a small overshoot brings the frequency characteristic closer toward the first order declining asymptote from the right, and the PDA output is loaded with excessively high frequency noise. The capacitance C

_{fb}is therefore usually set with a variable capacitor according to the frequency or time domain measurements.

_{fb}has been set, neither photodiode reverse bias nor any other items can be changed.

_{a}and the dark current I

_{0}, as in the condition in Equation (19). With the maximal allowed amplifier current I

_{a}= 5.47 μA, the condition in Equation (19) is fulfilled under incandescent lamp illumination. Under daylight, the currents are rather the same, which is still acceptable.

## 3. FSO Receiver Assembly

_{ref}. The optimal gain value of the VGA is set according to the obtained voltage difference V

_{diff}.

#### Detailed Circuit Diagram of the FSO Receiver

_{fb}= 2.53 pF and a non-zero capacitance of 1.5 pF in the variable capacitor. At the left of Figure 9 are the voltage reference MC1404U10 (ON Semiconductor, Phoenix, AZ, USA, 2006) (IC7) and reference voltage inverter OP27 (IC8). Both are low noise devices.

## 4. FSO Receiver Measurements

^{−1}data stream and x

^{23}+ x

^{5}+ 1 (n = 23) pseudorandom sequence. The calculations between the BER and the SNR parameters were provided by the “erfc” Excel function according to the following definition [33,34]:

^{−10}and 10

^{−11}, the time needed for measurements would be huge, reaching around 140 h (almost one week). The total measurement time would have to be extended at least to one month because several measurements are required.

^{7}+ x + 1 (n = 7) pseudorandom sequence or with the only one word (16 bits) repeated over and over again. However, the measurement was still unstable and synchronization was frequently lost. Even if the synchronization succeeded, the measured values would not be accurate due to the short data string and the presence of harmonic components in the spectrum of such testing signal. For the above reasons, two optical routes had to be used for measurement.

^{−3}. The SNR difference between only one and all six actively operating photodetectors was 6.41 dB, which is close to the theoretical value of 4.8 dB calculated above. As a reminder, the theoretical value of 4.8 dB was calculated as the SNR difference between only one and all six operating photodetectors with uniform signal coverage. In other words, this calculated difference covered only photodetector inherent noise suppression. The better result of 6.41 dB appears to be an inaccuracy arising from the unequal short and long optical routes. However, useful signal and noise processing differs, and no tool exists to precisely handle noise. Adding signals from the photodetectors means decreasing instead of increasing peak-to-peak total noise values. The measured result of 6.41 dB encourages more investigation with perhaps the same or more photodetector receivers.

^{−8}at ray trajectories equal to or longer than 6 m with a data stream as high as 2 Mbit·s

^{−1}.

_{ref}, comparator, variable gain amplifier, and its control voltage V

_{diff}, can be helpful in describing this. Adding noise increases its resultant power according to RMS. Peak-to-peak noise values are described by the noise power distribution function. The higher the noise RMS value, the higher the peak-to-peak values. However, we observed a difference in the photodetectors adding noise, which can be seen in Table 4 and from the graph in Figure 14.

_{diff}reduce VGA gain and vice versa. This means that an empty signal channel makes the V

_{diff}lowest and the VGA gain highest; similarly, a full signal channel makes the V

_{diff}highest and VGA gain lowest. The channel may be filled with a useful signal or with noise. According to Table 4, increasing the number of operating photodetectors (without a useful signal) cleared the channel, which was an unexpected but positive result.

## 5. Conclusions

^{−1}data stream.

^{−8}with FOVs highly overlapped. By contrast, the operation of only three photodetectors failed, with a high BER value of 4.79 × 10

^{−5}for the same configuration and FOVs only slightly overlapped.

## Author Contributions

## Funding

## Conflicts of Interest

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**Figure 1.**The truncated pyramid holder used for fixing six photodetectors PD

_{1}to PD

_{6}. Points A, B, C, and D, and S1, S2, and S mark the boundary and midpoints of one segment. The vector $\overline{n}$ means the normal vector of PD

_{1}photodetector active area. PD: photodetectors.

**Figure 3.**Shifts and rotations of square ABCD. (

**a**) Rotating the square on the x-axis, (

**b**) shifting the square along the y-axis, (

**c**) shifting the square along the z-axis, (

**d**) rotating the square on the z-axis.

**Figure 4.**Photodetector surface screening for segments SA, SB, SC, and SD over the φ

_{z}angle for (

**a**) the pyramid edge oriented toward the illumination source and (

**b**) the pyramid wall oriented toward the illumination source.

**Figure 6.**(

**a**) PDA direct topology; (

**b**) PDA feedback topology; (

**c**) PDA noise model including photodiode and amplifier intrinsic elements and discrete passive elements needed for operation.

**Figure 7.**Stability investigation of (

**a**) uncompensated, (

**b**) low frequency; and (

**c**) high frequency PDA using Bode plots.

**Figure 11.**(

**a**) Optical transmitter equipped with power LED VSMY99445 (Vishay, Malvern, PA, USA, 2015); (

**b**) assembled FSO receiver.

**Figure 12.**Synthesys BA400 BERT (SyntheSys Research, Menlo Park, CA, USA, 1992), PE codec, level and cabling translators, and Krohn-Hite 3202 band-pass filter (Krohn-Hite, Brockton, MA, USA, 1972) during PE spectrum evaluation.

**Table 1.**Darkness generation current I

_{dg}and total dark current I

_{0}= I

_{dg}+ I

_{pb}measurement results.

Photodiode | I_{dg}(Dark Room) | I_{0}(Incandescent Bulbs 200 lx) | I_{0}(LED Lamps 400 lx) | I_{0}(Daylight 550 lx *) |
---|---|---|---|---|

BPW34 | 62 pA | 26.6 μA | 4.18 μA | 19.3 μA |

BP104F | 46.9 pA | 20.8 μA | 132.3 nA | 7.09 μA |

Type | C_{a}[pF] | C_{in}[pF] | $\overline{{\mathit{e}}_{\mathit{n}}}$ $\left[\mathbf{n}\mathbf{V}\mathbf{\xb7}\mathbf{H}{\mathbf{z}}^{-\frac{1}{2}}\right]$ | f_{x}[MHz] | $\overline{{\mathit{e}}_{\mathit{n}}^{2}}{\mathit{\omega}}_{\mathit{x}}^{2}{\mathit{C}}_{\mathit{i}\mathit{n}}^{2}$ | R_{in}[kΩ] | I_{a}[μA] | f_{T}[MHz] | f_{p}[kHz] | R_{fb}[kΩ] | C_{fb}[pF] |
---|---|---|---|---|---|---|---|---|---|---|---|

OP27G | 8 | 27 | 3 | 1.5 | 0.583 × 10^{−24} | 28.4 | 1.82 | 8 | 281.3 | 20.96 | 5.06 |

OPA602BP | 1 | 20 | 13 | 1.5 | 6.005 × 10^{−24} | 2.76 | 18.7 | 6.5 | 346.2 | 22.99 | 4.62 |

OPA627BP | 8 | 27 | 5.2 | 1.5 | 1.751 × 10^{−24} | 9.46 | 5.47 | 16 | 140.6 | 41.92 | 2.53 |

Active Photodetector Count | 1 | 2 | 3 | 6 |
---|---|---|---|---|

Measured mean BER value [/] (short optical route) | 4.25 × 10^{−5} | 2.49 × 10^{−5} | 1.66 × 10^{−9} | / |

Corresponding SNR value [dB] (short optical route) | 17.91 | 18.19 | 21.45 | / |

Measured mean BER value [/] (long optical route) | / | / | 4.79 × 10^{−5} | 2.72 × 10^{−8} |

Corresponding SNR value [dB] (long optical route) | / | / | 17.85 | 20.72 |

Calculated SNR value [dB] related to the long optical route | 14.31 | 14.59 | 17.85 | 20.72 |

Operating Photodetector Count | 1 | 2 | 3 | 6 |
---|---|---|---|---|

V_{diff} voltage controlling VGA [V] | 1.002 | 0.935 | 0.896 | 0.867 |

© 2019 by the authors. Licensee MDPI, Basel, Switzerland. This article is an open access article distributed under the terms and conditions of the Creative Commons Attribution (CC BY) license (http://creativecommons.org/licenses/by/4.0/).

## Share and Cite

**MDPI and ACS Style**

Witas, K.; Nedoma, J.
Free Space Optic Receiver with Strongly Overlapped Photodetectors’ Field of View. *Appl. Sci.* **2019**, *9*, 343.
https://doi.org/10.3390/app9020343

**AMA Style**

Witas K, Nedoma J.
Free Space Optic Receiver with Strongly Overlapped Photodetectors’ Field of View. *Applied Sciences*. 2019; 9(2):343.
https://doi.org/10.3390/app9020343

**Chicago/Turabian Style**

Witas, Karel, and Jan Nedoma.
2019. "Free Space Optic Receiver with Strongly Overlapped Photodetectors’ Field of View" *Applied Sciences* 9, no. 2: 343.
https://doi.org/10.3390/app9020343