# A High Step-Down Interleaved Buck Converter with Active-Clamp Circuits for Wind Turbines

^{1}

^{2}

^{*}

## Abstract

**:**

## 1. Introduction

_{dc}or 12 V

_{dc}) and high output current to load. Therefore, a conventional interleaved buck converter (IBC) as shown in Figure 1 is widely adopted, because it has a simple structure, high output current density and low output current ripple. However, in high step-down voltage applications, it suffers from extremely short duty ratio and high component stresses, resulting in low conversion efficiency [5,6,7,8]. To alleviate these limitations, the IBCC is proposed, as shown in Figure 2 (where n

_{1}and n

_{11}are primary and secondary winding turns of coupled inductor L

_{1}, n

_{2}and n

_{22}are primary and secondary winding turns of coupled inductor L

_{2}). A coupled inductor has a simple winding structure, and it can extend duty ratio and reduce peak primary winding current. Thus, a coupled inductor based on converter is relatively attractive. Although the IBCC can yield high step-down voltage ratios, its leakage inductance (L

_{k}

_{1}and L

_{k}

_{2}) of coupled inductors not only increases voltage stresses on the active switches but induces significant switching losses too. To overcome these problems, the IBCC with passive-clamp circuits, as shown in Figure 3, it can recover the leakage energy and reduce voltage stresses of active switches, but the active switches are still operated in a hard-switching manner during turn-on transition [9,10,11,12].

_{1}and M

_{2}) and auxiliary switches (M

_{11}and M

_{22}), the extra resonant inductors (L

_{r}

_{1}and L

_{r}

_{2}) are usually required. In particular, the resonant inductors can also limit the rate of decrease of the free-wheeling diode currents at turn-off transition, reducing reverse-recovery losses a lot and further improving conversion efficiency [13,14,15,16,17,18].

## 2. Selection of Active-Clamp Circuit

_{cr}

_{(flyback)}can be expressed as:

_{i}is input voltage and D is duty ratio of main switch M

_{1}. Similarly, In Figure 6, the voltage of the clamp capacitor V

_{cr}

_{(boost)}can be expressed as:

## 3. Operational Principle

_{1}, L

_{2}, main switches M

_{1}, M

_{2}, auxiliary switches M

_{11}, M

_{22}, resonant inductors L

_{r}

_{1}, L

_{r}

_{2}, clamp capacitors C

_{r}

_{1}, C

_{r}

_{2}and free-wheeling diodes D

_{1}, D

_{2}. In order to analyze the ZVS feature of the main and auxiliary switches (M

_{1}, M

_{2}, M

_{11}and M

_{22}), their stray capacitors (C

_{M}

_{1}, C

_{M}

_{2}, C

_{M}

_{11}and C

_{M}

_{22}) will be considered at the steady-state operation of the circuit, as shown in Figure 7. In Figure 7, each set of the coupled inductors can be treated as a transformer with two magnetizing inductors. Its turns ratio is defined as:

- (1)
- clamp capacitors C
_{r}_{1}, C_{r}_{2}and output filter capacitor C_{o}are large enough so that the voltages across them are constant over a switching period, - (2)
- winding and turns of coupled inductors (L
_{1}and L_{2}) are identical, and - (3)
- all semiconductor components are ideal, except the MOSFETs (M
_{1}, M_{2}, M_{11}and M_{22}).

_{1}and auxiliary switch M

_{22}are turned on. Inductor current i

_{L}

_{1}flowing through the path V

_{i}-L

_{11}-L

_{r}

_{1}-M

_{1}-L

_{1}to the load is linearly increased. Clamping capacitor C

_{r}

_{2}begins releasing its stored energy by coupled-inductor L

_{2}and L

_{22}transferring to the load.

_{1}, main switch M

_{1}is turned off, resonant inductor L

_{r}

_{1}releases energy to stray capacitance C

_{M}

_{1}of M

_{1}and stray capacitance C

_{M}

_{11}of M

_{11}with a resonant manner. When time t = t

_{2}, stray capacitance C

_{M}

_{1}of M

_{1}will be charged toward to (V

_{i}+ nV

_{o}) while stray capacitance C

_{M}

_{11}of M

_{11}will be discharged down to zero. To achieve a ZVS feature for auxiliary switch M

_{11}, the energy stored in resonant inductor L

_{r}

_{1}should satisfy the following inequality:

**Figure 9.**Equivalent circuit modes of the IBCC with active-clamp circuits operating over one switching cycle. (

**a**) Mode 1 [t

_{0}≤ t < t

_{1}]; (

**b**) Mode 2 [t

_{1}≤ t < t

_{2}]; (

**c**) Mode 3 [t

_{2}≤ t < t

_{3}]; (

**d**) Mode 4 [t

_{3}≤ t < t

_{4}]; (

**e**) Mode 5 [t

_{4}≤ t < t

_{5}]; (

**f**) Mode 6 [t

_{5}≤ t < t

_{6}]; (

**g**) Mode 7 [t

_{6}≤ t < t

_{7}]; (

**h**) Mode 7 [t

_{7}≤ t < t

_{8}].

_{2}, voltage V

_{DS}

_{11}of M

_{11}is dropped to zero and V

_{DS}

_{1}of M

_{1}is reached to V

_{i}+ nV

_{o}. Main switch M

_{1}is turned off and free-wheeling diode D

_{1}is turned on. Current i

_{DS}

_{11}forces the body diode D

_{M}

_{11}of M

_{11}conducting to create a ZVS operation for M

_{11}. In this mode, inductor current i

_{L}

_{1}begins decreased through free-wheeling diode D

_{1}to the load. The inductor currents i

_{L}

_{1}and i

_{L}

_{2}can be expressed as follows:

_{r}

_{1}is recycled to clamp capacitor C

_{r}

_{1}. Due to the clamp capacitance of C

_{r}

_{1}being large enough, voltage V

_{Cr}

_{1}will keep constant.

_{22}is turned off at time t

_{2}, resonant inductor L

_{r}

_{2}resonates with C

_{M}

_{2}and C

_{M}

_{22}. Stray capacitance C

_{M}

_{22}of M

_{22}will be charged toward to V

_{Cr}

_{2}+ [n/(1 + n)](V

_{i}− V

_{o}), while stray capacitance C

_{M}

_{2}of M

_{2}will be discharged down to zero. To achieve a ZVS feature for main switch M

_{2}, the energy trapped in resonant inductor L

_{r}

_{2}should satisfy the following inequality:

_{3}, auxiliary switch M

_{11}is turned on under ZVS condition. When voltage V

_{DS}

_{2}of M

_{2}is dropped to zero and V

_{DS}

_{22}of M

_{22}is reached to V

_{Cr}

_{2}+ [n/(1 + n)](V

_{i}− V

_{o}) , auxiliary switch M

_{22}is turned off and current i

_{DS}

_{2}forces the body diode D

_{M}

_{2}of M

_{2}conducting to create a ZVS operation for M

_{2}. In this mode, inductor current i

_{L}

_{1}and i

_{L}

_{2}are continuously decreased through free-wheeling diode D

_{1}and D

_{2}to the load.

_{4}, main switch M

_{2}is turned on under ZVS condition and free-wheeling diode D

_{2}is turned off. Inductor current i

_{L}

_{2}flowing through the path of V

_{i}-L

_{22}-L

_{r}

_{2}-M

_{2}-L

_{2}to the load is linearly increased, and inductor current i

_{L}

_{1}continuously flowing through the path of V

_{o}-D

_{1}-L

_{1}is linearly decreases, which can be expressed as follows:

_{5}, main switch M

_{2}is turned off, resonant inductor L

_{r}

_{2}releases energy to stray capacitance C

_{M}

_{2}of M

_{2}and stray capacitance C

_{M}

_{22}of M

_{22}with a resonant manner. When time t = t

_{6}, the stray capacitance C

_{M}

_{2}will be charged toward to (V

_{i}+ nV

_{o}), while stray capacitance C

_{M}

_{22}will be discharged to zero. To achieve a ZVS feature for main switch M

_{22}, the energy stored in resonant inductor L

_{r}

_{2}should satisfy the following inequality:

_{6}, voltage V

_{DS}

_{22}of M

_{22}is dropped to zero and voltage V

_{DS}

_{2}of M

_{2}is reached to (V

_{i}+ nV

_{o}), main switch M

_{2}is turned off and free-wheeling diode D

_{2}is conducted. Current i

_{DS}

_{22}forces the body diode D

_{M}

_{22}of M

_{22}conducting and to create a ZVS operation for auxiliary switch M

_{22}. In this mode, inductor current i

_{L}

_{2}begins decreased through free-wheeling diode D

_{2}to the load. The inductor currents i

_{L}

_{1}and i

_{L}

_{2}can be expressed as follows:

_{r}

_{2}is recycled to clamp capacitor C

_{r}

_{2}. Since the clamping capacitor of C

_{r}

_{2}is large enough, voltage V

_{cr}

_{2}will keep constant.

_{7}, voltage V

_{DS}

_{1}of M

_{1}is dropped to zero and voltage V

_{DS}

_{11}of M

_{1}is reached to (V

_{i}+ nV

_{o}), main switch M

_{11}is turned off. Current i

_{DS}

_{1}forces the body diode D

_{M}

_{1}of M

_{1}conducting and to create a ZVS operation for main switch M

_{1}. When main switch M

_{1}starts conducting again at the end of mode 8, the converter operation over one switching cycle is completed.

## 4. Feature Analysis

#### 4.1. Voltage Gain and Duty Ratio

_{1}or M

_{2}), respectively.

_{i}= 150–250 V

_{dc}and output voltage V

_{o}= 12 V

_{dc}are considered. From Equations (13) and (14), we can sketch a set of curves showing the relationship between duty ratio D and voltage gain of V

_{o}/V

_{i}for different values of turns ratio n, as illustrated in Figure 10.

#### 4.2. Voltage Stress of Free-Wheeling Diode and Active Switch

_{1}and D

_{2}) and active switches (M

_{1}and M

_{2}) can be derived as:

**Figure 11.**Plots of voltage stress versus turns ratio n of the coupled inductor. (

**a**) free-wheeling diodes (D

_{1}and D

_{2}); (

**b**) active switches (M

_{1}and M

_{2}).

**Figure 12.**Performance comparison between the IBCC and the IBC. (

**a**) Duty ratio; (

**b**) voltage stress of the free-wheeling diode.

## 5. Power Losses Estimation and Experimental Results

- Input voltage: 150–200 V
_{dc}; - Output voltage: 12 V
_{dc}; - Maximum output current:20 A;
- Switching frequency: 75 kHz.

#### 5.1. Design Considerations of Key Components

_{1}and D

_{2}. In order to accommodate variations of loads, a proper turns ratio n of the coupled inductors is needed. From Figure 10, we can obtain a proper coupled-inductor turns ratio n = 8. Its maximum duty ratio is D

_{max}≈ 0.41 under input voltage V

_{i}= 150 V

_{dc}and minimum duty ratio is D

_{min}≈ 0.34 under input voltage V

_{i}= 200 V

_{dc}.

#### 5.1.1. Design of the Coupled Inductors

_{max}≈ 0.41. The maximum current ripple of the coupled inductors is designed with i

_{L}

_{(ripple)}= i

_{L}

_{1(ripple)}= i

_{L}

_{2(ripple)}= 10 A, and coupled inductance L = L

_{1}= L

_{2}can be determined as:

_{s}is the switching frequency of the active switches. Coupled inductance L = 3 μH is selected in the design. When the turns ratio n = (n

_{1}+ n

_{2})/n

_{1}is equal to 8, the coupled inductance L

_{11}= L

_{22}can be determined as:

_{max}= 200 mT, maximum winding factor K

_{w}

_{(max)}= 0.4 and maximum current density J

_{max}= 400 A/cm

^{2}. Thus, the area product of the core can be determined as:

_{a}is the window area of the core, A

_{e}is the effective cross-section area of the core, P

_{o}is the output power of the converter and η denotes the efficiency. From TDK datasheets, we select a proper size of core ETD-39 (A

_{e}= 1.25 cm

^{2}, W

_{a}= 2.57 cm

^{2}, V

_{e}= 11.5 cm

^{3}and A

_{L}= 3150 nH/N

^{2}) to reduce winding current density and core temperature.

_{11}or n

_{22}) of the coupled inductor (L

_{11}or L

_{22}) can be determined as:

_{11}is chosen as 35 turns. From turns ratio of the coupled inductors, n = (n

_{1}+ n

_{11})/n

_{1}, the turns (n

_{1}or n

_{2}) of the coupled inductance (L

_{1}or L

_{2}) can be correspondingly determined as:

#### 5.1.2. Selection of Power Switches and Diodes

_{1}and D

_{2}is:

_{1}and M

_{2}are:

_{1}or M

_{2}is turned on, the maximum switch current i

_{DS}

_{(max)}can be given as:

_{1}or D

_{2}is conducting, the maximum diode current i

_{D}

_{(max)}can be given as

_{1}and M

_{2}involves a trade-off between conduction loss and switching loss. The selection of MOSFETs with low R

_{ds}

_{(on)}will reduce conduction loss, but it will result in high parasitic capacitance. Switches with lower R

_{ds}

_{(on)}also imply larger die size and higher cost. For this application, we can select the proper MOSFETs as the IRFP460, which provide high enough safety margins with a drain-source breakdown voltage of 500 V. Several important parameters of the IRFP460 are listed as follows:

_{DSS}= 500 V, R

_{ds}

_{(on)}= 0.27 Ω, C

_{oss}= 870 pF and I

_{D}= 20 A

_{1}and D

_{2}, a 40 A/60 V Schottky diode which has the lowest forward voltage drop can be employed. The Schottky diode manufactured by International Rectifier is 40CPQ60 with a maximum dc reverse voltage V

_{RRM}= 60 V and a forward voltage drop V

_{F}

_{(max)}= 0.47 V, which is a good choice for D

_{1}and D

_{2}, was chosen.

#### 5.1.3. Consideration of ZVS Condition

_{1}, M

_{2}, M

_{11}and M

_{22}) at turn-on transition, the IBCC is necessary to store enough energy in resonant inductor (L

_{r}

_{1}or L

_{r}

_{2}). Because the voltage across main switch (M

_{1}or M

_{2}) is larger than that across auxiliary switch (M

_{11}or M

_{22}). Thus, the ZVS condition for all of the active switches is determined as follows:

_{r}= L

_{r}

_{1}= L

_{r}

_{2}, the clamp voltage of clamp capacitor V

_{cr}= V

_{cr}

_{1}= V

_{cr}

_{2}and the stray capacitor of the active switch is expressed as C

_{M}

_{1}= C

_{M}

_{2}and C

_{M}

_{11}= C

_{M}

_{22}).

_{r}and output current I

_{o}for different values of input voltage V

_{i}, as illustrated in Figure 13. From Figure 13, a proper value of resonant inductor L

_{r}should be selected for achieving ZVS feature.

**Figure 13.**Plots of resonant inductor L

_{r}versus output current I

_{o}for different values of input voltage V

_{i}.

#### 5.2. Power Losses Estimation

#### 5.2.1. Main Switches (M_{1} and M_{2})

_{DS}and channel resistance R

_{ds}

_{(on)}. In the proposed converter, the MOSFETs used are IRFP460, and their channel resistance R

_{ds}

_{(on)}= 0.27 Ω. In the proposed converter, the maximum duty ratio D

_{max}= 0.41 and switch current i

_{DS}

_{(max)}= 9.1 A. Thus, total power losses of the main switches can be determined as:

#### 5.2.2. Auxiliary Switches (M_{11} and M_{22})

#### 5.2.3. Free-Wheeling Diodes (D_{1} and D_{2})

_{F}= 0.47 V. Thus, total conduction losses of the free-wheeling diodes can be determined as:

#### 5.2.4. Coupled Inductors (L_{1} and L_{2})

_{max}= 200 mT. Their total winding turns are n

_{total}= (n

_{1}+ n

_{11}) = 40, primary winding turns are n

_{p}= n

_{1}= 5, and the turns ratio is n = n

_{total}/n

_{p}= 8. Thus, from Figure 15, we can obtain the core losses per volume, P

_{Coup}

_{(cv)}= 0.2 W/cm

^{3}at 60 °C. The core loss will be:

_{Coup}= 28 mΩ, and the copper losses can be estimated as:

_{i}= 150 V

_{dc}, and full-load condition is:

Components | Power Loss Calculation | Power Losses |
---|---|---|

Main switch (M_{1} and M_{2}) | ${P}_{total}^{main}=2[\frac{1}{3}{D}_{\mathrm{max}}{({i}_{DS(\mathrm{max})})}^{2}{R}_{ds(on)}]$ | 6 W |

Auxiliary switch (M_{11} and M_{22}) | ${P}_{total}^{aux}=2[\frac{1}{3}(1-{D}_{\mathrm{max}}){({i}_{DS(\mathrm{max})})}^{2}{R}_{ds(on)}]$ | 8.8 W |

Freewheel diode (D_{1} and D_{2}) | ${P}_{total}^{diode}=2[\frac{{I}_{o}}{2}{V}_{F}(1-{D}_{\mathrm{max}})]$ | 4.8 W |

Coupled inductor (L_{1} and L_{2}) | ${P}_{total}^{coup}=2({P}_{core}^{coup}+{P}_{copper}^{coup})$ | 10.2 W |

Total power losses | ${P}_{Total}^{Loss}={P}_{total}^{main}+{P}_{total}^{aux}+{P}_{total}^{coup}$ | 29.8 W |

Efficiency estimation | $\eta \%=\frac{{P}_{out}}{{P}_{out}+{P}_{Total}^{Loss}}$ | 89% |

#### 5.3. Experimental Results

_{1}or M

_{2}), from which it can be seen that the main switch has ZVS features during the turn-on transition. Figure 17 shows simulated and experimental voltage and current waveforms of the auxiliary switch (M

_{11}or M

_{22}). It can be also seen that the auxiliary switch has ZVS feature during turn-on transition. Figure 18 shows simulated and experimental inductor current waveforms of coupled inductor (i

_{L}

_{1}and i

_{L}

_{2}). Figure 19 shows measured current and voltage waveforms of the free-wheeling diode (D

_{1}or D

_{2}), from which it can be seen that the proposed IBCC with active-clamp circuits can limit the rate of decrease of the free-wheeling diode current, reducing reverse-recovery losses a lot. Figure 20 shows measured output voltage and current waveforms. Figure 21 shows efficiency measurements of the IBCC with active-clamp circuits, from which it can be seen that the maximum efficiency can reach as high as 91%. It can increase efficiency about 2% over the IBCC with passive-clamp circuits (as shown in Figure 3). The reason behind is that the active switches of IBCC with passive-clamp circuits are still operated in a hard-switching manner during turn-on transition.

**Figure 16.**Voltage and current waveforms of main switches (M

_{1}or M

_{2}). (

**a**) simulated results (V

_{DS}

_{1}: 200 V/div; i

_{DS}

_{1}: 10 A/div; Time: 5 μs/div); (

**b**) experimental results (V

_{DS}

_{1}: 200 V/div; i

_{DS}

_{1}: 10 A/div; Time: 5 μs/div); (

**c**) its expanded waveforms (V

_{DS}

_{1}: 100 V/div; i

_{DS}

_{1}: 5 A/div; Time: 0.5 μs/div).

**Figure 17.**Voltage and current waveforms of auxiliary switch (M

_{11}or M

_{22}). (

**a**) simulated results (V

_{DS}

_{1}

_{1}: 200 V/div; i

_{DS}

_{1}

_{1:}10 A/div; Time: 5 μs/div); (

**b**) experimental results (V

_{DS}

_{1}

_{1}: 200 V/div; i

_{DS}

_{1}

_{1}: 10 A/div; Time: 5 μs/div); (

**c**) its expanded waveforms (V

_{DS}

_{1}

_{1}: 100 V/div; i

_{DS}

_{1}

_{1}: 10 A/div; Time: 0.5 μs/div).

**Figure 18.**Waveforms of inductor current (L

_{1}and L

_{2}). (

**a**) simulated results (i

_{L}

_{1}: 10 A/div; i

_{L}

_{2}: 10 A/div; Time: 5 μs/div); (

**b**) experimental results (i

_{L}

_{1}: 10 A/div; i

_{L}

_{2}: 10 A/div; Time: 5 μs/div).

**Figure 19.**Measured voltage and current waveforms of free-wheeling diode (D

_{1}or D

_{2}; V

_{D}

_{1}: 20 V/div; i

_{D}

_{1}: 10 A/div; Time: 200 ns/div).

**Figure 20.**Measured output voltage and current waveforms (V

_{o}: 5 V/div; I

_{o}: 5 A/div; time: 500 ms/div).

**Figure 21.**Plots of efficiency versus output current for the IBCC with active-clamp circuits and without active-clamp circuits at input voltage 150 V

_{dc}.

## 6. Conclusions

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## Share and Cite

**MDPI and ACS Style**

Tsai, C.-T.; Shen, C.-L.
A High Step-Down Interleaved Buck Converter with Active-Clamp Circuits for Wind Turbines. *Energies* **2012**, *5*, 5150-5170.
https://doi.org/10.3390/en5125150

**AMA Style**

Tsai C-T, Shen C-L.
A High Step-Down Interleaved Buck Converter with Active-Clamp Circuits for Wind Turbines. *Energies*. 2012; 5(12):5150-5170.
https://doi.org/10.3390/en5125150

**Chicago/Turabian Style**

Tsai, Cheng-Tao, and Chih-Lung Shen.
2012. "A High Step-Down Interleaved Buck Converter with Active-Clamp Circuits for Wind Turbines" *Energies* 5, no. 12: 5150-5170.
https://doi.org/10.3390/en5125150