A High Step-Down Interleaved Buck Converter with Active-Clamp Circuits for Wind Turbines

In this paper, a high step-down interleaved buck coupled-inductor converter (IBCC) with active-clamp circuits for wind energy conversion has been studied. In high step-down voltage applications, an IBCC can extend duty ratio and reduce voltage stresses on active switches. In order to reduce switching losses of active switches to improve conversion efficiency, a IBCC with soft-switching techniques is usually required. Compared with passive-clamp circuits, the IBCC with active-clamp circuits have lower switching losses and minimum ringing voltage of the active switches. Thus, the proposed IBCC with active-clamp circuits for wind energy conversion can significantly increase conversion efficiency. Finally, a 240 W prototype of the proposed IBCC with active-clamp circuits was built and implemented. Experimental results have shown that efficiency can reach as high as 91%. The proposed IBCC with active-clamp circuits is presented in high step-down voltage applications to verify the performance and the feasibility for energy conversion of wind turbines.


Introduction
Limited fossil energy and serious greenhouse effect have forced most engineers to research renewable energy sources.The typical renewable energy sources include solar, wind and geothermal energy, which have the features of cleanliness, freedom from maintenance and abundance [1].Therefore, the development of renewable and clean energy sources to substitute for fossil fuels has been an important topic.Currently, wind is one of most widely utilized renewable energies.The wind turbine technology has been undergoing a dramatic development and now is the world's fastest growing energy form [2].However, due to the instability and intermittent characteristics of wind energy, it cannot provide a constant or stable power output.Thus, the power processor usually adopts a dc/dc converter as its energy processing system [2][3][4].
In wind energy conversion, most dc/dc converters are usually provided a low output voltage (24 V dc or 12 V dc ) and high output current to load.Therefore, a conventional interleaved buck converter (IBC) as shown in Figure 1 is widely adopted, because it has a simple structure, high output current density and low output current ripple.However, in high step-down voltage applications, it suffers from extremely short duty ratio and high component stresses, resulting in low conversion efficiency [5][6][7][8].To alleviate these limitations, the IBCC is proposed, as shown in Figure 2 (where n 1 and n 11 are primary and secondary winding turns of coupled inductor L 1 , n 2 and n 22 are primary and secondary winding turns of coupled inductor L 2 ).A coupled inductor has a simple winding structure, and it can extend duty ratio and reduce peak primary winding current.Thus, a coupled inductor based on converter is relatively attractive.Although the IBCC can yield high step-down voltage ratios, its leakage inductance (L k1 and L k2 ) of coupled inductors not only increases voltage stresses on the active switches but induces significant switching losses too.To overcome these problems, the IBCC with passive-clamp circuits, as shown in Figure 3, it can recover the leakage energy and reduce voltage stresses of active switches, but the active switches are still operated in a hard-switching manner during turn-on transition [9][10][11][12].Therefore, the conversion efficiency can be improved significantly.In Figure 4, to achieve a zero-voltage switching (ZVS) feature at turn-on transition for both main switches (M 1 and M 2 ) and auxiliary switches (M 11 and M 22 ), the extra resonant inductors (L r1 and L r2 ) are usually required.In particular, the resonant inductors can also limit the rate of decrease of the free-wheeling diode currents at turn-off transition, reducing reverse-recovery losses a lot and further improving conversion efficiency [13][14][15][16][17][18].

Selection of Active-Clamp Circuit
The active-clamp circuits discussed in this study can be selected as two circuits, in which one is a flyback-type clamp circuit, as shown in Figure 5, and the other is a boost-type clamp circuit, as shown in Figure 6.In Figure 5, by volt-second balance law, the voltage of the clamp capacitor V cr(flyback) can be expressed as: where V i is input voltage and D is duty ratio of main switch M 1 .Similarly, In Figure 6, the voltage of the clamp capacitor V cr(boost) can be expressed as: From Equations ( 1) and ( 2), we can see that voltage stress of the flyback-type clamp circuit is less than that of the boost-type one, making the flyback-type clamp circuit more attractive.In this study, only the flyback-type clamp is considered due to its obvious advantages.Thus, the IBCC with flyback-type clamp circuits, as shown in Figure 4, is proposed.

Operational Principle
The proposed IBCC (Figure 4) mainly includes two sets of coupled inductors L 1 , L 2 , main switches M 1 , M 2 , auxiliary switches M 11 , M 22 , resonant inductors L r1 , L r2 , clamp capacitors C r1 , C r2 and free-wheeling diodes D 1 , D 2 .In order to analyze the ZVS feature of the main and auxiliary switches (M 1 , M 2 , M 11 and M 22 ), their stray capacitors (C M1 , C M2 , C M11 and C M22 ) will be considered at the steady-state operation of the circuit, as shown in Figure 7.In Figure 7, each set of the coupled inductors can be treated as a transformer with two magnetizing inductors.Its turns ratio is defined as:  Under continuous inductor current operation, eight major operating modes are identified over one switching cycle.Figure 8 shows conceptual voltage and current waveforms of its key components and Figure 9 shows equivalent circuits of its operational modes: Mode 1 [Figure 9(a), t 0 ≤ t < t 1 ]: In this mode, main switch M 1 and auxiliary switch M 22 are turned on.Inductor current i L1 flowing through the path V i -L 11 -L r1 -M 1 -L 1 to the load is linearly increased.Clamping capacitor C r2 begins releasing its stored energy by coupled-inductor L 2 and L 22 transferring to the load.
Mode 2 [Figure 9(b), t 1 ≤ t < t 2 ]: At time t = t 1 , main switch M 1 is turned off, resonant inductor L r1 releases energy to stray capacitance C M1 of M 1 and stray capacitance C M11 of M 11 with a resonant manner.When time t = t 2 , stray capacitance C M1 of M 1 will be charged toward to (V i + nV o ) while stray capacitance C M11 of M 11 will be discharged down to zero.To achieve a ZVS feature for auxiliary switch M 11 , the energy stored in resonant inductor L r1 should satisfy the following inequality: Main switch M 1 is turned off and free-wheeling diode D 1 is turned on.Current i DS11 forces the body diode D M11 of M 11 conducting to create a ZVS operation for M 11 .In this mode, inductor current i L1 begins decreased through free-wheeling diode D 1 to the load.The inductor currents i L1 and i L2 can be expressed as follows: and : During this interval, the energy trapped in the resonant inductor L r1 is recycled to clamp capacitor C r1 .Due to the clamp capacitance of C r1 being large enough, voltage V Cr1 will keep constant.
When auxiliary switch M 22 is turned off at time t 2 , resonant inductor L r2 resonates with C M2 and C M22 .Stray capacitance C M22 of M 22 will be charged toward to will be discharged down to zero.To achieve a ZVS feature for main switch M 2 , the energy trapped in resonant inductor L r2 should satisfy the following inequality: auxiliary switch M 22 is turned off and current i DS2 forces the body diode D M2 of M 2 conducting to create a ZVS operation for M 2 .In this mode, inductor current i L1 and i L2 are continuously decreased through free-wheeling diode D 1 and D 2 to the load.
Mode 5 [Figure 9(e), t 4 ≤ t < t 5 ]: At time t = t 4 , main switch M 2 is turned on under ZVS condition and free-wheeling diode D 2 is turned off.Inductor current i L2 flowing through the path of V i -L 22 -L r2 -M 2 -L 2 to the load is linearly increased, and inductor current i L1 continuously flowing through the path of V o -D 1 -L 1 is linearly decreases, which can be expressed as follows: and: Mode 6 [Figure 9(f), t 5 ≤ t < t 6 ]: At time t = t 5 , main switch M 2 is turned off, resonant inductor L r2 releases energy to stray capacitance C M2 of M 2 and stray capacitance C M22 of M 22 with a resonant manner.When time t = t 6 , the stray capacitance C M2 will be charged toward to (V i + nV o ), while stray capacitance C M22 will be discharged to zero.To achieve a ZVS feature for main switch M 22 , the energy stored in resonant inductor L r2 should satisfy the following inequality: Mode 7 [Figure 9(g), t 6 ≤ t < t 7 ]: At time t = t 6 , voltage V DS22 of M 22 is dropped to zero and voltage V DS2 of M 2 is reached to (V i + nV o ), main switch M 2 is turned off and free-wheeling diode D 2 is conducted.Current i DS22 forces the body diode D M22 of M 22 conducting and to create a ZVS operation for auxiliary switch M 22 .In this mode, inductor current i L2 begins decreased through free-wheeling diode D 2 to the load.The inductor currents i L1 and i L2 can be expressed as follows: and: The energy trapped in the resonant inductor L r2 is recycled to clamp capacitor C r2 .Since the clamping capacitor of C r2 is large enough, voltage V cr2 will keep constant.
Mode 8 [Figure 9(h), t 7 ≤ t < t 8 ]: main switch M 11 is turned off.Current i DS1 forces the body diode D M1 of M 1 conducting and to create a ZVS operation for main switch M 1 .When main switch M 1 starts conducting again at the end of mode 8, the converter operation over one switching cycle is completed.

Feature Analysis
The proposed IBCC with active-clamp circuits can extend duty ratio of the active switches and reduce component stress.This section describes the feature analysis and efficiency estimation for the proposed IBCC.The feature analysis includes voltage gain, duty ratio, and voltage stress of free-wheeling diode:

Voltage Gain and Duty Ratio
From the key waveforms of the converter shown in Figure 8 and by applying the volt-second balance law, the voltage gain and duty ratio can be derived as: (13) and: where D is the duty ratio of the main switch (M 1 or M 2 ), respectively.
For example, input voltage V i = 150-250 V dc and output voltage V o = 12 V dc are considered.From Equations ( 13) and ( 14), we can sketch a set of curves showing the relationship between duty ratio D and voltage gain of V o /V i for different values of turns ratio n, as illustrated in Figure 10.

Voltage Stress of Free-Wheeling Diode and Active Switch
According to description of Mode 2, the voltage stress of the free-wheeling diodes (D 1 and D 2 ) and active switches (M 1 and M 2 ) can be derived as: ( 1) From Equations ( 15) and ( 16), we can sketch a set of curves showing the free-wheeling diode voltage stress versus turns ratio n under different input voltages, as shown in Figure 11.To objectively judge the merits of the proposed converter, performance comparison between the proposed converter and the IBC is shown in Figure 12.From these plots, it can be seen that the proposed IBCC yields higher duty ratio and lower diode voltage stress over the IBC.

Power Losses Estimation and Experimental Results
To verify the performance of the proposed IBCC with active-clamp circuits is higher than that of the IBCC with passive-clamp circuits, a 240 W prototype of the proposed converter was designed and built, as shown in Figure 7.Its specifications are listed as follows:  Input voltage: 150-200 V dc ;  Output voltage: 12 V dc ;  Maximum output current:20 A;  Switching frequency: 75 kHz.
As followed are the design considerations, power losses estimation and experimental results for the proposed IBCC with active-clamp circuits.

Design Considerations of Key Components
A larger duty ratio D corresponds to a larger coupled-inductor turns ratio n, which results in a lower voltage stress on free-wheeling diodes D 1 and D 2 .In order to accommodate variations of loads, a proper turns ratio n of the coupled inductors is needed.From Figure 10, we can obtain a proper coupled-inductor turns ratio n = 8.Its maximum duty ratio is D max ≈ 0.41 under input voltage V i = 150 V dc and minimum duty ratio is D min ≈ 0.34 under input voltage V i = 200 V dc .

Design of the Coupled Inductors
Once the coupled-inductor turns ratio n = 8 selected, the maximum duty ratio can be determined as D max ≈ 0.41.The maximum current ripple of the coupled inductors is designed with i L(ripple) = i L1(ripple) = i L2(ripple) = 10 A, and coupled inductance L = L 1 = L 2 can be determined as: where f s is the switching frequency of the active switches.Coupled inductance L = 3 μH is selected in the design.When the turns ratio n = (n 1 + n 2 )/n 1 is equal to 8, the coupled inductance L 11 = L 22 can be determined as: ( 1) From TDK datasheets, we choose an optimum ferrite material PC40 and maximum flux B max = 200 mT, maximum winding factor K w(max) = 0.4 and maximum current density J max = 400 A/cm 2 .Thus, the area product of the core can be determined as: where W a is the window area of the core, A e is the effective cross-section area of the core, P o is the output power of the converter and η denotes the efficiency.From TDK datasheets, we select a proper size of core ETD-39 (A e = 1.25 cm 2 , W a = 2.57 cm 2 , V e = 11.5 cm 3 In this design, n 11 is chosen as 35 turns.From turns ratio of the coupled inductors, n = (n 1 + n 11 )/n 1 , the turns (n 1 or n 2 ) of the coupled inductance (L 1 or L 2 ) can be correspondingly determined as:

Selection of Power Switches and Diodes
According to Equations ( 15) and ( 16), the maximum voltage stresses imposed on both free-wheeling diodes D 1 and D 2 is: and both active switches M 1 and M 2 are: When active switch M 1 or M 2 is turned on, the maximum switch current i DS(max) can be given as: when free-wheeling diode D 1 or D 2 is conducting, the maximum diode current i D(max) can be given as Selection of switching devices M 1 and M 2 involves a trade-off between conduction loss and switching loss.The selection of MOSFETs with low R ds(on) will reduce conduction loss, but it will result in high parasitic capacitance.Switches with lower R ds(on) also imply larger die size and higher cost.For this application, we can select the proper MOSFETs as the IRFP460, which provide high enough safety margins with a drain-source breakdown voltage of 500 V. Several important parameters of the IRFP460 are listed as follows: V DSS = 500 V, R ds(on) = 0.27 Ω, C oss = 870 pF and I D = 20 A According to (22)~(25), with the selection of free-wheeling diodes D 1 and D 2 , a 40 A/60 V Schottky diode which has the lowest forward voltage drop can be employed.The Schottky diode manufactured by International Rectifier is 40CPQ60 with a maximum dc reverse voltage V RRM = 60 V and a forward voltage drop V F(max) = 0.47 V, which is a good choice for D 1 and D 2 , was chosen.

Consideration of ZVS Condition
To achieve the ZVS feature for all of the active switches (M 1 , M 2 , M 11 and M 22 ) at turn-on transition, the IBCC is necessary to store enough energy in resonant inductor (L r1 or L r2 ).Because the voltage across main switch (M 1 or M 2 ) is larger than that across auxiliary switch (M 11 or M 22 ).Thus, the ZVS condition for all of the active switches is determined as follows: ( 1) ] where Δi is the peak current of the coupled inductors, and it can be expressed as: (Note: In the analysis of ZVS condition for the proposed converter, the resonant inductor is expressed as L r = L r1 = L r2 , the clamp voltage of clamp capacitor V cr = V cr1 = V cr2 and the stray capacitor of the active switch is expressed as According to (26), we can plot the curves showing the relationships between resonant inductor L r and output current I o for different values of input voltage V i , as illustrated in Figure 13.From Figure 13, a proper value of resonant inductor L r should be selected for achieving ZVS feature.

Power Losses Estimation
Power losses of the proposed converter are estimated to verify the measured efficiency.The key component values of the experimental converter are shown in Figure 14, from which power losses are evaluated as follows: The main switches of the proposed converter are designed to achieve ZVS during turn-on transition, so that their switching losses can be negligible.Only the conduction losses are taken into account in the estimation.Conduction losses of the main switches can be determined from their switch current i DS and channel resistance R ds(on) .In the proposed converter, the MOSFETs used are IRFP460, and their channel resistance R ds(on) = 0.27 Ω.In the proposed converter, the maximum duty ratio D max = 0.41 and switch current i DS(max) = 9.1 A. Thus, total power losses of the main switches can be determined as: The auxiliary switches of the proposed converter are also designed to achieve ZVS at turn-on transitions, so that their switching losses can also be neglected.Thus, total power losses of the auxiliary switches is determined as: The free-wheeling diodes are selected as 40CPQ60 Schottky diodes, and their forward voltage drop V F = 0.47 V. Thus, total conduction losses of the free-wheeling diodes can be determined as: According to the datasheets of the TDK Company, the cores of the coupled inductors are designed as PC-40 ETD-39 and maximum flux B max = 200 mT.Their total winding turns are n total = (n 1 + n 11 ) = 40, primary winding turns are n p = n 1 = 5, and the turns ratio is n = n total /n p = 8.Thus, from Figure 15, we can obtain the core losses per volume, P Coup(cv) = 0.2 W/cm 3 at 60 °C.The core loss will be:

Experimental Results
Figure 16 shows simulated and experimental voltage and current waveforms of the main switch (M 1 or M 2 ), from which it can be seen that the main switch has ZVS features during the turn-on transition.Figure 17 shows simulated and experimental voltage and current waveforms of the auxiliary switch (M 11 or M 22 ).It can be also seen that the auxiliary switch has ZVS feature during turn-on transition.Figure 18 shows simulated and experimental inductor current waveforms of coupled inductor (i L1 and i L2 ). Figure 19 shows measured current and voltage waveforms of the free-wheeling diode (D 1 or D 2 ), from which it can be seen that the proposed IBCC with active-clamp circuits can limit the rate of decrease of the free-wheeling diode current, reducing reverse-recovery losses a lot. Figure 20 shows measured output voltage and current waveforms.Figure 21 shows efficiency measurements of the IBCC with active-clamp circuits, from which it can be seen that the maximum efficiency can reach as high as 91%.It can increase efficiency about 2% over the IBCC with passive-clamp circuits (as shown in Figure 3).The reason behind is that the active switches of IBCC with passive-clamp circuits are still operated in a hard-switching manner during turn-on transition.

Conclusions
In this paper, an IBCC with active-clamp circuits for wind turbine conversion is proposed.The proposed converter can provide a proper duty ratio for high step-down voltage applications, resulting in low component stresses on active switches.By adopting active-clamp circuits, energy trapped in the leakage inductance of the coupled inductors can be recovered, ZVS features can be achieved and voltage spikes can be suppressed effectively for active switches.Therefore, the conversion efficiency of the proposed IBCC with active-clamp circuits can be increased significantly.In the study, analysis of the proposed IBCC with active-clamp circuits has been presented in detail, including operational principles, feature characteristics and power losses estimation.A 240 W model of the proposed IBCC with active-clamp circuits has been built and implemented.In Figure 21, experimental results can be seen showing that the maximum efficiency can reach as high as 91%.In high step-down voltage applications, the proposed IBCC with active-clamp circuits is relatively attractive for wind energy conversion.

Figure 1 .
Figure 1.Topology of conventional IBC for wind turbine applications.

Figure 2 .
Figure 2. Topology of the IBCC for wind turbine applications.

Figure 3 .
Figure 3. Topology of IBCC with passive-clamp circuits for wind turbine applications.

Figure 4 .
Figure 4. Topology of IBCC with active-clamp circuits for wind turbine applications.

Figure 7 .
Figure 7.The proposed IBCC with active-clamp circuits for wind turbine applications.

Figure 8 .
Figure 8. Key waveforms of the IBCC with active-clamp circuits.

Figure 10 .
Figure 10.Plots of V o /V i versus duty ratio D.

Figure 11 .
Figure 11.Plots of voltage stress versus turns ratio n of the coupled inductor.(a) free-wheeling diodes (D 1 and D 2 ); (b) active switches (M 1 and M 2 ).

Figure 12 .
Figure 12.Performance comparison between the IBCC and the IBC.(a) Duty ratio; (b) voltage stress of the free-wheeling diode.

Figure 13 .
Figure 13.Plots of resonant inductor L r versus output current I o for different values of input voltage V i .

Figure 14 .
Figure 14.Experimental circuit of the IBCC with active-clamp circuits.

Figure 21 .
Figure 21.Plots of efficiency versus output current for the IBCC with active-clamp circuits and without active-clamp circuits at input voltage 150 V dc .
and A L = 3150 nH/N 2 ) to reduce winding current density and core temperature.By applying Faraday's law, turns (n 11 or n 22 ) of the coupled inductor (L 11 or L 22 ) can be determined as:

Table 1 .
Power losses under full load condition on the key components are summarized in Table1.The estimated efficiency of the proposed converter with the active-clamp circuits at input voltage V i = 150 V dc , and full-load condition is: Efficiency estimation of the proposed IBCC with active-clamp circuits.
Figure 15.Typical core loss data of the TDK PC-40.