A Novel Three-Phase Six-Switch PFC Rectifier with Zero-Voltage-Switching and Zero-Current-Switching Features

Abstract

A novel three-phase power-factor-correction (PFC) rectifier with zero-voltage-switching (ZVS) in six main switches and zero-current-switching (ZCS) in the auxiliary switch is proposed, analyzed, and experimentally verified. The main feature of the proposed auxiliary circuit is used to reduce the switching loss when the six main switches are turned on and the one auxiliary switch is turned off. In this paper, a detailed operating analysis of the proposed circuit is given. Modeling and analysis are verified by experimental results based on a three-phase 7 kW rectifier. The soft-switched PFC rectifier shows an improvement in efficiency of 2.25% compared to its hard-switched counterpart at 220 V under full load.

1. Introduction

Power electronic converters play a critical role in the energy industry due to their ability to optimally control and condition the power they deliver to a load. In addition, they are required to control and condition the power they draw from energy sources to support their optimal operation. This is achieved by compliance to EMI and harmonic standards such as EN6100-3-2 and efficiency standards such as 80Plus [1]. Soft-switching technologies are a primary enabler for improving efficiency by minimizing switching losses and reducing EMI and harmonics by “soft” ending the edges of the switching transitions [2,3,4,5,6,7,8,9,10,11,12]. References [8] and [9] report soft-switching techniques that includes zero-voltage-switching (ZVS) and zero-current-switching (ZCS). Three-phase rectifiers with active power-factor-correction (PFC) control achieve an improved power factor and lower harmonic content [10,11,12,13]. Active PFC rectifiers using a boost (current source) front end achieve better input current wave-shaping and lower harmonic distortion compared to their buck-derived counterparts [14]. Three single-phase PFC rectifiers are used in [15] to synthesize a three-phase PFC rectifier. Reference [16] reports the use of space vector modulation (SVM) to achieve a high power factor in a three-phase six-switch rectifier. Soft-switching techniques employed in three-phase rectifiers are reported in [17,18,19] to improve efficiency and EMI performance. Soft-switching using a passive lossless snubber is presented in [17]. Although this approach can improve the efficiency, the circuit suffers from higher component stress. In [18], an active snubber is used to achieve soft-switching at the expense of higher control complexity and switching stress in the auxiliary switch. In [19,20,21,22], the zero-voltage-transition and control technique was applied in a three-phase PFC rectifier. Although the main switches can achieve ZVS at turn-on, the auxiliary switch was hard-switched operated at turn-off.

A conventional three-phase six-switch PFC rectifier is shown in Figure 1. A novel soft-switched three-phase active rectifier using an active auxiliary circuit is proposed in this paper. The principal performance improvement is the achievement of ZVS at turn-on for the six rectifier switches and ZCS at turn-off for the one auxiliary switch. A detailed description of the operation of the proposed soft-switched rectifier is presented in Section 2. Validation of the design through simulation and experimental results are shown in Section 3 followed by concluding remarks in Section 4.

2. Proposed Three-Phase Six-Switch Soft-switching PFC Rectifier

The proposed three-phase six-switch soft-switching PFC rectifier is shown in Figure 2. The circuit inside the dotted box is a soft-switching assist circuit to achieve ZVS in the main switches and ZCS in the auxiliary switch. The soft-switching assist circuit consists of the auxiliary switch S_A, resonant inductor L_R, transformer T_r, barrier diode D_R1, clamp circuit R_C–D_C–C_C, and resonant capacitor (the capacitance employs the parasitic capacitance of main switch).

Three phase line voltages V_RN, V_SN, V_TN for a balanced three-phase system are shown in Figure 3. The 60° symmetry in the three-phase voltages is evident from Figure 3. The operation of the three-phase PFC using the 60° symmetry is described in detail in [11].

In order to simplify the analysis, Interval 1 (0°–60°) can be selected for the analysis of the switching cycles as the operation over the rectifier is identical in the other 60° segments. The following assumptions are made to support the operating analysis:

(1)

Input inductance L_B is large enough to allow the input current to be considered as a current source over a switching period;

(2)

Input capacitance C_L is large enough to be equivalent to the ideal voltage source V_O; and

(3)

The output capacitance of the clamp circuit C_C is large enough to allow its voltage V_C to be considered a voltage source over a switching period.

Under the assumptions listed above, the simplified circuit diagram is shown in Figure 4 and the voltage polarity and current direction for each main component are defined.

A detailed description of circuit operation is provided in this section. The key waveforms of the circuit for Interval 1 are shown in Figure 5, and equivalent circuits for each operating mode are shown in Figure 6. There are 12 operating modes to be analyzed over a switching cycle.

2.1. Mode 0: (t ≦ T₀)

This mode is based on the analysis of the switching cycle in Interval 1 (V_RN > 0, V_TN > 0, and V_SN < 0). Before T₀, as in Figure 6a, the diode D₁, D₆, and D₅ are in the state of conduction. The main switches S₁ to S₆ and auxiliary switch S_A are turned off. The currents i_R and i_T flow through diode D_B to the load and return to the AC source as the current i_S. Under this condition, the voltage across the active rectifier bridge is V_X = V_O.

2.2. Mode 1 (T₀ < t ≦ T₁)

At T₀, the auxiliary switch S_A is turned on to go into Mode 1. The current i₁ of resonant inductor L_R starts to increase, and current i₁ flows through the primary winding N₁ of the transformer T_r. The induced current i₂ and excitation current i_m outflow through secondary coil N₂, as shown in Figure 6b. The voltage on the seconding winding N₂ is the output voltage V_O. The voltage V₁ and V₂ across the windings of transformer T_r are obtained as follows:

$$V_2=V_O$$

(1)

$$V_1=\frac{N_1}{N_2}{V}_2=n{V}_O$$

(2)

The current in the resonant inductor i₁ increases linearly with the slope given by

$$\frac{d{i}_1}{dt}=\frac{V_O-V_1}{L_R}=\frac{V_O-n{V}_O}{L_R}=(1-n)\frac{V_O}{L_R}$$

(3)

Similarly to i₁, the excitation current i_m also displays a linear increase, and the slope is

$$\frac{d{i}_m}{dt}=\frac{V_O}{L_m}$$

(4)

When the current i₁ ascends to i_S current, this mode ends. The time interval is given as below:

$$t_{01}=\frac{i_s}{V_o(1-n)L_R}$$

(5)

2.3. Mode 2 (T₁ < t ≦ T₂)

At t = T₁, the DC link diode current i_b reaches zero. The reverse recovery current of diode D_B flows through diode D_B in a negative direction. The resonant inductor current keeps increasing, as shown in Figure 6c.

2.4. Mode 3 (T₂ < t ≦ T₃)

At t = T₂, the parasitic capacitance of diode D_b along with the resonant capacitor C_R, which includes the parasitic capacitor of the main switches and the resonant inductor L_R, start resonating, as shown in Figure 6d. When the equivalent voltage V_X of the main switch is decreased to zero at t = T₃, the mode ends. The equivalent voltage V_X and resonant current i₁ are shown in Equations (6) and (7).

$$V_X=V_O-(1-n)V_O(1-\cos ({\omega }_{R}t))$$

(6)

$$i_1=i_s+i_{RR}+\frac{(1-n)V_O}{Z_C}\sin ({\omega }_{R}t)$$

(7)

$$C_R=C_2+C_3+C_4$$

(8)

$${\omega }_R=\frac{1}{\sqrt{L_R(C_R+C_b)}}$$

(9)

$$Z_C=\sqrt{\frac{L_R}{C_R+C_b}}$$

(10)

2.5. Mode 4: (T₃ < t ≦ T₄)

When t > T₃, the bridge rectifier voltage V_X is decreased to zero and the auxiliary switch S_A continues to conduct. The corresponding equivalent circuit is shown in Figure 6e. The body diodes D₄, D₃, and D₂ of the main switches S₄, S₃, and S₂ are conducting. Turning on the main switches S₄, S₆, and S₂ when the bridge voltage reaches zero achieves ZVS turn-on. The detection circuitry to turn on the main switches at zero voltage also enables the minimization of duty-cycle loss and, thus, loss of efficiency. After the main switches turn on at ZVS, the resonant inductor current i₁ decreases linearly with the slope given by Equation (11). When the current of the main switches S₄ and S₂ reaches zero at t = T₄, the mode ends.

$$\frac{d{i}_1}{dt}=-\frac{n{V}_O}{L_R}$$

(11)

2.6. Mode 5: (T₄ < t ≦ T₅)

As shown in Figure 6f, when t > T₄, then S₄, S₆, and S₂ keep conducting. The current i_l is continuously decreased to zero until t = T₅.

2.7. Mode 6: (T₅ < t ≦ T₆)

When t > T₅, the input currents i_R and i_T flow through the main switches S₄ and S₂, as shown in Figure 6g. When the current i_a flows through the auxiliary switch S_A, it consists mostly of the magnetizing current, i_m, of the transformer. If the magnetizing inductance L_m is designed to be relatively large, the current i_a of the auxiliary switch S_A is extremely close to zero. When T₅ < t ≦ T₆, the auxiliary switch is set to be turned off so that it can effectively achieve the purpose of ZCS.

2.8. Mode 7: (T₆ < t ≦ T₇)

When t = T₆, the auxiliary switch S_A is turned off, as shown in Figure 6h. Subsequently, the magnetizing current i_m of the transformer charges the parasitic capacitance C_oss1 of the auxiliary switch S_A so that the auxiliary switch voltage will increase continuously.

2.9. Mode 8: (T₇ ≦ t ≦ T₈)

At t = T₇, the auxiliary switch voltage V_SA increases to V_O + V_C and the clamp diode D_C is conducting. The magnetizing current i_m discharges through the clamp circuit D_C–V_C, as shown in Figure 6i. The slope of the excitation current in this mode is given by

$$\frac{d{i}_m}{dt}=-\frac{V_C}{L_m}$$

(12)

2.10. Mode 10: (T₈ ≦ t ≦ T₉)

At t = T₈, the magnetizing current i_m is decreased to zero which resets the transformer, as shown in Figure 6j.

2.11. Mode 10: (T₉ < t ≦ T₁₀)

At t = T₉, the main switch S₄ is turned off. The input current i_R charges the parasitic capacitance of the main switch S₄ and the equivalent voltage V_X of the main switch is increased, as shown in Figure 6k.

2.12. Mode 11: (T₁₀ ≦ t ≦ T₁₁)

At t = T₁₀, the equivalent voltage V_X of the main switch is increased to V_O and the diode D_B is conducting as shown in Figure 6l. Subsequently, the antiparallel diode D₁ of the main switch S₁ is conducting. The input current i_R flows through diodes D₁ and D_B and flows back from i_S through the load.

2.13. Mode 12: (T₁₁ ≦ t ≦ T₁₂)

At t = T₁₁, the main switch S₂ is turned off. The input current i_T starts to charge the parasitic capacitance of the main switch S₂, as shown in Figure 6m.

3. Experimental Verifications

Based on the design described, a prototype was built. When the line voltage in the three-phase input was 220 V, namely, the phase voltage was 127 V, the switching frequency 40 kHz, and the output load 7 kW, then the measured waveforms for three-phase V_RN, V_SN, V_TN, line voltage and the line current were at low line and full load. The current waveform, as in Figure 7, is practically sinusoidal with low THD and a high power factor (the A-THD is shown in Figure 8 and power factor is shown in Figure 9).

Figure 10 shows the simulation waveforms using Isspice at half load and full load. It can be seen that the main switches S₄ and S₂ on the bottom sides turned on when their V_X was down to zero by the resonant circuit, after the auxiliary switch S_A was turned on.

Measured waveforms of the gate drive signals and voltage across the main switch are shown in Figure 11. The captured waveforms indicate the need to turn on the auxiliary switch S_A before the turning on the main switches S₄ and S₂ to achieve ZVS.

As shown in Figure 12, when the current i₁ of resonant inductor L_R decreases to zero, the auxiliary switch S_A can be turned off under ZCS conditions.

A comparison between the hard-switched and proposed soft-switched rectifier was performed at a load of 7 kW over an input voltage range of 190–250 V. The hard-switched rectifier was tested by simply disabling the soft-switch assist circuitry. Figure 13 shows the efficiency improvement from the soft-switched rectifier. The largest efficiency difference between hard-switch and soft-switch rectifier was 2.55% when the input line voltage was 220 V, and Figure 14 shows the soft-switched rectifier efficiency from 1–7 kW.

4. Conclusions

A novel three-phase rectifier with zero-voltage-switching and zero-current-switching features was proposed. The design has been validated with simulation and experimental data captured on a 7 kW three-phase rectifier prototype. Efficiency improvement between hard-switch and soft-switch rectifiers peaks at 2.55% when the input line voltage is 220 V at full load. Mathematical equations to explain circuit operation have been derived and analyzed under a sequence of operating modes. The experimental results have confirmed the proposed design of the soft-switched rectifier in achieving high efficiency, high power factor, and low THD.