Compact Single-Stage Micro-Inverter with Advanced Control Schemes for Photovoltaic Systems

Abstract

This paper proposes a grid-connected single-stage micro-inverter with low cost, small size, and high efficiency to drive a 320 W class photovoltaic panel. This micro-inverter has a new and advanced topology that consists of an interleaved boost converter, a full-bridge converter, and a voltage doubler. Variable switching frequency and advanced burst control schemes were devised and implemented. A 320 W prototype micro-inverter was very compact and slim with 60-mm width, 310-mm length, and 30-mm height. In evaluations, the proposed micro-inverter achieved CEC weighted efficiency of 95.55%, MPPT efficiency >95% over the entire load range, and THD 2.65% at the rated power. The proposed micro-inverter is well suited for photovoltaic micro-inverter applications that require low cost, small size, high efficiency, and low noise.

1. Introduction

The photovoltaic (PV) generation is emerging as a future energy system because of its installation convenience, no-noise, infinite, and eco-friendly characteristics [1,2,3,4]. It is classified into the centralized power system and the distributed power system depending on the scale of solar power generation [5]. The centralized power system has a simple circuit structure with PV strings as the input energy source, but it has a disadvantage that the power generation is considerably lowered when some panels of the PV string are shaded. On the other hand, in the distributed power system, the optimal power extraction is possible because the maximum power point tracking (MPPT) control can be applied to each PV panel with a micro-inverter connected. So, it can minimize the loss of power generation caused by the shading effect. However, one micro-inverter is required for each PV panel, so implementation of this strategy is expensive. Therefore, many attempts have recently been made to lower the cost of micro-inverters.

In general, considering the cost, micro-inverters have been designed to use circuit architectures with a flyback converter [6,7,8,9,10], which provides galvanic isolation with fewer switches than other designs. Although the flyback converter has the advantage of circuit simplicity and low cost, the design must use a transformer with a high turns ratio to achieve a high voltage-conversion ratio from low dc voltage on a single PV panel. In the transformer, the high turns ratio causes a large leakage inductance which increases the stress on semiconductor switches. Moreover, due to low utilization of the transformer, this topology is most suitable for low-power applications <200 W. Recently, multi-phase interleaved technology has been applied to solar power generation from PV panels that output ≥320 W, but this technology requires large and expensive components.

This paper proposes a low-cost, slim, single-stage micro-inverter to drive a 320-W-class PV panel. The proposed micro-inverter has an interleaved structure based on the boost half-bridge (BHB) converter [11] with a cascaded voltage doubler. The interleaved BHB has an inversely-coupled inductor for the voltage step-up operation. The coupled inductor can reduce input ripple current and can be reduced in size. The voltage doubler increases the ac output voltage from the interleaved converter. Therefore, the transformer can have a lower turns ratio in the interleaved BHB than in a flyback converter and can be reduced in size. In the proposed micro-inverter, semiconductor switches achieve turn-on zero-voltage-switching (ZVS) and turn-off zero-current-switching (ZCS) by exploiting the resonance between the leakage inductance of the transformer and output capacitors of the voltage doubler, without additional components.

This paper also presents two advanced control algorithms. First, a variable switching frequency control scheme was implemented to reduce total harmonic distortion (THD) by reducing output ripple current. Then an advanced burst control scheme was implemented to improve power-conversion efficiency at light loads. By distributing output current temporally at light loads, input ripple voltage can be reduced. Therefore, the size of decoupling capacitors is reduced and MPPT efficiency is improved compared with the conventional burst control [12,13]. Section 2 describes the circuit structure and operating principles of the proposed micro-inverter, Section 3 gives the proposed control schemes, Section 4 shows experimental results using a 320-W prototype micro-inverter, and Section 5 concludes the paper.

2. Circuit Structure and Operating Principles of the Proposed Micro-inverter

The proposed micro-inverter (Figure 1) consists of an interleaved boost converter, a full-bridge converter, and a voltage doubler. The portion that is composed of the interleaved boost and full-bridge converters is based on a boost half-bridge topology. The interleaved boost converter consists of an inversely-coupled inductor L_B, four switches S₁–S₄, and a storage capacitor C_S. The full-bridge converter consists of a transformer T₁ and the same four switches S₁–S₄ as the interleaved boost converter. The voltage doubler has four switches S₅–S₈ and two capacitors C₁ and C₂.

In the proposed interleaved boost converter, two inductors L₁ and L₂ form L_B (Figure 2) by using a single magnetic core instead of two separate magnetic cores used in the conventional interleaved boost converter [14]. L_B has a turns ratio of 1:1; L₁ and L₂ each have self-inductance L. The mutual inductance M between L₁ and L₂ is represented as:

$$M=kL,\hspace{0.17em}(k<0),$$

(1)

where k is the coupling coefficient. The voltage drops of L₁ and L₂ are given, respectively, by

$$v_1=L\frac{d{i}_1}{dt}-M\frac{d{i}_2}{dt},$$

(2)

and

$$v_2=L\frac{d{i}_2}{dt}-M\frac{d{i}_1}{dt}.$$

(3)

Using Equations (2) and (3) and $v_m=-Md\left(i_1+i_2\right)/dt$ yields

$$v_1-v_m=(L+M)\frac{d{i}_1}{dt}$$

(4)

and

$$v_2-v_m=(L+M)\frac{d{i}_2}{dt}.$$

(5)

S₁, the body diode of S₂, and L₁ form one boost power stage. S₃, the body diode of S₄ and L₂ form the other boost power stage. The two boost power stages form an interleaved boost converter and two outputs operate out of phase. When S₁ or S₃ is turned on, voltage $v_{IN}$ is applied to L₁ or L₂, respectively. When S₁ or S₃ is turned off, voltage $v_{IN}-v_{Cs}$ is applied to L₁ or L₂, respectively. The energy accumulated during the on-state for each boost power stage is transferred into C_S. There are four cases of the voltage v₁ of L₁ and the voltage v₂ of L₂ depending on the states of S₁ and S₃. Using Equations (4) and (5), the equivalent inductance for each case is obtained (

Table 1. Equivalent inductances in the interleaved boost converter.

Symbol	Value	Condition
Leq1	$\frac{L^2-M^2}{L+DM/1-D}$	$v_1=v_{IN},\hspace{0.17em}{v}_2=v_{IN}-v_{Cs}$
Leq2	$L+M$	$v_1=v_2=v_{IN}$
Leq3	$L+M$	$v_1=v_2=v_{IN}-v_{Cs}$
Leq4	$\frac{L^2-M^2}{L+(1-D)M/D}$	$v_1=v_{IN}-v_{Cs},\hspace{0.17em}{v}_2=v_{IN}$

). M < 0 in Equation (1), so Table 1 demonstrates that appropriate design of the inversely coupled inductor can reduce the input ripple current of the micro-inverter [15].

The full-bridge converter shares four switches S₁–S₄ with the interleaved boost converter, and its input power comes from C_S. The leakage inductance L_lk of T₁ and capacitors C₁ and C₂ in the voltage doubler form an LC resonant circuit. The LC resonant current flows through the primary and secondary sides of T₁ with turns ratio n₁:n₂. This current causes the body diode of each switch to conduct before the turn-on gate signal is applied, thus achieving zero-voltage-switching (ZVS) for S₁–S₄.

In the voltage doubler, S₅–S₈ rectify current on the secondary side of T₁. When grid voltage is positive, both S₅ and S₈ are turned on, and both S₆ and S₇ act as diodes. When grid voltage is negative, both S₆ and S₇ are turned on, and both S₅ and S₈ act as diodes. The energy transferred to the voltage doubler through T₁ is stored in C₁ and C₂. C₁ and C₂ are connected in series, and the output voltage of the micro-inverter is the sum of the voltage v_C1 of C₁ and the voltage v_C2 of C₂.

In the proposed micro-inverter, variable-switching-frequency control is used, and the output voltage is a sinusoidal grid voltage. However, for simplicity, the analysis is based on the assumption that the micro-inverter generates a constant output voltage with a fixed switching frequency at a certain point in the analysis. In addition, the electrical losses of all components are ignored, and the following conditions are assumed: $2\pi \sqrt{L_{lk}\left(C_1+C_2\right)}>D{T}_s$ and $n^2{L}_m>>L_{lk}$, where L_m is the magnetizing inductance and T_s is the switching period. The operation cycle S₁–S₄ is the same regardless of the polarity of the grid voltage, so the analysis considers only positive grid voltage.

The operating waveforms (Figure 3) of the proposed micro-inverter depend on the duty ratio D. First, operational states are analyzed for D ≤ 0.5 (Figure 3a and Figure 4).

State 1 (t₀–t₁): At t = t₀, S₁ is turned on, v_DS1 = 0, and i_SW1 < 0. S₄ remains in the turn-on state, and both S₂ and S₃ remain in the turn-off state. For T₁, the voltage v_Lm across L_m is equal to v_Cs, and the secondary voltage v_s proportional to the turns ratio n₁:n₂ is generated on the secondary side of T₁. The magnetizing current i_Lm is increased and is given by:

$$i_{Lm}(t)=i_{Lm}(t_0)+\frac{v_{Cs}}{L_m}(t-t_0).$$

(6)

Resonance is generated by L_lk on the secondary side of T₁ and capacitors C₁ and C₂, and the state equation is given by

$$L_{lk}\frac{d{i}_s}{dt}=n{v}_{Lm}-v_{C1},$$

(7)

$$i_s=C_1\frac{d{v}_{C1}}{dt}-C_2\frac{d{v}_{C2}}{dt}=(C_1+C_2)\frac{d{v}_{C1}}{dt}.$$

(8)

Using Equations (7) and (8), the secondary current i_s of T₁ is obtained as

$$i_s(t)=\frac{n{v}_{Lm}-v_{C1}}{Z_r}\sin [{\omega }_r(t-t_0)],$$

(9)

where

$$Z_r=\sqrt{\frac{L_{lk}}{C_1+C_2}}$$

(10)

is the resonant impedance and

$${\omega }_r=\frac{1}{\sqrt{L_{lk}(C_1+C_2)}}$$

(11)

is the resonant angular frequency.

From Equations (6) and (9), the primary current i_p of T₁ is obtained as

$$i_p(t)=i_{Lm}(t_0)+\frac{v_{Cs}}{L_m}(t-t_0)+\frac{n^2{v}_{Lm}-n{v}_{C1}}{Z_r}\sin [{\omega }_r(t-t_0)].$$

(12)

From Table 1, the currents i_L1 and i_L2 of the coupled inductor are obtained as

$$i_{L1}(t)=i_{L1}(t_0)+\frac{v_{IN}}{L_{eq1}}(t-t_0),\hspace{0.17em}\hspace{0.17em}\hspace{0.17em}\hspace{0.17em}\hspace{0.17em}\hspace{0.17em}{i}_{L2}(t)=i_{L2}(t_0)+\frac{v_{IN}-v_{Cs}}{L_{eq1}}(t-t_0).$$

(13)

State 2 (t₁–t₂): At t = t₁, S₁ is turned off and S₄ remains in the turn-on state. Both S₂ and S₃ remain in the turn-off state. This interval is a dead time to prevent shoot-through before S₂ is turned on. During this state, the drain-source voltage of S₁ increases from 0 V to v_Cs and that of S₂ decreases from v_Cs to 0 V by charging and discharging parallel capacitance across each switch, respectively.

State 3 (t₂–t₃): At t = t₂, S₂ is turned on, v_DS2 = 0, and i_SW2 < 0. S₄ remains in the turn-on state, and both S₁ and S₃ remain in the turn-off state. For T₁, the voltage v_Lm across L_m is 0 V and the voltage v_lk across L_lk is –v_C1. The amplitude of i_Lm remains unchanged during state 3 as:

$$i_{Lm}(t)=i_{Lm}(t_2)=i_{Lm}(t_0)+\frac{v_{Cs}}{L_m}(t_2-t_0).$$

(14)

i_s begins to decrease because the energy stored in L_lk is transferred to C₁, and is given by

$$i_s(t)\cong i_s(t_2)-\frac{v_{C1}}{L_{lk}}(t-t_2)=\frac{n{v}_{Lm}-v_{C1}}{Z_r}\sin [{\omega }_r(t_2-t_0)]-\frac{v_{C1}}{L_{lk}}(t-t_2).$$

(15)

From Equations (14) and (15), i_p is obtained as

$$i_p(t)=i_{Lm}(t_0)+\frac{v_{Cs}}{L_m}(t_2-t_0)+\frac{n^2{v}_{Lm}-n{v}_{C1}}{Z_r}\sin [{\omega }_r(t_2-t_0)]-\frac{n{v}_{C1}}{L_{lk}}(t-t_2).$$

(16)

From Table 1, i_L1 and i_L2 are obtained as

$$i_{L1}(t)=i_{L1}(t_2)+\frac{v_{IN}}{L_{eq3}}(t-t_2),\hspace{0.17em}\hspace{0.17em}\hspace{0.17em}\hspace{0.17em}\hspace{0.17em}\hspace{0.17em}{i}_{L2}(t)=i_{L2}(t_2)+\frac{v_{IN}-v_{Cs}}{L_{eq3}}(t-t_2).$$

(17)

State 4 (t₃–t₄): At t = t₃, S₄ is turned off and S₂ remains in the turn-on state. Both S₁ and S₃ remain in the turn-off state. This time interval is a dead time to prevent shoot-through before S₃ is turned on. During this state, the drain-source voltage of S₄ increases from 0 V to v_Cs and that of S₃ decreases from v_Cs to 0 V.

The proposed micro-inverter has an interleaved structure, so both the operating principle of the next half cycle for D ≤ 0.5 and the operating principle for D > 0.5 are the same as the above analysis except for the switches used. Thus, further analysis for the others is not given.

The voltage gain G_v of the proposed micro-inverter is twice the product of the boost converter voltage gain and the full bridge converter voltage gain:

$$G_v=\frac{V_{grid}}{V_{IN}}=2\cdot \frac{1}{1-D}\cdot 2nD=\frac{4nD}{1-D}.$$

(18)

3. The Proposed Control Schemes

The main controller (Figure 5) for the proposed micro-inverter takes as analog-to-digital inputs the grid voltage v_grid, the grid current i_g, the input voltage V_IN and the input current I_IN. The MPPT controller is based on the perturb and observe (P&O) MPPT algorithm [16]. This controller determines the amplitude of the reference grid current I_{g_ref} by using I_IN and V_IN to maximize solar power generation. In the P&O MPPT algorithm used (Figure 6), I_{g_ref} is increased if ΔP_IN > 0 and ΔV_IN > 0 or if ΔP_IN < 0 and ΔV_IN < 0. I_{g_ref} is decreased if ΔP_IN > 0 and ΔV_IN < 0 or if ΔP_IN < 0 and ΔV_IN > 0. This process is repeated until the maximum power point (MPP) is reached, i.e., ΔP_IN = 0.

The phase-locked loop (PLL) generates the phase information |sin θ^*| by using v_grid. In the PLL, virtual voltage v_q1 is derived from v_grid for phase detection.

$$v_{q1}(s)=G_{PLL}(s)v_{grid}(s)=V_{grid}(-\frac{1}{s+\omega }+\frac{s}{s^2+{\omega }^2}+\frac{\omega }{s^2+{\omega }^2}),$$

(19)

where G_PLL(s) is PLL gain and V_grid is the amplitude of v_grid.

From the inverse Laplace transform of v_q1(s),

$$v_{q1}(t)=V_{grid}(-e^{\omega t}+\cos \omega t+\sin \omega t)\approx V_{grid}(\cos \omega t+\sin \omega t),$$

(20)

where ωt is the actual phase of the grid.

Using equation (20), the other virtual voltage v_q2 is obtained as

$$v_{q2}(t)=v_{q1}(t)-v_{grid}(t)=V_{grid}\sin \omega t.$$

(21)

v_grid and v_q2 are transformed into the synchronous reference frame as follows:

$$\left[\begin{array}{l}{v}^e{}_{grid}\\ v^e{}_{q2}\end{array}\right]=\left[\begin{array}{l}\hspace{0.17em}\hspace{0.17em}\cos {\theta }^{*}\hspace{0.17em}\hspace{0.17em}\sin {\theta }^{*}\\ -\sin {\theta }^{*}\hspace{0.17em}\hspace{0.17em}\cos {\theta }^{*}\end{array}\right]\left[\begin{array}{l}{v}_{grid}\\ v_{q2}\end{array}\right],$$

(22)

where θ^* is a phase output from the PLL. From Equation (22),

$$v^e{}_{grid}=V_{grid}\cos (\omega t-{\theta }^{*})\approx V_{grid},$$

(23)

$$v^e{}_{q2}=V_{grid}\sin (\omega t-{\theta }^{*})\approx V_{grid}(\omega t-{\theta }^{*}).$$

(24)

The PLL generates θ^* to follow ωt through PI control inside the PLL. The reference current signal i_{g_ref} is the product of I_{g_ref} and |sin θ^*|:

$$i_{g\_ref}=I_{g\_ref}\left|\sin {\theta }^{*}\right|.$$

(25)

The proportional-integral (PI) controller determines the duty ratio variation ΔD by using the difference between i_{g_ref} and |i_g| as follows:

$$\Delta D=K_P(i_{g\_ref}-|i_g|)+K_I{\displaystyle \sum (i_{g\_ref}-|i_g|)}$$

(26)

ΔD compensates for the voltage drop of L_lk, so that i_g follows i_{g_ref}. The nominal duty ratio

$$D_n=\frac{\left|v_{grid}\right|}{G_v}=\frac{\left|v_{grid}\right|}{4n{V}_{Cs}}$$

(27)

provides stable system dynamics for nonlinear sinusoidal waves which are difficult to control using only ΔD. The total duty ratio

$$D=D_n+\Delta D=\frac{\left|v_{grid}\right|}{4n{V}_{Cs}}+K_P(i_{g\_ref}-|i_g|)+K_I{\displaystyle \sum (i_{g\_ref}-|i_g|)}$$

(28)

where D_n is duty ratio generated by the grid voltage and ΔD is a duty ratio variation generated by the grid current. D is given to the pulse-width-modulation (PWM) controller. The PWM controller generates gate signals for switches to track the reference power.

Operating modes (Figure 7) depend on the grid current level when grid voltage is positive. When i_g is low, the proposed micro-inverter operates in discontinuous conduction mode (DCM) because i_g becomes zero before the end of the switching cycle with the period T_s. When i_g is high, continuous conduction mode (CCM) is applied.

If a fixed switching frequency is used for the operating modes, especially the DCM mode, two problems occur: (1) High grid current ripples at low grid currents increase total harmonic distortion (THD); (2) as the output power decreases, the total DCM operating time can increase over the total CCM operating time, and the power conversion efficiency of the micro-inverter can be reduced by high current stress. To solve these problems, this paper proposes two advanced control schemes: Variable-switching-frequency (VSF) control and the advanced burst (AB) control.

3.1. Variable Switching Frequency Control

During T_s, i_s of T₁ in DCM and CCM modes vary with D (Figure 8). As D decreases, the energy stored in L_lk decreases, so time required to demagnetize L_lk decreases. Therefore, the micro-inverter is operated in DCM mode. From Equations (9) and (15), the operating condition for DCM is given by

$$0>\frac{n{v}_{Lm}-v_{C1}}{Z_r}\sin {\omega }_{r}D{T}_s-\frac{v_{C1}{T}_s}{2L_{lk}}(1-2D)$$

(29)

Existing methods to optimize the DCM mode duration have drawbacks. One method is to increase the value of L_lk; a large L_lk increases the inductive energy and increases the demagnetizing time, but this solution requires a large transformer with a large number of windings. Another solution is to increase the switching frequency f_s; this approach can also increase the power density, but high f_s causes high switching loss. Thus, this paper presents VSF control, which minimizes switching loss without increasing the transformer size. VSF control varies f_s depending on the magnitude |i_g| of the grid current.

Fixed-switching-frequency (FSF) control and VSF controls have distinct attributes (Figure 9). FSF control changes only D depending on v_grid (Figure 9a). In contrast, VSF control changes both D and f_s depending on v_grid (Figure 9b). When v_grid is near zero, the switching loss is very small because i_g is close to zero. Therefore, when VSF control is used, f_s is increased to the maximum switching frequency f_max and the time interval between demagnetizings of L_lk is reduced (Figure 9b). As v_grid increases, f_s is decreased to the minimum switching frequency f_min to reduce switching losses. f_s is given by

$$f_s=f_{\max }-(f_{\max }-f_{\max })\frac{v_{grid}}{V_{grid}}$$

(30)

where V_grid is the peak value of v_grid.

3.2. Advanced Burst Control

When solar power generation and load are very small, micro-inverters operate only intermittently to supply the desired power to the grid on an average power basis. This intermittent operation is called “burst control”. For the burst control, the micro-inverter supplies i_g to the grid only during the ON state, and stops running during the OFF state. The burst control improves power-conversion efficiency by reducing the ripple of i_g and switching loss when the load is small.

In the conventional burst control scheme, positive and negative grid currents are consecutively supplied to the grid during one ON-state period (Figure 10). Then OFF-state periods follow the ON-state period. During the OFF state, no power is output, so output occurs only during the ON state, and the energy flowing out of C_IN is also concentrated. Therefore, the input ripple voltage ΔV_IN is increased, the MPPT efficiency is reduced, and additional time is required to charge the input capacitor C_IN for the next operation.

To further improve the performance of burst control, this paper proposes AB control, which supplies positive grid current during the first ON-state (Figure 10). The negative grid current is supplied during the ON-state that immediately follows the first ON-state period. Then, OFF-state periods follow the ON-state periods. This scheme has the effect of distributing the output current temporally compared with the conventional burst control scheme. Therefore, in the proposed micro-inverter with the advanced burst control scheme, the MPPT efficiency can be improved, and the input capacitance C_IN can be reduced due to the reduced ΔV_IN.

4. Experimental Results

The proposed grid-connected micro-inverter (Figure 11) was designed to operate at the rated power 320 W, V_IN = 25~52 V_DC, I_IN.max = 12 A_DC, and f_s = 60~90 kHz. The grid voltage was 220 V_rms, the grid frequency was 60 Hz, and grid current supplied by the proposed micro-inverter is 0~1.45 A_rms. The proposed micro-inverter was implemented using the circuit parameters given in

Table 2. Hardware specifications and circuit parameters.

Unit Type	Symbol	Value	Note
Micro Inverter	Po	320 W	Output power
	CIN	9900 µF	Input capacitor
	L1,L2	190 µH	Self inductance (k = −0.947)
	S1–S4	BCS035N10NS5	MOSFET (VDS = 100 V, ID = 100 A)
	Cs	60 µF	Storage capacitor
	Lm	600 µH	Magnetizing inductance
	Llk	100 µH	Leakage inductance
	S5–S8	IPB65R150	MOSFET (VDS = 650 V, ID = 22.4 A)
	C1,C2	100 nF	Doubler capacitors
	fs	60~90 kHz	Switching frequency
	vgrid	220 Vrms/60 Hz	Grid voltage
	ig	~1.45 Arms/60 Hz	Grid current
	VIN	25~52 VDC	Operating voltage range
	IIN.max	12 ADC	Max input current
PV module	VPV	40.9 V	Open circuit voltage
	VMP	34 V	MPP voltage
	IPV	10.05 A	Short circuit current
	IMP	9.38 A	MPP current

. The microcontroller used was a MN103DF35 (PANASONIC). For the PI controller in the main controller, K_P and K_I were experimentally optimized and set to 9.5 and 200, respectively. The sampling frequency for analog signals is 20 kHz, and the resolution of the analog-to-digital converter is 12 bits. The turns ratio of L_B is 10:10 and that of T₁ is 6:19. The resonant frequency f_r = 35.5 kHz from L_lk = 100 µH and C₁ = C₂ = 100 nF. The MOSFET package of S₁–S₄ is PG-TDSON-8 and that of S₅–S₈ is D²PAK. Capacitors C_s, C₁ and C₂ are MPP-film type. The fabricated micro-inverter was compact and slim with 60-mm width, 310-mm length, and 30-mm height.

Instead of an actual PV module, the photovoltaic simulator ETS600X14CPVF TerraSAS from AMETEK was used as an input source. The solar cell I-V characteristic curve for the experiment was based on that of the NeON®2 PV module from LG electronics.

Gate-source and drain-source voltages were obtained for S₁ and S₂ at D ≤ 0.5 (Figure 12a) and at D > 0.5 (Figure 12b) at V_IN = 34 V and v_grid = 220 V_rms /60 Hz. The drain-source voltage v_DS1 of S₁ drops to 0 V before the gate signal v_S1 is applied, so S₁ turns on with ZVS. S₂ is complementary to S₁ and achieves a ZVS turn-on. The operation of S₃ and S₄ is out of phase with that of S₁ and S₂, so S₃ and S₄ can also achieve the ZVS turn-on.

Waveforms (Figure 13) were obtained for v_grid and i_g at V_IN = 34 V and v_grid = 220V_rms / 60 Hz for output power P_o = 320 W and 64 W. To maximize efficiency, the proposed micro-inverter operates in normal mode at P_o ≥ 110 W and in AB control mode at P_o < 110 W. The boundary of the output power at which the proposed micro-inverter switches from the normal mode to AB control mode and vice versa is selected to be in a range where the peak value of i_g does not exceed the rated grid current.

Waveforms were obtained for the fixed-frequency (Figure 14a) and the variable-switching-frequency (Figure 14b) controls. Gate signals of S₁ and S₃, i_g and v_grid were measured at V_IN = 34 V, v_grid = 220 V_rms / 60 Hz, and output power P_o = 320 W. When fixed-switching frequency control was used, i_g was distorted near zero-crossing, and THD was increased to 5.79%. In contrast, when variable switching frequency control was used, the distortion of i_g was improved near zero-crossing, and THD was reduced to 2.65%, which is below the requirement for distributed power. The switching frequency f_s decreased as i_g increased, so switching loss was also reduced.

ΔV_IN is higher when conventional burst control is used (Figure 15a) than when AB control is used (Figure 15b), because AB control reduces the energy supplied by C_IN during one ON-state period. At V_IN = 34 V, v_grid = 220 V_rms / 60 Hz, and P_o = 32 W, ΔV_IN was 4.2 V when conventional burst control was used, but 2.4 V when AB control was used.

The MPPT efficiency of the proposed micro-inverter was measured (Figure 16) in the range of irradiance from 50 W/m² (P_o = 16 W)–1000 W/m² (P_o = 320 W). In the proposed control scheme, for P_o < 110 W (burst mode), the MPPT efficiency was kept >95% because ΔV_IN and ΔI_{g_ref} are reduced. However, in the conventional control scheme, the MPPT efficiency was reduced to ~88% because fluctuation of I_{g_ref} increased. During burst mode, the maximum MPPT efficiency was >99% for the proposed control scheme but <97.5% for the conventional control scheme.

In a micro-inverter, one of the most important factors is the power conversion efficiency ɳ_e for 50~75% load under actual solar irradiation. Therefore, the California Energy Commission (CEC) weighted efficiency to represent this fact has been widely used to measure the performance of micro-inverters. The power conversion efficiency ɳ_e (Figure 17) was measured for the proposed micro-inverter; the result indicate that the CEC weighted efficiency [17,18] is 95.55%, in which ɳ_e(10%) = 91.71%, ɳ_e(20%) = 94.42%, ɳ_e(30%) = 95.28%, ɳ_e(50%) = 96.06%, ɳ_e(75%) = 95.8%, and ɳ_e(100%) = 95.72%. The maximum ɳ_e is 96.06% for P_o = 160 W.

5. Conclusions

A compact single-stage micro-inverter with advanced control schemes for PV systems is described. The proposed micro-inverter achieved a high voltage-conversion ratio and high efficiency by using a new topology that consists of an interleaved boost converter, a full-bridge converter, and a voltage doubler. The leakage inductance of the transformer and the capacitors of the voltage doubler ensure ZVS condition without any additional components. A variable-switching-frequency control scheme is applied to the micro-inverter to decrease THD by reducing the grid ripple current. An advanced burst-control scheme increases MPPT efficiency with smaller input ripple voltage than the conventional burst control causes. A fabricated 320-W prototype micro-inverter was very compact and slim with 60-mm width, 310-mm length, and 30-mm height. It achieved CEC weighted efficiency of 95.55%, MPPT efficiency > 95% over the entire load rage, and THD 2.65% at V_IN = 34 V, v_grid = 220 V_rms/60 Hz, and P_o = 320 W. These results show that the proposed micro-inverter is well suited for PV micro-inverter applications that require low cost, small and slim size, high efficiency, and low noise.