A Compact and Wideband Beam-Scanning Antenna Array Based on SICL Butler Matrix

Abstract

A compact and wideband beamforming antenna array based on a substrate-integrated coaxial line (SICL) Butler matrix at 60 GHz is proposed in this paper. The cavity-backed patch antenna loading double-ridged horn antenna is designed to enhance a gain of 5.4 dB and a bandwidth of 2.7 GHz. Different phase centers of double-ridged horn elements are formed into a non-uniform array to reduce sidelobes by −7.9 dB. By introducing the defected ground structure (DGS) for a broadband coupler, a rotationally symmetric SICL Butler matrix is designed with a 55–70 GHz bandwidth and compact dimensions of 63 × 65 × 0.512 mm³. To validate the design, a prototype was fabricated and measured. The experimental results show a wideband −10 dB impedance bandwidth of 23.3% (55.4–70 GHz) with measured gains ranging from 15 to 16.1 dBi at 62 GHz. The one-dimensional beam scanning covers ±32°. The simulation and measurement results are in good agreement.

1. Introduction

In recent years, beam-scanning antenna arrays have been widely used in fifth-generation mobile communications to enhance data transmission and coverage capabilities [1]. In beamforming antenna arrays, the feeding network plays a crucial role in distributing the desired signal magnitude and phase delay among individual array elements. Commonly employed topologies include the series-feed, parallel-feed, and matrix feeding networks. Compared with conventional series and parallel-feed networks, the matrix feeding network functions as a multiple-input and multiple-output system, comprising components such as couplers, phase shifters, crossovers, power dividers, and switches. Among these, the Butler matrix (BM) stands out for its symmetrical structure, maintaining an equal number of inputs and outputs [2]. Compared to lens-based beamforming networks, the Butler matrix utilizes fewer couplers and phase shifters, resulting in a more compact design [3,4,5,6].

A variety of Butler matrix configurations have been designed to function as indispensable components in multibeam antenna systems. To fulfill the miniaturization requirements, dual-layer BMs [7,8,9,10] have been implemented. A multilayer interconnection structure is employed to construct the vertical-plane BM, replacing the crossover, which achieved the goal of miniaturization. However, it introduced the problem of design complexity. In [9], a novel 4 × 4 BM was developed by utilizing couplers with −45° and −90° phase differences. This design eliminated the need for a phase shifter. This novel Butler matrix offers significant improvements in phase and amplitude performance while maintaining a compact topology. Moreover, certain new structures demonstrated compact designs, such as the substrate-integrated waveguide (SIW) [11,12,13], half-mode substrate-integrated waveguide (HMSIW) structure [14], ridge gap waveguide [15,16], groove gap waveguide (GGW) [17], SICL [18,19], and single-pole double-throw switching structure [20].

Meanwhile, the BM produced a high sidelobe level (SLL) to multibeam array antenna, because the conventional BM provides equal power but different phases at all outputs. The peak sidelobe levels (PSLs) cannot be better than −10 dB. Tapered amplitude distribution [21,22,23] for the antenna is a significant method for SLL suppression. In one solution to implement this method, unequal split power dividers are added to the circuits. Another efficient work is using attenuators as external channels of the BM [24,25]. However, external attenuators always bring significant insertion losses when operated at the mm wave band.

To address the aforementioned challenge, this paper introduces a compact and wideband beam-scanning antenna array based on the SICL Butler matrix. The 4 × 4 rotationally symmetric SICL Butler matrix is designed, fabricated, and measured for verification. Furthermore, it is connected to a double-ridged horn antenna array to demonstrate its performance as a beamforming network.

The organization of this paper is as follows: An introduction is given in Section 1. Section 2 contains detailed information about the design of the double-ridged horn antenna element. In Section 3, the 3 dB three-branch-line coupler with DGS is combined into the wide bandwidth SICL BM. The non-uniform double-ridged hybrid horn antenna array is given in Section 4. Simulated and measured results are provided in Section 5. Finally, in Section 6, the conclusions are presented.

2. Design of Cavity-Backed Ridged Hybrid Horn Antenna Element

The perspective view of a SICL is shown in Figure 1, which composes two substrate layers (substrate 1 and substrate 3) and one bonding film layer (substrate 2). There are two metallic layers printed on the top of substrate 1 and the bottom of substrate 3. The inner conductor as a signal line is etched on the top of substrate 3. Two additional side walls made of via holes are added around two sides of the inner conductor to form the SICL structure. SICL configuration can be recognized as a planar coaxial line structure, which has the advantages of non-dispersion and miniaturization. The dielectric substrates 1 and 3 are Rogers 5880 (tan δ = 0.0009, ε_r = 2.2, h = 0.254 mm), and bonding film (substrate 2) is Rogers 4450F (tan δ = 0.004, ε_r = 3.52, h₁ = 0.1 mm). The simulated length of the SICL and SIW is 20 mm, and the cross-section sizes are 1.3 mm and 2.65 mm, respectively. The simulated |S₂₁| of the SICL and SIW is −0.22 dB and −0.4 dB up to 70 GHz, as shown in Figure 2. The insertion loss of the SICL is better than SIW.

The proposed antenna element comprises a double-ridged horn, cavity-backed patch, and SICL feeding line, as depicted in Figure 3. The double-ridged horn is on the top of substrate 1. The patch is located on the top metal layer of substrate 1. The SICL signal line serves as a feeding line located under the patch. Considering the size of the cavity-backed patch, w and d₅ are the width and length for the inner rectangular aperture. The distance between antenna elements is generally between λ₀/2 and λ₀. The H-shaped double-ridged horn is designed with the horn aperture width (d₄) and the angle (θ_r). The curvature of the ridge along the longitudinal direction (x-axis) is determined by a modified exponential function [26]:

$$y(x)=a{e}^{-kx}\hspace{1em}0\le x\le s$$

(1)

where x represents the axial length along the horn, starting from the end of the ridge’s straight section, and s denotes the axial length of the ridge.

The design of the internal ridges within the pyramidal horn cavity serves a dual purpose. On one hand, it guides the transmission of the current further. On the other hand, it ensures impedance matching and gradual transition to the external air for effective impedance matching. These dimensions can be seen in

Table 1. The optimal dimensions of antenna element (Unit: mm).

Parameters	l1	l2	l3	l4	l5	l6	l7	l8	w1	w2	w3	w4
Values	5.74	1.86	0.22	0.83	1.3	0.2	0.59	7.18	2.83	2.3	0.73	2.11
Parameters	w5	w6	d1	d2	d3	d4	d5	θr	s	l	w	d
Values	0.36	0.25	6	10	3	7	1.7	24°	7.32	15	3.5	2

. In Figure 4, the simulated −10 dB impedance bandwidths of the cavity-backed patch-loaded and -unloaded double-ridged horn are 9.6 GHz (55–64.6) and 12.3 GHz (54.6–66.9 GHz), respectively. The bandwidth of the cavity-backed patch antenna loading ridged horn is 2.7 GHz larger than that of the cavity-backed patch antenna without a double-ridged horn. The resonance within the double-ridged horn generates additional resonance points, thereby expanding the antenna’s bandwidth. The simulated gain of the loaded ridged horn is 10.8 dBi at 60 GHz, which increased 5.1 dB.

3. Design of Compact 4 × 4 Rotationally Symmetric Butler Matrix Based on SICL

The schematic of the 4 × 4 symmetrical Butler matrix is composed of four 3 dB couplers, two 45° phase shifters, and two crossovers, as shown in Figure 5a. The rotational symmetry structure avoids using crossovers for miniaturized design, as shown in Figure 5b. The proposed rotationally symmetric Butler matrix, with its highly integrated design and placement beneath the radiators, enhances the compactness and interference-free characteristics of the presented antenna design.

To facilitate a clearer understanding of this design, the output ports are categorized into two groups: P5–P6 and P7–P8, with a progressive phase increment α assigned to the elements within each group. To meet the required phase difference between the adjacent ends of the two groups, the following criterion is established [27]:

$$\alpha =\left(\beta +2K\pi \right)/4$$

(2)

$$\beta =\left(2M+1\right)\pi$$

(3)

where β represents the phase difference between the outputs of the power splitters and K and M are integers. The phase increment α can take one of the following values: ±45° or ±135°, while β is set to π for simplification. Consequently, K is assigned the values −2, −1, 0, and 1, which correspond to the phases of P3, P1, P4, and P2, respectively.

3.1. Wideband 3 dB Three-Branch-Line Coupler with DGS

A planar 3 dB three-branch-line coupler with a DGS is proposed, as shown in Figure 6. A DGS can be realized by etching three rectangular slots on the top and bottom grounds. The DGS section increases the electrical length and provides a wider effective characteristic impedance compared to that of a conventional microstrip. The rectangular shape of the DSG is optimized to control the design of the coupler, as shown in

Table 2. The dimensions of 3 dB coupler with DGS (Unit: mm).

Parameters	Cw1	Cw2	Cw3	Cw4	CL1	CL2	Sw1	Sw1	SL
Values	0.33	0.23	0.1	0.22	1.82	0.92	0.21	0.29	0.75

. The phase difference between Port 2 and Port 3 is 90° ± 1°. The |S₂₁| and |S₃₁| are less than −3.5 dB that induces 0.5 dB of loss. The bandwidth of this three-branch 3 dB coupler ranges from 55 to 70 GHz (21%), as shown in Figure 7.

3.2. WR-15-to-SICL Transition

Figure 8a shows the overall structure of the SICL to waveguide (WR-15). The coupling aperture is etched on the top of the cavity, which consists of many metallic via holes. The inner conductor probe has a wider width than the SICL transition and makes the shape of the rectangular to achieve impedance matching. Figure 8b shows that the simulated |S₁₁| is less than −20 dB and |S₂₁| is more than −1 dB when the frequency ranges from 57 GHz to 68 GHz. This design can provide the BM with a convenient connection to waveguide measurement instruments in millimeter-wave applications.

3.3. 4 × 4 SICL Butler Matrix

Figure 9 shows the topology of the proposed 4 × 4 rotationally symmetric BM, which comprises four 3 dB three-branch-line couplers with a DGS, two −45° phase shifters, and four WR-15-to-SICL transitions [28]. A designed 4 × 4 rotationally symmetric SICL BM occupies an area of 63 × 65 × 0.518 mm³. The simulated results of the SICL Butler matrix are shown in Figure 10. The simulated −15 dB impedance bandwidth is 17% (56–68 GHz). The magnitudes of |S₅₁|, |S₆₁|, |S₇₁|, |S₈₁|;|S₅₂|, |S₆₂|, |S₇₂|, and |S₈₂| are less than −6.8 ± 0.8 dB. Due to the symmetry of the SICL Butler matrix, the insertion loss of other ports is omitted. The phase differences among different output ports are −45°, 45°, −135°, and 135° with phase fluctuations within ±5° over the frequency range. The phase differences of the designed BM meet the requirements of a beam-scanning antenna array.

4. Design of Non-Uniform Double-Ridged Horn Antenna Array

In Figure 11, the cavity-backed antenna and Butler matrix are positioned within the substrate, while the dual-ridged horn antenna array is secured to the substrate using screw holes. The BM generates a beam-scanning performance, but also produces high sidelobes. The traditional uniform horn antenna array has the same aperture and element distance. The non-uniform horn antenna array consists of two types of double-ridged horn apertures with S₁ and S₂. The aperture of the double-ridged horn antenna elements on both sides is extend outward, which is equivalent to introducing a phase difference between the double-ridged horn elements of different apertures. The non-uniform element distribution improves the sound field pattern and optimizes the energy distribution, which achieving a low sidelobe effect. W_out and L_out are the width and length for the inner rectangular aperture of the horn array. These dimensions can be seen in

Table 3. The dimensions of non-uniform antenna array (Unit: mm).

Parameters	Lout	Wout	Hin	Lin	Win	S1	S2
Values	22	7	1	14.5	1.7	3.5	4

.

The beams of the non-uniform double-ridged horn antenna array can scan up to angles of −31°, −7°, 8°, and 32° at 62 GHz and a half-power beamwidth (HPBW) of 19°~23° and an SLL lower than −14.4 dB, as illustrated in Figure 12. The non-uniform double-ridged horn antenna array reduces the sidelobe level by −7. dB compared to the uniform array.

5. Measurement and Discussion

The fabricated prototype of the proposed beam-scanning antenna array is shown in Figure 13a,b. In this processing accuracy, the minimum line width for multi-layer boards is 3.5 mil (0.0889 mm) and the minimum line spacing is 6 mil (0.153 mm). Therefore, this processing accuracy can meet the minimum line width design requirement. The gain measurement setup is depicted in Figure 13c and operates in a microwave anechoic chamber, with each input port fed individually while the remaining ports are terminated using 50-ohm loads. The size of the 4 × 4 rotationally symmetric BM and non-uniform double-ridged horn antenna array is 63 × 65 × 0.512 mm³ and 55 × 17.7 × 11.5 mm³, respectively.

From 55.4 to 70 GHz, the measured |S₁₁|, |S₂₂|, |S₃₃|, and |S₄₄| are better than −10 dB as shown in Figure 14a. The simulated beam directions are −31°, −7°, 8°, and 32° with gains of 16.5, 17.1, 17.2, and 16.1 dBi at 62 GHz. The measured beam directions are −30°, −9°, 10°, and 30°. The measured gains fed in four input ports are 15.4, 16, 16.1, and 15 dBi at 62 GHz. The measured gains of the beams are slightly lower than the simulated gains due to losses from the waveguides used in the measurement connections. The measured loss of the straight waveguide is approximately −0.52 dB at 62 GHz. Due to the CCR antenna measurement, the overall loss is doubled up to −1.04 dB at 62 GHz. Therefore, the −1~−1.4 dB decrease in gain between the simulation and measurement is due to waveguides. The simulated results, after accounting for measurement losses, closely match the measurement results.

The characteristics and performance metrics of the proposed and previously reported millimeter-wave beam-scanning antenna arrays are summarized in

Table 4. Comparison among different millimeter-wave beam-scanning antenna arrays.

Ref.	Type	BM	Size (λg)	Bandwidth	Gain (dBi)	Beam (°)
[4]	MS	1 × 4	6 × 6 × 0.03	8.83%(@60GHz)	7.7~8.9	±14, ±40
[5]	SIW	4 × 4	11 × 4.2 × 0.05	13%(@30GHz)	9.7~12	±12, ±51
[10]	SIW	4 × 4	37.5 × 24	22.5%(@37.5GHz)	10.4–12.8	±13, ±38
[15]	RGW	4 × 4	9 × 5.4	21.25%(@30GHz)	10.2~11.3	±13, ±36
[16]	RGW	4 × 4	14.4 × 26 × 0.6	21.5%(@61.5GHz)	25.1~26.3	±17, ±43
This work	SICL	4 × 4	11 × 10 × 2.03	23.3%(@62GHz)	15.1~16.5	−30°, −9°, 10°, 32°

. The operating bandwidth of the proposed array is broader compared to other designs operating at 62 GHz. Furthermore, due to the low-loss characteristics of the SICL Butler matrix, high gains and compact performances are achieved in this work. This indicates that it is of value for 5G wireless communication.

6. Conclusions

A wideband beam-scanning antenna array based on a compact 4 × 4 SICL BM is proposed for 60 GHz applications. The design integrates a non-uniform double-ridged horn antenna array and a 4 × 4 rotationally symmetric BM, for compact and efficient implementation. Beam-scanning performance is achieved through both simulations and measurements. The simulated and measured results exhibit good agreement with a 23.3% bandwidth and a 15 to 16.1 dBi gain. The measured beam directions are −30°, −9°, 10°, and 32°. Compared to other beam-scanning antenna arrays, the proposed antenna array offers unique advantages of a wideband and high gains. This makes it highly suitable for base-station applications in 5G wireless communication systems.