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Article

Compact Circularly Polarized Cavity-Backed Crossed-Dipole Antenna with Ultra-Wide Bandwidth for Integrated GNSS–SatCom Terminals

1
School of Information and Communication, Guilin University of Electronic Technology, Guilin 541004, China
2
School of Artificial Intelligence, Guilin University of Aerospace Technology, Guilin 541004, China
3
Sichuan Technology Innovation Center for Intelligent Sensing & Computing Chip and System, Chengdu 610016, China
*
Author to whom correspondence should be addressed.
Electronics 2025, 14(16), 3193; https://doi.org/10.3390/electronics14163193
Submission received: 21 July 2025 / Revised: 8 August 2025 / Accepted: 9 August 2025 / Published: 11 August 2025

Abstract

As wireless systems evolve toward multiband, multifunctional convergence and high-throughput services, the demand for ultra-wideband circularly polarized (CP) antennas for multi-standard terrestrial–satellite terminals continues to grow; however, because of the dispersive nature of the three-quarter-ring phase shifter, the relative bandwidth achievable with conventional crossed-dipole antennas rarely exceeds 100%. This paper presents a compact left-hand circularly polarized (LHCP) crossed-dipole antenna that combines a cavity-backed ground, ground-slot perturbations, and parasitic patches to simultaneously broaden the impedance and axial-ratio bandwidths. The fabricated prototype achieves an impedance bandwidth (IMBW) of 0.71–3.89 GHz (138%) and a 3 dB axial-ratio bandwidth (ARBW) of 0.98–3.27 GHz (108%), while maintaining gains above 3.5 dBic across most of the frequency range. The good agreement validates the multi-technique co-design and shows that the compact architecture (0.302 λ × 0.302 λ × 0.129 λ) breaks classical crossed-dipole limits. The antenna provides a scalable building block for wideband conformal arrays in next-generation integrated GNSS–SatCom systems.

1. Introduction

Circularly polarized (CP) antennas are indispensable components of modern wireless communication systems and offer unrivaled performance advantages in Global Navigation Satellite Systems (GNSS), 5 G millimeter-wave communications, the Internet of Things (IoT), radio-frequency identification (RFID), and radar-sensing applications [1,2,3,4,5]. Compared with the linearly polarized counterparts, CP antennas exhibit lower polarization-mismatch loss, immunity to Faraday rotation, enhanced multipath-interference rejection, and greater flexibility in polarization alignment, which together render them especially suitable for reliable communication in complex electromagnetic environments [1]. As wireless systems evolve toward multiband, multifunctional convergence and high-throughput services, antenna platforms that efficiently support spectral fusion across multiple standards not only reduce the number of radiating elements required in a terminal but also facilitate the seamless integration of multifunctional modules [4,6,7].
Although many antenna types can generate CP radiation, the crossed dipole remains highly attractive for multi-standard, multifunctional platforms as it offers an exceptionally wide impedance bandwidth (IMBW) together with an equally broad 3 dB axial-ratio bandwidth (ARBW) [8,9,10,11,12,13,14]. In the classical crossed-dipole configuration, the inner and outer conductors of a coaxial cable independently excite the orthogonal dipole arms, while a three-quarter-arc microstrip phase-delay line imposes a 90° phase shift between them to realize CP radiation. This compact, low-complexity feed arrangement eliminates the need for additional circuitry and has therefore become one of the most widely adopted architectures for CP radiators. Consequently, extensive research over the past several decades has advanced crossed-dipole CP antennas in terms of wideband characteristics [8,12,15,16], multiband functionality [17,18,19], high gain [20,21,22], low profile [18,23,24], broad beamwidth [7,10,18], and radiation-reconfigurable capability [25,26,27].
However, because microstrip lines are dispersive, a three-quarter-arc microstrip phase-delay line provides an exact 90° phase shift only at its center frequency; as the operating frequency deviates from this value, the phase error increases, and the antenna’s axial-ratio performance deteriorates. Consequently, the attainable 3 dB ARBW is usually limited to only 10–20% of the center frequency. To overcome this limitation, substantial bandwidth enhancement has been realized through (i) reshaping the dipole arms [4,8,28,29,30], (ii) adding coplanar parasitic patches of various shapes [4,13,22,31,32], (iv) integrating multifarious back cavities [10,12,28], (iv) embedding square resonant rings [17,19,32,33], and (v) etching defected-ground structures [12,33,34], which collectively expand both the IMBW and the 3 dB ARBW of crossed-dipole antennas. In practice, multiple techniques are frequently combined to achieve still broader bandwidth.
Although state-of-the-art broadband circularly polarized antennas already deliver IMBWs approaching 110% and 3 dB ARBWs near 100%, the pursuit of even higher performance remains vigorous. To push this frontier further, we propose an ultra-wideband (UWB) circularly polarized crossed-dipole antenna that surpasses conventional limits through a multidimensional co-design methodology. Building on a rectangular crossed-dipole architecture, the antenna employs a metallic quasi-back cavity to broaden its IMBW, while a defected-ground structure (DGS) lowers the resonant frequency. Tightly packed patches enhance inter-patch coupling and mitigate the phase dispersion introduced by the three-quarter ring; moreover, nearby parasitic elements further improve the axial-ratio characteristic and widen the 3 dB ARBW. Through holistic optimization, the antenna achieves an impressive 138% IMBW spanning 0.71–3.89 GHz and a 108% 3 dB ARBW covering 0.98–3.27 GHz, yielding a 108% overlapping bandwidth.

2. Antenna Design

2.1. Antenna Configuration

The conventional half-wavelength dipole antenna operates in its fundamental resonant mode and is therefore intrinsically narrowband. Conversely, broadband dipole designs evolved from the centrally fed three-dimensional biconical antenna, whose IMBW grows with the effective width of its dipole arms. Planar biconical structures with a finite number of edges appear as bow-tie, diamond, or hexagonal shapes, whereas those with many edges approximate circular or elliptical geometries. These planar antennas feature wide arms that support both the fundamental resonant mode and closely spaced higher-order modes, thereby enabling broadband operation.
To preserve broadband performance while reducing the overall dimensions, we employ a tightly packed array of rectangular radiating patches. This compact layout leverages strong inter-patch coupling to mitigate the phase dispersion introduced by the three-quarter-ring microstrip feed line. A slotted ground plane incorporating DGSs, together with vertical parasitic patches, creates a defected resonator cavity that excites additional resonant modes and thus broadens the IMBW. Additional parasitic patches placed adjacent to the dipole arms further extend the 3 dB ARBW. The final antenna configuration is illustrated in Figure 1, and the corresponding geometric dimensions are summarized in Table 1.

2.2. Design Evolution

The widespread adoption of crossed-dipole antennas stems from their streamlined feed scheme: the intrinsic 180° phase difference between the inner and outer conductors of a coaxial cable, combined with a pair of three-quarter-ring microstrip phase shifters that provide an additional 90° delay, yields the sequential 90° phase progression required for circularly polarized excitation. However, the three-quarter-ring microstrip section produces an exact 90° phase shift only at its design frequency; signals above this frequency incur phase delays exceeding 90°, those below it incur delays less than 90°, and the deviation from the ideal increases with greater frequency offset. For example, a three-quarter-ring line that yields a 90° phase delay at 2 GHz increases to 135° at 3 GHz and decreases to 45° at 1 GHz, making it challenging to preserve high-quality circular polarization across an ultra-wide bandwidth. Accordingly, we enhance the conventional crossed-dipole topology by introducing tight coupling between the radiating elements and adding supplementary parasitic patches, thereby realizing an UWB, circularly polarized antenna with superior performance.
To elucidate the operating mechanism, the step-by-step design procedure is presented in Figure 2, whereas the corresponding reflection coefficient, peak gain, and axial-ratio responses are plotted in Figure 3. As illustrated in Figure 2a, the baseline prototype (Antenna 1) comprises a compact rectangular crossed-dipole radiator backed by a planar ground plane. To broaden the IMBW, metal vertical plates are added, each shorted to the ground plane to create a defected resonator cavity that excites additional modes; the resulting configuration (Figure 2b) expands the bandwidth from 1.40–3.37 GHz to 0.81–3.25 GHz. Next, four slot-line perturbations are etched into the ground plane to create a DGS (Figure 2c); although this modification only modestly affects the absolute bandwidth, it downshifts the impedance band, extending it to 0.72–3.18 GHz. Although the IMBW is now wide, the 3 dB axial-ratio bandwidth remains limited; therefore, a set of rectangular parasitic elements is placed above the dipole layer to enhance circular-polarization performance, yielding the final design (Figure 2d) with an IMBW of 0.73–3.11 GHz and a 3 dB axial-ratio bandwidth of 0.99–3.29 GHz.

2.3. Antenna Analysis

Based on the Antenna 4 configuration, further parametric optimization was carried out using a full-wave electromagnetic solver, and the resulting geometric parameters are summarized in Table 1. The simulated reflection coefficient, peak gain, total efficiency, and axial ratio derived from the optimized parameters correspond to the curves for Antenna 4 shown in Figure 3a–d. As can be seen from the figures, the simulated IMBW, defined by |S11| < −10 dB, is 0.72–3.25 GHz, and the axial-ratio bandwidth (AR < 3 dB) extends from 0.99 to 3.27 GHz; consequently, the effective operating band is 0.99–3.27 GHz.
As a key component of wireless communication systems, an antenna converts electromagnetic energy bidirectionally between guided and free-space modes; consequently, radiation gain, polarization, and radiation-pattern characteristics constitute its primary performance metrics. Figure 3b shows that the peak gain remains above 4 dBic throughout the 0.70–3.25 GHz band. Figure 4 further demonstrates that the proposed cavity-backed crossed-dipole antenna delivers inherently stable radiation; however, in the UWB regime, the electrical wavelength decreases as the frequency increases. Over this interval, λ decreases from 303 mm to 91.7 mm, increasing the electrical size from 0.31 λ to 1.01 λ. Accordingly, as the frequency rises, the antenna gain increases, the half-power beamwidth narrows, and the side-lobe and back-lobe levels rise slightly, as corroborated by the 3D radiation patterns in Figure 4.
The axial ratio, another fundamental performance metric of an antenna, is chiefly determined by the temporal evolution of the surface currents on the radiating elements—specifically, by the periodic rotation of the resultant surface-current vector. Although the metal cavity-backed enclosure of the proposed antenna radiates some energy, the dominant radiation originates from the dipole patches; hence, our analysis focuses on their surface-current distributions. The surface-current distributions of the proposed antenna at 1 GHz, 2 GHz, and 3 GHz for phase angles of 0°, 90°, 180°, and 270° are illustrated in Figure 5, Figure 6 and Figure 7. These results show that loading asymmetric parasitic patches alters part of the current path and thereby enhances the CP performance.
Owing to the rotationally symmetrical arrangement of all radiating patches, variations in the feed phase induce corresponding changes in the surface-current directions, so that the resultant surface-current vector rotates clockwise and thereby generates left-hand circular polarization (LHCP). As demonstrated by the surface-current distributions in Figure 5a–d, the current on every radiating patch rotates in concert with the feed phase. Specifically, each 90° increment in the feed phase produces a matching 90° rotation of the resultant current vector. A schematic inset in the upper-right corner of each subfigure depicts these directions: black arrows denote the current vectors on individual patches, whereas the red arrow indicates the resultant vector. The surface-current distributions in Figure 6, Figure 7, Figure 8 and Figure 9 exhibit the same behavior.
Figure 10 illustrates the axial-ratio response of the proposed antenna in terms of θ. As the frequency increases, the 3 dB axial-ratio beamwidth narrows, and the sidelobes become increasingly erratic near the upper end of the band. This behavior mirrors the half-power beamwidth (HPBW) variation: at higher frequencies, the larger electrical size excites non-uniform surface currents on the radiating patches, thereby degrading the axial-ratio performance.

3. Experimental Results and Discussion

3.1. Antenna Fabrication

To experimentally verify the proposed antenna, a prototype was fabricated and measured; the assembled prototype is shown in Figure 11a–d. The radiating patch section, shown in Figure 11a,b, is fabricated on a ZYF300CA dielectric substrate composed of PTFE resin, glass-fiber cloth, and ceramic filler; the substrate exhibits a relative permittivity (εr) of 2.94 and a loss tangent of 0.0016 at 10 GHz. The metallic resonant cavity was machined from a 0.5 mm thick oxygen-free copper sheet using CNC milling for the outline shown in Figure 11c, and it was subsequently folded along the designated bend lines to produce the configuration shown in Figure 11d.
Following the assembly of the radiating patches, metallic cavity back, coaxial line, and SMA connector, the completed prototype is depicted in Figure 11e. The coaxial feed penetrates the central circular aperture of the back cavity; its outer conductor is soldered to both the cavity walls and the rear radiating patch, whereas its inner conductor is soldered to the front radiating patch. To enhance mechanical stability, the edges of the radiating patches are bonded with epoxy resin to the upper rims of the cavity’s vertical metal walls.
During assembly, the coaxial cable is inserted through the center aperture of the cavity back; its outer conductor is soldered to the cavity wall and to the viahole in the bottom patch, whereas its inner conductor is soldered to the viahole in the top patch. To enhance mechanical stability, the edges of the radiating patches are bonded with epoxy resin to the upper rims of the cavity’s vertical metal walls. The fully assembled prototype—comprising the radiating patches, back-cavity coaxial feed, and SMA connector—is illustrated in Figure 11e.

3.2. Reflection Measurement

The assembled prototype was measured for its reflection coefficient using a Ceyear 3656B vector network analyzer. The measured data were then exported and compared with the simulated results, and the corresponding traces are plotted in Figure 12. The two curves exhibit good agreement, with the measurements being marginally superior to the simulations. Specifically, although the simulated S11 rises slightly above −10 dB in the 3.25–3.46 GHz range, the measured S11 remains below −10 dB; consequently, the antenna achieves an IMBW of 0.71–3.89 GHz, corresponding to a relative bandwidth of 138.2%.

3.3. Pattern Measurement

The radiation pattern of the proposed crossed-dipole antenna was measured in a microwave anechoic chamber, as shown in Figure 13a. As depicted in Figure 13b, the measured peak gain exceeds 3.5 dBic and remains essentially flat above 1 GHz, while staying above 2.7 dBic below 1 GHz; the peak measured gain reaches 7.07 dBic at 3 GHz. Figure 13b also compares the simulated and measured peak gains from 0.70 to 3.50 GHz, demonstrating close agreement between the two traces.
Figure 14a–e display the two-dimensional far-field radiation patterns measured at 1.0, 1.5, 2.0, 2.5, and 3.0 GHz on the φ = 0° (XOZ) and φ = 90° (YOZ) planes, while the corresponding legend is provided in Figure 14f. With increasing frequency, the antenna gain rises, whereas the HPBW decreases. This behavior reflects the inherent properties of the antenna: as frequency rises, its electrical size (expressed in wavelengths) enlarges—from 0.302 λ at 0.98 GHz to 0.918 λ at 3.27 GHz. Additionally, although the measured and simulated radiation patterns remain broadly consistent across the band, their agreement deteriorates at higher frequencies—a divergence attributable to fabrication and assembly tolerances that become electrically significant at shorter wavelengths.
Figure 15 compares the measured and simulated axial-ratio characteristics of the proposed antenna from 0.70 to 3.50 GHz along the boresight (φ = 0°, θ = 0°), showing that the axial ratio remains below 3 dB throughout the 0.98–3.27 GHz range. Figure 16a–e show the axial-ratio characteristics measured at 1.0, 1.5, 2.0, 2.5, and 3.0 GHz on the φ = 0° (XOZ) and φ = 90° (YOZ) planes, and the corresponding legend is given in Figure 16f. The measured and simulated axial-ratio characteristics agree closely overall, although minor discrepancies appear near the upper end of the band. These deviations are attributed to fabrication tolerances, which become more significant at higher frequencies.
Table 2 compares the performance metrics of the proposed antenna with those of previously reported wideband CP antennas. By judiciously integrating several bandwidth-extension techniques, the proposed antenna achieves substantially broader IMBW and 3 dB ARBW than other previously reported ones. Although its footprint is larger than that of the design in [12] and its profile height slightly exceeds those in [7,18], the proposed antenna still surpasses these references in both the IMBW and ARBW. Accordingly, the proposed antenna delivers highly competitive bandwidth performance while retaining a reasonably compact form factor.

4. Conclusions

An ultra-wideband, LHCP crossed-dipole antenna has been designed and experimentally verified. Vertical coupling plates that form a cavity-backed ground, tight coupling between adjacent patches, and precisely tuned ground slots and parasitic patches work in concert to broaden the impedance and axial-ratio bandwidths while maintaining a low profile. Simulated results predict |S11| < −10 dB across 0.72–3.25 GHz and an axial ratio ≤ 3 dB over 0.99–3.27 GHz. Measurements confirm |S11| < −10 dB from 0.71 to 3.89 GHz (≈ 138%) and a 3 dB ARBW of 0.98–3.27 GHz (≈ 108%), with gains exceeding 3.5 dBic; these results show good agreement between simulation and experiment. They surpass previous crossed-dipole limits and demonstrate that coordinated multi-technique integration—tight coupling, parasitic patches, ground slots, and cavity-backed assistance—can overcome circular-polarization bandwidth ceilings. The compact architecture (0.302 λ × 0.302 λ× 0.129 λ) is tailored for multifunction terminals that fuse global navigation satellite services with broadband satellite communications, providing a foundation for arrays requiring stable CP radiation over wide frequency spans. Future work will refine the feed to further mitigate phase dispersion and unlock still wider axial-ratio bandwidth and beamwidth.

Author Contributions

Conceptualization, X.J. and K.M.; methodology, K.M. and L.P.; measurement and validation, Q.L. and Z.L.; writing—original draft preparation, K.M.; writing—review and editing, R.F.; funding acquisition, X.J. All authors have read and agreed to the published version of the manuscript.

Funding

This research was funded by the National Natural Science Foundation of China (Grant No. 62461012).

Data Availability Statement

The original contributions presented in this study are included in the article. Further inquiries can be directed to the corresponding author.

Conflicts of Interest

The authors declare no conflicts of interest.

Abbreviations

The following abbreviations are used in this manuscript:
CPcircularly polarized
LHCPleft-hand circularly polarized
IMBWimpedance bandwidth
ARBWaxial-ratio bandwidth
DGSdefected-ground structure
UWBultra-wideband
sim.simulated
mea.measured

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Figure 1. Configuration of the proposed antenna. (a) Top view of patch. (b) Three-dimensional view.
Figure 1. Configuration of the proposed antenna. (a) Top view of patch. (b) Three-dimensional view.
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Figure 2. Evolution stages of the proposed antenna (a) Step 1 (Antenna 1). (b) Step 2 (Antenna 2). (c) Step 3 (Antenna 3). (d) Step 4 (Antenna 4).
Figure 2. Evolution stages of the proposed antenna (a) Step 1 (Antenna 1). (b) Step 2 (Antenna 2). (c) Step 3 (Antenna 3). (d) Step 4 (Antenna 4).
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Figure 3. Performance comparison between the antennas of the four steps. (a) S11. (b) Peak gain. (c). Total efficiency. (d) Axial ratio.
Figure 3. Performance comparison between the antennas of the four steps. (a) S11. (b) Peak gain. (c). Total efficiency. (d) Axial ratio.
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Figure 4. Simulated radiation patterns of the proposed antenna: (a) 1.0 GHz, (b) 1.5 GHz, (c) 2.0 GHz, (d) 2.5 GHz, (e) 3.0 GHz.
Figure 4. Simulated radiation patterns of the proposed antenna: (a) 1.0 GHz, (b) 1.5 GHz, (c) 2.0 GHz, (d) 2.5 GHz, (e) 3.0 GHz.
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Figure 5. Simulated surface-current vectors on the patches at 1 GHz under various phases. (a) φ = 0°, (b) φ = 90°, (c) φ = 180°, (d) φ = 270°, (e) scale plate.
Figure 5. Simulated surface-current vectors on the patches at 1 GHz under various phases. (a) φ = 0°, (b) φ = 90°, (c) φ = 180°, (d) φ = 270°, (e) scale plate.
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Figure 6. Simulated surface-current vectors on the patches at 1.5 GHz under various phases. (a) φ = 0°, (b) φ = 90°, (c) φ = 180°, (d) φ = 270°, (e) scale plate.
Figure 6. Simulated surface-current vectors on the patches at 1.5 GHz under various phases. (a) φ = 0°, (b) φ = 90°, (c) φ = 180°, (d) φ = 270°, (e) scale plate.
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Figure 7. Simulated surface-current vectors on the patches at 2 GHz under various phases. (a) φ = 0°, (b) φ = 90°, (c) φ = 180°, (d) φ = 270°, (e) scale plate.
Figure 7. Simulated surface-current vectors on the patches at 2 GHz under various phases. (a) φ = 0°, (b) φ = 90°, (c) φ = 180°, (d) φ = 270°, (e) scale plate.
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Figure 8. Simulated surface-current vectors on the patches at 2.5 GHz under various phases. (a) φ = 0°, (b) φ = 90°, (c) φ = 180°, (d) φ = 270°, (e) scale plate.
Figure 8. Simulated surface-current vectors on the patches at 2.5 GHz under various phases. (a) φ = 0°, (b) φ = 90°, (c) φ = 180°, (d) φ = 270°, (e) scale plate.
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Figure 9. Simulated surface-current vectors on the patches at 3 GHz under various phases. (a) φ = 0°, (b) φ = 90°, (c) φ = 180°, (d) φ = 270°, (e) scale plate.
Figure 9. Simulated surface-current vectors on the patches at 3 GHz under various phases. (a) φ = 0°, (b) φ = 90°, (c) φ = 180°, (d) φ = 270°, (e) scale plate.
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Figure 10. Simulated axial ratio of the proposed antenna: (a) 1.0 GHz, (b) 1.5 GHz, (c) 2.0 GHz, (d) 2.5 GHz, (e) 3.0 GHz, (f) legend.
Figure 10. Simulated axial ratio of the proposed antenna: (a) 1.0 GHz, (b) 1.5 GHz, (c) 2.0 GHz, (d) 2.5 GHz, (e) 3.0 GHz, (f) legend.
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Figure 11. Components and assembled prototype of the proposed antenna. (a) Top side of the patch. (b) Bottom side of the patch. (c) Copper sheet of the cavity back. (d) Metallic cavity back. (e) Three-dimensional view of the assembled antenna.
Figure 11. Components and assembled prototype of the proposed antenna. (a) Top side of the patch. (b) Bottom side of the patch. (c) Copper sheet of the cavity back. (d) Metallic cavity back. (e) Three-dimensional view of the assembled antenna.
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Figure 12. Comparison of reflection between the measurement and simulation.
Figure 12. Comparison of reflection between the measurement and simulation.
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Figure 13. Radiation-pattern measurement setup and corresponding results for the proposed antenna: (a) antenna under test; (b) comparison of measured and simulated peak gain.
Figure 13. Radiation-pattern measurement setup and corresponding results for the proposed antenna: (a) antenna under test; (b) comparison of measured and simulated peak gain.
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Figure 14. Comparison of normalized patterns between the measurement and simulation of the proposed antenna: (a) 1.0 GHz, (b) 1.5 GHz, (c) 2.0 GHz, (d) 2.5 GHz, (e) 3.0GHz, (f) legend.
Figure 14. Comparison of normalized patterns between the measurement and simulation of the proposed antenna: (a) 1.0 GHz, (b) 1.5 GHz, (c) 2.0 GHz, (d) 2.5 GHz, (e) 3.0GHz, (f) legend.
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Figure 15. Measured versus simulated axial-ratio–frequency response of the proposed antenna.
Figure 15. Measured versus simulated axial-ratio–frequency response of the proposed antenna.
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Figure 16. Comparison of axial ratio between the measurement and simulation of the proposed antenna: (a) 1.0 GHz, (b) 1.5 GHz, (c) 2.0 GHz, (d) 2.5 GHz, (e) 3.0 GHz, (f) legend.
Figure 16. Comparison of axial ratio between the measurement and simulation of the proposed antenna: (a) 1.0 GHz, (b) 1.5 GHz, (c) 2.0 GHz, (d) 2.5 GHz, (e) 3.0 GHz, (f) legend.
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Table 1. Dimensions of the proposed antenna (unit: mm).
Table 1. Dimensions of the proposed antenna (unit: mm).
ParameterValueParameterValueParameterValue
a92.42ma30.07wa42.84
pa45.45mb11.00dl26.87
pb21.49mx18.3dg2.49
r14.45my6.86
r25.05h39.50
Table 2. Comparison between the proposed and the previously reported crossed-dipole CP antennas.
Table 2. Comparison between the proposed and the previously reported crossed-dipole CP antennas.
Ref.Antenna TypeSize (λL3) *IMBW
(S11 < −10dB) (GHz (%))
ARBW
(AR < 3dB)
(GHz (%))
Peak Gain
(dBic)
[2]magneto-electric dipole0.788 × 0.788 × 0.1432.03–3.56 (54.9)2.15–3.35 (43.6)10.3
[6]dielectric patch antenna0.965 × 0.758 × 0.2204.63–7.58 (48.3)5.15–7.36 (35.3)8.47
[7]dielectric resonator antenna0.538 × 0.538 × 0.0941.01–1.94 (63.05)1.09–1.40 (24.9)5.06
[8]crossed dipole with post fence0.41 × 0.41 × 0.130.87–3.17 (113.8)0.98–2.99 (101.3)6.2.
[12]cavity-backed crossed dipole0.240 × 0.240 × 0.0911–3.1 (102.4)1.17–2.85 (73.1)3.8
[15]cavity-backed crossed dipole0.483 × 0.483 × 0.1830.9–2.95 (106.5)1–2.87 (96.6)12
[16]cavity-backed crossed bowtie dipole0.44 × 0.44 × 0.2242.21–6.06 (93.1)2.4–6.4 (91.2)8.6
[18]two-pair crossed dipoles0.353 × 0.353 × 0.0971.06–1.89 (55.8)1–2.1 (70.9)5.2
[30]backed-plane crossed dipole0.455 × 0.455 × 0.1891.95–3.05 (44)2.2–3 (33)6.74
This workcavity-backed crossed dipole0.302 × 0.302 × 0.1290.71–3.89 (138)0.98–3.27 (108)7.07
* λL denotes the free-space wavelength associated with the antenna’s lowest operating frequency.
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MDPI and ACS Style

Mo, K.; Jiang, X.; Peng, L.; Fang, R.; Liu, Q.; Li, Z. Compact Circularly Polarized Cavity-Backed Crossed-Dipole Antenna with Ultra-Wide Bandwidth for Integrated GNSS–SatCom Terminals. Electronics 2025, 14, 3193. https://doi.org/10.3390/electronics14163193

AMA Style

Mo K, Jiang X, Peng L, Fang R, Liu Q, Li Z. Compact Circularly Polarized Cavity-Backed Crossed-Dipole Antenna with Ultra-Wide Bandwidth for Integrated GNSS–SatCom Terminals. Electronics. 2025; 14(16):3193. https://doi.org/10.3390/electronics14163193

Chicago/Turabian Style

Mo, Kunshan, Xing Jiang, Ling Peng, Rui Fang, Qiushou Liu, and Zhengde Li. 2025. "Compact Circularly Polarized Cavity-Backed Crossed-Dipole Antenna with Ultra-Wide Bandwidth for Integrated GNSS–SatCom Terminals" Electronics 14, no. 16: 3193. https://doi.org/10.3390/electronics14163193

APA Style

Mo, K., Jiang, X., Peng, L., Fang, R., Liu, Q., & Li, Z. (2025). Compact Circularly Polarized Cavity-Backed Crossed-Dipole Antenna with Ultra-Wide Bandwidth for Integrated GNSS–SatCom Terminals. Electronics, 14(16), 3193. https://doi.org/10.3390/electronics14163193

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