Design and In Vivo Measurement of Miniaturized High-Efficient Implantable Antennas for Leadless Cardiac Pacemaker

Abstract

Deeply implanted biomedical devices like leadless pacemakers require an antenna with minimal volume and high radiation efficiency to ensure reliable in-body communication and long operational time within the human body. This paper introduces a novel implantable antenna designed to significantly reduce the spatial requirements within an implantable capsule while maintaining high radiation efficiency in lossy media like heart tissue. The design principles of the proposed antenna are outlined, followed by antenna parameters and an equivalent circuit study that demonstrates how to fine-tune the antenna’s resonant frequency. The radiation characteristics of the antenna are thoroughly investigated, revealing a radiation efficiency of up to 28% at the Medical Implant Communication System (MICS) band and 56% at the 2.4 GHz ISM band. The transmission efficiency between two deeply implanted antennas within heart tissue has been improved by more than 15 dB compared to the current state of the art. The radiation and transmission performance of the proposed antennas has been validated through comprehensive simulations using anatomical human body models, phantom measurements, and in vivo animal experiments, confirming their superior radiation performance.

1. Introduction

The Leadless Pacemaker Capsule (LPC) represents a promising advancement in the field of deeply implantable medical devices, aiding in the regulation and maintenance of cardiac rhythms for patients with cardiopathy [1]. Globally, approximately 700,000 pacemaker implantations are performed annually [2]. Unlike traditional pacemakers, which require wire leads to connect various implantable capsules and a subcutaneous control device, leadless pacemakers utilize a wireless connection. This innovation not only reduces the risk of infections but also minimizes surgical invasiveness for patients [3]. As illustrated in Figure 1, optimal cardiac monitoring typically requires the implantation of at least two leadless capsules within the heart; in some cases, three deeply implanted pacemakers may be necessary. Each capsule must reliably communicate with others, either by functioning independently as a master node or through a single master capsule acting as an intermediary for slave capsules and external devices. This application requires a highly reliable in-body communication link, which demands antennas with high radiation efficiency. Moreover, longevity is a critical attribute for leadless pacemakers, which are expected to operate within the heart for periods ranging from 8 to 10 years. Consequently, minimizing the antenna’s size within the pacemaker’s limited volume is essential to maximize space for the battery. For typical antennas, the radiation efficiency decreases with the size. Thus, the development of miniaturized antennas with enhanced radiation efficiency is pivotal for the success of deeply implanted devices.

Designing the antenna for a Leadless Pacemaker Capsule (LPC) that operates within heart tissue—characterized by high permittivity and conductivity—poses significant challenges. The electrical properties of human tissues demand specialized design principles for implantable antennas, distinct from those used in free-space environments. The high conductivity of human tissues, especially water-rich ones such as the heart, blood, and muscles, significantly distorts the antenna’s field distribution. Consequently, traditional free-space antenna concepts, such as radiation patterns and efficiency, cannot be directly applied to implantable antennas. These parameters must be redefined when the antenna is embedded within a lossy medium [5,6,7,8,9,10,11]. Insulation is another critical factor in the design of implantable antennas, as the thickness and electrical properties of the insulating material significantly influence the near-field distribution [12,13,14]. Literature reviews suggest that magnetic antennas generally achieve higher radiation efficiency than electric antennas in lossy media, especially those with high conductivity. This increased efficiency is attributed to the magnetic field predominance in the near field of magnetic antennas, which results in lower thermal losses compared to electric antennas, where the electric field dominates [15,16,17].

Furthermore, the radiation pattern of an implantable antenna cannot be straightforwardly defined as it would be in free space, since it also depends on the dimensions and electrical properties of the surrounding media, not solely on the antenna itself [5]. In the context of the LPC, where intra-body communication prevails, merely illustrating the radiation pattern and gain proves insufficient. These parameters are subject to change depending on the dimensions of the surrounding medium. Consequently, it is crucial to develop methodologies for assessing the radiation performance of antennas within such environments. This paper proposes high-radiation-efficiency implantable antennas and details the methods for evaluating the radiation and transmission performance of antennas within lossy media.

Numerous conformal antennas have been developed to conserve the internal space of implantable capsules [18,19,20,21,22,23,24]. However, implementing conformal antennas on the titanium shell of LPCs presents challenges, as it can compromise the mechanical properties of titanium, which are crucial for the capsule’s longevity. As discussed, the antenna for LPCs must meet the requirements of high radiation efficiency in a lossy medium and a compact size. Therefore, miniaturized implantable antennas have been the focus of several studies [25,26,27,28,29,30,31,32,33]. For instance, in [25], a multilayer structure is employed to reduce the antenna size, resulting in a multilayer helical antenna with a 12 mm diameter, designed for 2.4 GHz ISM band ingestible capsule endoscope systems. Similarly, a folded structure [26] is used to design a compact folded antenna measuring 20.3 mm × 0.8 mm × 0.8 mm, operating within the UHF band (0.951–0.956 GHz). In [27], an electrically coupled loop antenna, which is a dual of the planar inverted-F antenna (PIFA), is proposed. This antenna is later utilized in implanted devices within the human body, proving to be a strong candidate for a miniaturized, high-efficiency radiation antenna [28]. The size of the dual-band antenna is greatly reduced by inserting multi-slots into the patch and ground plane in asymmetric patterns. For size reduction, a modified square ring with a mitered cross inside is adopted as the main radiator. By splitting one arm of the mitered cross, the compactness can be further improved while exciting the potential CP radiation [29]. In our previous work [30], we designed an implanted spiral antenna with a modified ground plane, optimized for leadless cardiac pacemakers in the MICS band. Building on that design, this paper introduces an improved model that increases the antenna’s radiation efficiency at MICS from 3.3% to 28%, along with a more than 15 dB enhancement in forward transmission.

Part of the results presented in this paper are also included in the first author’s doctoral thesis [31]. The manuscript is organized as follows: Section 2 describes the antenna design principle and studies the equivalent circuit of the proposed antenna. Section 3 outlines the methods for evaluating the radiation efficiency of antennas within a lossy medium and assesses the radiation performance of the proposed antennas by comparison with ideal small electric and magnetic antennas. Section 4 discusses the transmission performance of the antennas in the heart tissue. In vitro phantom measurement and in vivo animal experiment are carried out to verify the performance of designed antennas in Section 5.

2. Antenna Design and Equivalent Circuit

2.1. Design Principle

As illustrated in Figure 2a, the proposed antenna comprises three distinct copper layers. The fundamental working principle involves manipulating the imaginary component of the input impedance by introducing distributed capacitance and adjusting the real component via the coupling mechanism.

The top copper layer features an open-circuited low impedance transmission line, which connects to the feeding probe to facilitate coupling feed to the middle layers, as shown in Figure 2b. Additionally, the upper substrate, made of RO3003 with a dielectric constant of 3.0 and a thickness of 0.254 mm, exhibits low permittivity. These properties are intended to minimize ohmic losses and stored energy within the antenna. Consequently, the dimensions of the top layer, the thickness of the upper substrate, and its permittivity are crucial for scaling the input impedance. Between the upper substrate and the middle copper layer lies the prepreg, which has a relative permittivity of 4.4 and serves to bond the two substrates together, as shown in Figure 2a. During fabrication, pressure is applied to the prepreg to fill the gap between the upper and bottom substrates. The thickness of the prepreg after fabrication is 0.1 mm, and this dimension is incorporated into the simulation.

The middle copper layer, depicted in Figure 2c, is designed as a circular arc loop structure terminated at the bottom copper layer, effectively functioning as a small inductor. In this configuration, the inductance—represented by the imaginary part of the impedance—increases with higher operating frequencies. And the middle copper layer is fed through capacitive coupling from the top copper layer. Figure 2d presents the bottom structure of the antenna. The capacitance between the middle and bottom layers facilitates miniaturization of the antenna size; therefore, a high permittivity and thin substrate are employed, specifically RO3010 with a dielectric constant of 10.2 and a thickness of 0.127 mm. The loop formed by the middle and bottom layers serves as the radiating element of the antenna, with its radius primarily determining the resonant frequency. The width of the bottom layer, denoted as $W_b$, is adjusted to modify the distributed capacitance, thereby ensuring that the imaginary part of the impedance is zero.

Figure 3 illustrates the equivalent circuit of the proposed antenna design. In this schematic, $R_r$ represents the radiation resistance, $R_L$ denotes the loss resistance, $L_A$ is the inductance of the loop, $L_i$ indicates the inductance of the loop conductor, and $C_r$ is the parallel capacitance. Additionally, the diameter of the antenna plays a crucial role in determining the inductances $L_A$ and $L_i$.

To enable the antenna to resonate at various frequency bands, the impedance can be adjusted by modifying the dimensions of the feed head and loop line, thereby tuning $Z_{in}$ to 50 ohms. It is important to note that, in order to increase the antenna’s radiation efficiency, the lossy resistance ($R_L$) should be minimized by selecting a substrate with a low loss tangent. The radiation resistance ($R_r$) can be tuned to 50 ohms by adjusting the coupling between the top copper layer and the middle coupling layer.

Furthermore, the parallel capacitance ($C_r$) can be controlled by choosing an appropriate substrate thickness and adjusting the overlap size between the bottom copper layer and the middle layer. As a type of loop antenna, the inductances ($L_A$ and $L_i$) are significantly affected by the antenna’s size. Further details will be discussed in the following sections.

2.2. Discussion and Simulation

In the initial stages of the antenna design, simulations were conducted with the antennas encapsulated in a 0.5 mm thick layer of Polymethyl methacrylate (PMMA), a biomedical material. These were then implanted within a spherical heart phantom, with a diameter corresponding to the effective wavelength. The electrical properties of the heart phantom are frequency-dependent and were assigned based on Gabriel’s four Cole–Cole models [32]. For this particular antenna design, critical parameters include not only the diameter but also the central angle of the circular lines on the three copper layers. Initial investigations have focused on the effects of the width and central angle of the circular line of the top layer on the antenna’s input impedance. Figure 4 illustrates how the simulated S11 and input impedance vary with the radius angle. As previously discussed, the primary role of the top layer is to establish a coupling feed structure, enhancing the real part of the antenna’s impedance to achieve 50 Ohms. Figure 4b reveals that as the central angle of the circular line on the top copper layer decreases, the real part of the input impedance remains relatively constant, while the zero-crossing points of the imaginary part’s curves shift slightly towards higher frequencies, resulting in a deterioration of the S11. This observation suggests that the coupling feed structure, while increasing the radiation resistance, $R_r$, also introduces a subtle capacitive effect to the antenna.

Figure 5a clearly demonstrates that the resonant frequency shifts to a higher frequency as the central angle of the bottom layer decreases, a trend also observable in the impedance characteristics depicted in Figure 5b. The primary function of the bottom layer is to introduce a parallel distributed capacitor, thereby shifting the zero-crossing point of the imaginary part of the impedance curve to a lower frequency. Additionally, the widths of the circular lines across the three copper layers influence the resonant frequency. A decrease in width leads to a reduction in the capacitance of the distributed capacitor between the bottom and middle layers, consequently causing the resonant frequency to shift to a higher frequency, as shown in Figure 6. From Figure 6a, it is also evident that the resonance quality deteriorates slightly with the reduction in width. This effect arises because the width of the circular line affects the coupling between the top and middle layers.

The proposed design concept has been successfully implemented to model antennas operating across various potential frequency bands for a leadless pacemaker. Figure 7 depicts four distinct antennas tailored to function at the Medical Implant Communication System (MICS); Wireless Medical Telemetry Service (WMTS); Industrial, Scientific, and Medical (ISM) 902 MHz; and 2.4 GHz frequency bands. These antennas have diameters of 12 mm, 8.6 mm, 6.6 mm, and 5 mm, which correspond to 0.04, 0.044, 0.05, and 0.1 times the effective wavelength, respectively. Figure 8 presents the simulated reflection coefficients for these configurations. As previously discussed, the diameter of the antenna primarily dictates its resonant frequency. Additionally, the widths and curves of the copper circular line are strategically employed to fine-tune this resonant frequency. Owing to the antenna’s simplistic structure, adjusting its resonance to align with different frequency bands is straightforward.

3. Radiation Performance of the Proposed Antenna

3.1. Radiation Pattern

To evaluate the radiation pattern of the proposed antenna, the surrounding medium is configured as depicted in Figure 9. This medium is a sphere with a radius equivalent to the effective wavelength and is filled with homogeneous heart tissue. The homogeneous heart tissue used in simulations had the following electrical parameters at 403 MHz: relative permittivity εr = 66.0 and conductivity σ = 0.966 S/m. At 2.4 GHz, the parameters are: εr = 54.9 and σ = 2.22 S/m. The normalized radiation pattern, illustrated in Figure 10, resembles an unbalanced doughnut. This asymmetry arises because the loop portion of the proposed antenna, comprising the middle and bottom layers as shown in Figure 2, is not symmetrical. The antenna is immersed in a lossy medium; consequently, its gain is influenced not solely by the antenna’s properties but also by the dimensions and loss characteristics of the surrounding medium [33,34,35]. As the size of the phantom increases, the gain of the antenna decreases. Therefore, the conventional definition of antenna gain in free space cannot be directly applied to antennas operating within a lossy medium.

3.2. Radiation Efficiency

Radiation efficiency is a far-field concept that denotes the ratio of the power radiated into the far-field zone to the power inputted into the antenna. For an antenna operating in free space, its radiation efficiency is primarily influenced by structural losses, which include mismatch, ohmic, and dielectric losses of the antenna itself. However, for an antenna operating in a lossy medium, the high conductivity of the surrounding medium affects the radiation efficiency. In this case, the efficiency is influenced not only by the mismatch, ohmic, and dielectric losses of the antenna structure but also by the near-field loss caused by the coupling between the electric field and the conductivity of the surrounding medium.

We compared the radiation efficiency of the proposed antenna with that of ideal electric and magnetic dipoles. At 403 MHz and 2.4 GHz, the radii of the ideal electric and magnetic dipoles are 6 mm and 2.5 mm, respectively, to match those of the designed antennas, and the material used is a perfectly electrically conducting material. In the full-wave simulation, the radiated power of the proposed antenna is determined by numerically integrating the Poynting vector over the spherical surface surrounding the antenna. Figure 11 presents the comparison results of the radiated power for the proposed antennas, ideal electric dipole, and magnetic dipole. In Figure 11a, at 403 MHz, the radiated power of the ideal magnetic dipole shows an approximate 5.8 dB improvement over the ideal electric dipole with a 6.5 mm lossless sphere. The discrepancy between the radiated power of the proposed antenna and the ideal magnetic dipole is less than 2.7 dB, primarily due to the loss introduced by the insulation layer. Figure 11b illustrates the comparison results at 2.4 GHz, indicating that as the frequency increases, the improvement of the ideal magnetic dipole becomes less pronounced, around 1 dB.

By compensating the radiated power with $e^{2\alpha r}$, the radiation efficiency of the proposed antenna can be determined. The parameter $\mathrm{\alpha }$ represents the attenuation factor, which characterizes the rate of electromagnetic wave energy loss during propagation through the medium, r denotes the propagation distance of electromagnetic waves radiated by the antenna in lossy media, typically taken as the radius to any spherical surface in the far-field region. In traditional free space, antenna radiation efficiency can be directly calculated by integrating the Poynting vector over a closed far-field surface. However, in lossy media, far-field power attenuates with distance, resulting in inconsistent integration results at different distances. By introducing the factor $e^{2\alpha r}$, we compensate for the radiated power at any far-field spherical surface, ensuring the compensated power remains constant regardless of distance. This enables accurate extraction of the antenna’s intrinsic radiation power and subsequent calculation of precise radiation efficiency.

For the proposed antenna operating at 403 MHz, the calculated radiation efficiency is approximately 28%, compared to 49% for an ideal magnetic dipole of the same size, due to structural and near-field losses. At 2.4 GHz, the radiation efficiency of the proposed antenna is approximately 56%, compared to 79% for an ideal magnetic dipole of the same size. The electrical size of the proposed antennas operating at 403 MHz and 2.4 GHz is 0.07${\lambda }_{eff}$ and $0.16{\lambda }_{eff}$, respectively. ${\lambda }_{eff}$ refers to the effective wavelength in heart tissue, calculated as ${\lambda }_{eff}={\displaystyle \frac{{\lambda }_0}{\sqrt{{\epsilon }_r}}}$, where ${\lambda }_0$ is the free-space wavelength and ${\epsilon }_r$ is the relative permittivity of heart tissue. The value of ${\lambda }_{eff}$ ssat 403 MHz and 2.4 GHz are 91.5 mm and 16.5 mm, respectively.

The larger electrical size of the proposed antenna operating at 2.4 GHz results in higher radiation efficiency than the antenna operating at 403 MHz.

3.3. Specific Absorption Rate

The SAR (Specific Absorption Rate) distribution of the proposed antenna in heart tissue was investigated through simulation at 403 MHz and 2.4 GHz. As shown in Figure 12, the SAR distribution on the cross-section of the model is concentrated in the near-field region of the antenna, with SAR values gradually attenuating as the distance from the antenna increases. According to IEEE/IEC standards, the safety threshold for local SAR averaged over 10 g of tissue is 2 W/kg, and human tissue is considered safe when the induced SAR values are below this safety guideline at the prescribed transmission power of 20 mW.

At 403 MHz, the simulated SAR distribution is concentrated around the implanted antenna with a maximum value of 0.920 W/kg, which is substantially below the safety threshold, demonstrating compliance with safety regulations for implantable medical devices. The SAR field exhibits an elliptical distribution pattern, with the highest intensity region (shown in red) closely surrounding the antenna structure. SAR values decrease rapidly beyond the immediate vicinity of the antenna, with the outermost blue region showing values below 0.093 W/kg, indicating effective power dissipation within the heart tissue. This SAR distribution pattern suggests that electromagnetic energy is effectively confined to the local region around the antenna. At 2.4 GHz, the SAR distribution demonstrates a more compact and localized pattern compared to the 403 MHz case, with a peak SAR value of 0.622 W/kg, maintaining values significantly below the regulated limits throughout the tissue volume and complying with safety standards. The SAR field exhibits a more uniform and symmetrical distribution around the antenna, with concentric regions of decreasing intensity from the center. The outer field extends approximately the same distance as the 403 MHz case, but with a more rapid decay rate, which is characteristic of higher frequency operation in lossy media.

4. Transmission Performance of the Proposed Antenna

The evaluation of the transmission coefficient between two antennas deeply implanted within both a homogeneous phantom and an anatomical human model is detailed herein. Figure 13 illustrates the simulation setup used to evaluate the transmission coefficient between two antennas embedded in a homogeneous medium. The simulation was conducted with Ansys HFSS 2023. Each antenna maintains a distance of one effective wavelength from the surface of the phantom sphere, with the distance between the antennas varying. The boundary conditions were defined as radiation boundaries, making the simulation domain equivalent to the size of the spherical heart tissue. This configuration mitigates the impact of boundary-related calculation inaccuracies. The electric properties of the homogeneous phantom, which are frequency-dependent, closely resemble those of heart tissue.

As depicted in Figure 14, to assess the transmission coefficient within the anatomical human model, two antennas are embedded deep within the human heart. This simulation was conducted with CST Microwave Studio, and the anatomical human model used is a female model named Nelly. This arrangement models the operational context of a leadless pacemaker capsule, with one antenna fixed at the cardiac apex and the other adjustable to vary the interspatial distance. The antenna was positioned inside the heart, which was surrounded by various tissues, including muscle, lungs, and bone. The boundary conditions in CST were defined as open boundaries, with the simulation background set to air. This configuration closely mimics the realistic environment in which the implanted antennas operate.

Figure 15 illustrates the comparative analysis of transmission coefficients for two antennas, specifically designed and deeply implanted, within both homogeneous heart tissue and an anatomical human model at frequencies of 403 MHz and 2.4 GHz. For the antenna operating within the MICS band, the simulated S21 (transmission coefficient) in the anatomical human model aligns closely with that observed in the homogeneous tissue model. As the distance increases, the S21 value in the anatomical model exceeds that in the homogeneous tissue. This discrepancy is attributed to reflections and scattering caused by the complex internal structure and organs of the human body, particularly when the antenna is positioned near the heart wall. Regarding the antenna designed for the 2.4 GHz ISM band, the trend of the transmission coefficients in the anatomical model mirrors that observed in the homogeneous tissue. However, slight variations in the values occur due to misalignment between the two antennas, which notably affects transmission in the higher frequency band. This impact is compounded by higher path loss at elevated frequencies.

5. In Vitro and In Vivo Measurement

The fabricated antennas operating within the MICS and 2.4 GHz ISM bands are shown in Figure 16a. It is important to note that the performance of these antennas can be influenced by the coaxial cable connected to them, as documented in reference [30]. Consequently, as depicted in Figure 16b, efforts have been made to minimize the length of the coaxial cable to mitigate its impact on antenna performance. The antennas were tested using a liquid phantom contained within a box designed to simulate the electrical properties of heart tissue [36].

For the fabrication process, traditional tin soldering was employed to connect the inner conductor of the coaxial cable to the top layer of the antenna and the outer metal shield to the bottom layer. By performing measurements using a standard-calibrated vector network analyzer (VNA), Figure 17 illustrates the comparison between the measured and simulated values of |S11|. As observed in the figure, the measured |S11| for the antenna (12 mm diameter) operating within the MICS band aligns closely with the simulated results. However, for the 2.4 GHz ISM band antenna with its considerably smaller dimensions (5 mm diameter), the manufacturing challenges are more pronounced. The post-fabrication measurements revealed that the actual manufactured antenna was approximately 0.5 mm larger than the designed dimensions, which explains the observed downward frequency shift in Figure 17b.

As previously discussed, the transmission coefficients between antennas of different deeply implanted pacemaker capsules play a crucial role in assessing the intra-heart communication links. In the experimental setup, to mitigate the coupling effect inherent between the antenna feeding cables, the commercial transceiver chips TI CC1101 and TI CC2500 were employed, as illustrated in Figure 18a,b. These chips operate within the MICS band and the 2.4 GHz ISM band, respectively. Both transceivers’ evaluation boards feature input and output impedances of 50 Ohms, allowing for the direct connection of the custom-fabricated antennas, which are soldered to short feeding coaxial cables. The transmission power settings range from −30 dBm to 10 dBm, facilitating the detection of the Received Signal Strength Indicator (RSSI) values.

For the measurement of in-body to in-body transmission, two identical antennas, integrated with the transceivers, were utilized. As shown in Figure 18c, the two implantable antennas were immersed inside the phantom. One was maintained in a relatively fixed position, while the other was manually adjusted along a line of sight to document the coupling between the antennas at various separation distances within a homogeneous phantom. The experimental setup facilitated the recording of received power as a function of the distance between the two in-body antennas. Figure 19 illustrates a comparison between the simulated and measured results at 403 MHz and 2.4 GHz during in-body to in-body communication.

For the MICS band antennas, there is a strong concordance between the measured and simulated results, despite a minor disparity. The slopes of the S21 curves at 403 MHz, for both measured and simulated data, are nearly identical. In the case of the 2.4 GHz ISM band antennas, the slopes of the measured and simulated curves also exhibit a resemblance; however, the discrepancies are more pronounced compared to those observed in the MICS band antennas. Several factors contribute to this disparity: (a) the structural losses in the fabricated antenna are more substantial than those predicted by simulations, particularly in high frequency bands such as the 2.4 GHz ISM band; (b) the received signal strength indicator (RSSI) is measured not directly from the antenna port but from the input port of the CC1101, thereby introducing losses attributable to the strip line, SMA connector, and coaxial cable; (c) the conductivity of the phantom slightly exceeds that of heart tissue, which may affect the results; and (d) the S11 in the 2.4 GHz ISM case does not closely match the simulation.

The in vivo experiment was conducted at the Intervention Centre, Rikshospitalet, Oslo University Hospital, Norway, which holds full authorization to carry out animal research. For the animal experiment, a 65 kg pig was used under normal room temperature conditions (approximately 25 degrees Celsius), and it was under general anesthesia during the clinical experiment. The in vivo experiment and animal treatment are performed per the strict clinical and ethical standards defined according to European Union and Norwegian laws. Throughout the duration of the experiment, the pig remained alive with all organs functioning normally.

Identical antennas were implanted for both transmitting and receiving purposes. The transmitting antenna was securely affixed to the lower apex of the heart using surgical thread, while the position of the receiving antenna was systematically altered in a clockwise direction around the heart. After recording the data at one location, the receiving antenna was moved to the next to characterize the transmission link across different paths. This adjustment created varying distances between the transmitter and receiver, as depicted in Figure 20. Signal strength was assessed under two different conditions: with the chest open and with the chest closed—the latter scenario simulating the practical application for implanted devices. The distance between the transmitting and receiving antennas was manually estimated using a ruler.

Figure 21 presents both the simulated and measured results, illustrating a decrease in signal strength with increasing distance. Notably, the signal strength in closed chest measurements closely aligns with those conducted with an open chest, as depicted in the same figure. However, minor discrepancies between the simulated and measured results are evident. These discrepancies are likely attributable to inaccuracies in distance measurement and the dynamic nature of the experiment involving a real, beating heart. The motion associated with the heart’s expansion and contraction during pumping may cause slight shifts in the positions of the transmitting and receiving antennas.

6. Analysis and Discussion

Table 1. Comparison of the proposed antennas with other reported antennas in the literature.

ISM band	Ref	Vol. (mm3)	Eff. %	Depth (mm)	Measured Atten. in-Body to in-Body (IB2IB) (dB)	Tissue	Band Width (MHz)
	[37]	24	N/A	N/A	N/A	Skin	100
	[38]	43.15	N/A	N/A	N/A	Skin	200
	[39]	248.92	N/A	N/A	N/A	Skin	240
	[40]	6	0.1	N/A	N/A	Heart	N/A
	[41]	80	1	35	N/A	2/3 Muscle	40
	[42]	48.6	N/A	4	N/A	Heart	134
	T.W	6.08	56	100	60@ 50 mm	Heart	100
	[43]	193.2	0.44	3	N/A	Skin	20
MICS band	[44]	2356.5	N/A	50	N/A	Muscle	50
	[45]	479	N/A	3	N/A	Muscle	35
	[46]	642.62	N/A	4	N/A	Skin	50
	[47]	365	N/A	N/A	N/A	Head	30
	[48]	967.74	N/A	1	N/A	Muscle	20
	[49]	204.8	N/A	14	N/A	Brain	40
	[50]	197.04	N/A	4	N/A	Head	70
	[51]	797.96	N/A	15	N/A	Muscle	96
	[52]	47.7	0.19	55	N/A	Heart	84
	P.W [30]	22.20	3.3	100	50@50 mm	Heart	30
	T.W	45.24	28	100	32@50 mm	Heart	30

provides a comprehensive comparison of the performance of our innovative antenna design with several antennas reported in the recent literature, highlighting its notable advantage in terms of volume and radiation efficiency inside the lossy medium. Most studies primarily report gain and in-body to off-body transmission; however, in-body to in-body transmission is rarely addressed. The gain of an implantable antenna is highly dependent on its implantation site, as factors such as tissue composition, body shape, and size significantly affect antenna performance. Therefore, comparing the gain of antennas implanted in different body locations may not yield meaningful results. In this paper, the radiation performance of our antenna is validated through both simulated and measured transmission coefficients in homogeneous tissue, an anatomical human model, and a living animal.

7. Conclusions

In conclusion, we propose a novel design principle for implantable antennas in high-loss environments, addressing the need for miniaturization and high radiation efficiency, particularly for leadless pacemakers. The design, based on an equivalent circuit model, allows easy tuning to different resonant frequencies. Antennas operating in the MICS, WMTS, 902 MHz ISM, and 2.4 GHz ISM bands were developed using this straightforward and adaptable approach. A comparison with ideal magnetic and electric Hertzian dipoles demonstrates the superior transmission performance and high radiation efficiency of the proposed antenna. At 403 MHz, the calculated radiation efficiency is approximately 28%, while at 2.4 GHz, it reaches 56%, significantly outperforming existing designs. The transmission efficiency between two deeply implanted antennas within heart tissue has been improved by more than 15 dB compared to the current state of the art [29]. The resonant frequency and transmission performance of the antenna were verified with in vitro and in vivo experiments.