Non-Isolated Current-Fed Series Resonant Converter with Hybrid Control Algorithms for DC Microgrid

Abstract

Module-level power electronics can optimize the power generation of photovoltaic panels, which requires a simple topology structure, high power conversion efficiency, and wide voltage gain, corresponding to irradiance variations. This paper proposes a non-isolated current-fed L-C series resonant converter and its control algorithms. The proposed converter can obtain a buck–boost voltage gain with the hybrid control algorithm utilizing the switching frequency modulation and asymmetric pulse width modulation. The proper power stage design can guarantee the soft-switching capability for the entire operating range. Also, the control strategy is designed to obtain a small rms current. The operational principle of proposed converter is analyzed in order to design the resonant tank and control strategy. The performance of designed converter can be verified with the experimental results with a 500 W prototype converter.

1. Introduction

The limited availability of fossil fuels and their significant contribution to greenhouse gas emissions have presented humanity with a critical issue [1]. The evidence of climate change is becoming more apparent, with escalating global temperatures, extreme weather occurrences, and ecological disturbances impacting ecosystems and human welfare. To address these pressing issues, it is imperative to shift towards cleaner and renewable energy sources that effectively reduce our carbon footprint. Solar power, derived from the virtually unlimited energy emitted by the sun, emerges as a promising solution to this pressing challenge.

Solar power offers numerous advantages as a compelling source of renewable energy [2]. It is abundant and virtually inexhaustible, providing a sustainable solution to meet our growing energy demands. Solar energy generation produces no greenhouse gas emissions or air pollutants, improving air quality and mitigating the adverse effects of climate change. Additionally, solar power’s decentralization and versatility enable distributed energy generation, enhancing energy independence, and reducing reliance on centralized grids. Its scalability and modular nature allow for deployment across various scales and applications, catering to diverse energy needs.

Solar power, despite its advantages, faces limitations that must be considered. The intermittent nature of solar energy, dependent on sunlight availability, poses challenges for meeting continuous energy demands. Energy storage technologies, like batteries, are essential for overcoming this limitation. Additionally, utility-scale solar installations require significant land area, potentially competing with other land uses. Careful planning is necessary to mitigate conflicts. The upfront cost of solar power systems can be substantial, although long-term financial benefits, such as reduced energy costs and potential revenue generation, should be acknowledged.

In terms of power generation, PV systems have a wide variation in the power generation according to the solar irradiances [3]. Especially, the partial shading among PV panels can degrade the total power generation [4]. The conventional string and central inverters are affected by the shading of solar panels in terms of power conversion, which causes a large loss in power generation from a long-term perspective. To solve the partial shading problem, the PV systems using the module level power electronics (MLPE) was introduced [5,6,7,8]. Figure 1 shows the example of PV system structure with MLPE. The MLPE implements the maximum power point tracking (MPPT) for each PV panel, which can generate the maximum power of PV systems at the partial shading. The MLPE requires the wide voltage gain according to the solar irradiance, because the irradiance change induces the maximum power point variation at the PV V-I curve. Also, the MLPE requires the soft-switching operation and small rms current to minimize the power losses on the power converter.

In the previous research, various MLPEs were introduced to overcome the partial shading issue. In [9,10,11,12,13,14,15], the non-isolated buck and boost converters were introduced for the MLPE. This is a simple and basic structure in the DC/DC converter and has high cost-effectiveness. In [16,17,18], the non-inverting buck–boost converter was developed with wide voltage gain compared with buck and boost converters. In [19,20,21], isolated-type DC/DC converters were introduced to achieve the high voltage gain and galvanic isolation. The resonant type converters were introduced to obtain the soft-switching capability [22,23,24]. Previous research is focused on the power loss reduction with soft switching capability and wide voltage gain with a wide PV panel voltage.

In the resonant converters, the current-fed series resonant converter was introduced to obtain the wide voltage gain and soft-switching capability. In [25], the hybrid modulation, using the pulse width modulation (PWM) and phase shift modulation (PSM), was applied to reduce the rms current. In [26], hybrid PWM modulation was invented to reduce the current ripple and bidirectional power flow. In [27], a PWM modulation and auxiliary inductor was utilized to implement the bidirectional power flow with a soft switching capability. In [28], the PSM between primary switching legs and frequency modulation (FM) was utilized to implement wide voltage gain and soft-switching capability. Previous research revealed high power conversion efficiency based on the soft-switching capability and small rms current. However, the complex topology is proper for the high power applications. It is not appropriate for MLPE application, since it requires a simple structure for high cost-effectiveness and high power density.

In this paper, a non-isolated current-fed series resonant converter is proposed in order to achieve the buck–boost voltage gain, soft switching capability, and small rms current with the hybrid modulation based on the asymmetric PWM and FM. Figure 2 shows the topology structure of non-isolated current-fed series resonant converter. The operational principle of the current-fed series resonant converter is analyzed with the designed control algorithms. The power stage and control algorithm are designed to achieve the desired voltage gain, soft switching for the entire operating condition, and small rms current. The experimental results verify the performance of the proposed method with a 500 W prototype converter.

2. Analysis of Proposed Converter

The non-isolated current-fed series resonant converter, as shown in Figure 2, is integrated with the boost converter and series resonant converter. The boost operation includes Q1 and Q2 power switches, boost inductance (L_in), and link capacitance (C_link). The resonant operation includes the Q1 and Q2, C_link, resonant inductance (L_r), capacitance (C_r), rectifying diodes (D1, D2), and output capacitance (C_o). In the steady state condition, it has four operating modes, as shown in Figure 3 and Figure 4.

2.1. Mode 1 (t₀ − t₁)

The Mode 1 is the turn-on duration of the power MOSFET Q2, which is described in Figure 3a. The PV voltage (V_s) and boost inductance increase the input current (i_in), as follows:

$$i_{in}\left(t\right)=I_{in}\left(t_o\right)+\frac{V_s}{L_{in}}t$$

(1)

where I_in(t₀) is the input current at t₀. The resonant current (i_r) flows through the L_r, C_r, Q2, and D2, which can be derived as follows:

$$i_r\left(t\right)=I_r\left(t_o\right)\cos {\omega }_s\left(t-t_0\right)-\frac{v_{cr}}{Z_r}\sin {\omega }_s\left(t-t_0\right)$$

(2)

where I_r(t₀) is the resonant current at t₀, v_cr is the voltage of the resonant capacitor, Z_r is the impedance of resonant tank, and ω_s is the switching angular frequency. This duration does not transfer the power to the output load.

2.2. Mode 2 (t₁ − t₂)

Mode 2 is the turn-off duration of the power MOSFET Q1 and Q2, which are described in Figure 3b. The input and resonant current charge the link capacitor (C_link). Also, they can discharge the parasitic capacitor (C_p) of Q1, which can achieve the zero voltage switching (ZVS) condition. After the discharge of C_p, the current flows through the junction diode of Q1.

2.3. Mode 3 (t₂ − t₃)

Mode 3 is the turn-on duration of power MOSFET Q1, which is described in Figure 3c. The relationship between PV and DC link voltage (V_link) can decrease the input current, which becomes a negative value to achieve the ZVS condition of Q2. The input current can be derived as follows:

$$i_{in}\left(t\right)=I_{in}\left(t_2\right)+\frac{V_s-V_{link}}{L_{in}}t$$

(3)

where I_in(t₂) is the input current at t₂, and V_link is the DC link capacitor voltage. The resonant current flows through the L_r, C_r, D1, R_o, and C_link, which can be derived as follows:

$$i_r\left(t\right)=I_r\left(t_2\right)\cos {\omega }_s\left(t-t_2\right)+\frac{V_s-v_{cr}-V_o}{Z_r}\sin {\omega }_s\left(t-t_2\right)$$

(4)

where I_r(t₂) is the resonant current at t₂. This duration transfers the power to the output load.

2.4. Mode 4 (t₃ − t₄)

Mode 4 is the turn-off duration of the power MOSFET Q1 and Q2, which are described in Figure 3d. The sum of input and resonant current is the current of Q2. The negative current of Q2 is able to fully discharge the parasitic capacitor of Q2. After the discharge of C_p, the input current flows through the junction diode of Q2. Mode 4 shows the negative current of the input inductance, which contributes to achieving the zero voltage condition (ZVS) of Q2. In Mode 4, the PV voltage and current are positive and negative, respectively. Therefore, the negative current induces the reactive power, which does not transfer the power to the load.

3. Design Methodology

The proper resonant tank and input inductance designs can obtain the desired voltage gain and ZVS condition. Also, they utilize the hybrid control algorithm of the asymmetric PWM and FM. The proper duty ratio and switching frequency can obtain the minimum input and resonant currents, which increases the power conversion efficiency. The specifications of target PV systems are shown in

Table 1. Specification of PV and MLPE.

Parameter		Value
PV voltage	Open circuit	49.5 V
	MPP	15–42.1 V
Maximum Power of PV		455 W
Output Voltage of MLPE		<80 V
Maximum short circuit current		15 A
Maximum efficiency		97.5%

.

3.1. Control Algorithm

The proposed converter utilizes the hybrid control algorithm with asymmetric PWM and FM. Those two modulations have different voltage gains in the proposed converter. In the resonant tank, the switching frequency variation changes the impedance of the resonant tank, which induces the voltage gain variation. The voltage gain from V_link to V_o can be derived as follows:

$$G_{FM}\left({\omega }_s\right)=\frac{V_o}{V_{link}}=\left|\frac{R}{R+j{\omega }_s{L}_r+1/j{\omega }_s{C}_r}\right|$$

(5)

Figure 5a shows the voltage gain using (5). It has same voltage gain of a conventional series resonant converter. The proposed converter operates at an inductive impedance area. At the series resonant frequency (f_r), the converter has unified voltage gain. A switching frequency that is higher than f_r decreases the voltage gain. The voltage gain from V_s to V_link has a unified voltage gain using FM.

The current-fed structure operates as a boost converter between V_s and V_link. The voltage gain from V_s to V_link using an asymmetric duty ratio can be derived as follows:

$$G_{CF}(D)=\frac{V_{link}}{V_s}=\frac{1}{1-D}$$

(6)

where D is the duty ratio of Q2. In the resonant tank, the voltage gain using the asymmetric duty cycle can be derived as follows:

$$G_{RT}\left(D\right)=\frac{V_o}{V_{link}}\cong \frac{R_o}{Z_r+R_o}$$

(7)

From (7), the resonant tank cannot regulate the output voltage using asymmetric PWM. From Equations (5)–(7), the total voltage gain using the FM and asymmetric PWM can be derived as follows:

$$G_T\left(D,{\omega }_s\right)=\frac{V_o}{V_s}=\frac{1}{1-D}·\left|\frac{R}{R+j{\omega }_s{L}_r+1/j{\omega }_s{C}_r}\right|$$

(8)

Figure 5b shows the total voltage gains. A buck–boost voltage gain for the wide input voltage variation of PV panel is obtained.

3.2. Soft Switching Capability

The proper power stage design can achieve zero voltage switching (ZVS) capability for Q1 and Q2. The ZVS condition can be achieved with the discharge of the parasitic capacitor. Also, it requires the current flow of junction diode for the dead time duration. Mode 2 and Mode 4 show the theoretical waveforms for the dead time duration. The switch currents of Q1 and Q2 can be derived through input and resonant currents. The ZVS condition of Q1 can be analyzed via Mode 2 operation. For this duration, the input and resonant currents can be derived as follows:

$$i_{in}\left(t_1\right)=\frac{V_s}{(1-D{)}^2{R}_{eq}}+\frac{V_{s}D{T}_s}{2L_{in}}$$

(9)

$$i_r\left(t_1\right)\cong \frac{v_{cr}}{Z_r}\sin {\omega }_s\left(t_1-t_0\right)$$

(10)

where R_eq is the equivalent resistance and T_s is the switching period. Using Equations (9) and (10), the switch current at Mode 4 can be derived as follows:

$$i_{Q1}\left(t_1\right)\cong i_{in}\left(t_1\right)+i_r\left(t_1\right)$$

(11)

The ZVS condition of Q2 can be obtained with the analysis of Mode 4 duration. For this duration, the input and resonant currents can be derived as follows:

$$i_{in}\left(t_3\right)=\frac{V_s}{(1-D{)}^2{R}_{eq}}-\frac{V_{s}D{T}_s}{2L_{in}}$$

(12)

$$i_r\left(t_3\right)\cong \frac{V_s-v_{cr}-V_o}{Z_r}\sin {\omega }_s\left(t_3-t_2\right)$$

(13)

Using Equations (12) and (13), the switch current at Mode 4 can be derived as follows:

$$i_{Q2}\left(t_3\right)\cong i_{in}\left(t_3\right)+i_r\left(t_3\right)$$

(14)

The ZVS condition requires the proper current and dead time duration to discharge the parasitic capacitor. Using Equations (11) and (14), the ZVS condition can be derived as follows:

$$I_{Q1}\ge \frac{2V_{link }{C}_p}{t_{dt}},\left|I_{Q2}\right|\ge \frac{2V_{link}{C}_p}{t_{dt}}$$

(15)

where C_p is the drain-source capacitor of MOSFET. When the dead time duration is the fixed value, the proper switch current can be obtained via Equation (15) to discharge the parasitic capacitor of Q1 and Q2.

The switch currents flow through the junction diode of Q1 and Q2 before the turn-on state in order to achieve the ZVS condition. For Mode 2 and Mode 4, the negative current of Q1 and positive current of Q2 cannot achieve the ZVS condition. The switch currents at t₂ and t₄ are the worst conditions for the ZVS condition, which can be derived as follows:

$$I_{Q1}\left(t_4\right)=I_{in,max}+\frac{V_s-V_{link}}{L_{in}}{t}_{dt}+I_{r,max}\sin \left(\omega t_{dt}-\phi \right)$$

(16)

$$I_{Q2}\left(t_2\right)=I_{in,min}+\frac{V_s}{L_{in}}{t}_{dt}+I_{r,max}\sin \left(\omega t_{dt}-\phi \right)$$

(17)

$$I_{r,max}=\frac{2\pi I_o}{\left[1-\cos \left(2\pi D\right)\right]}, \phi ={\mathrm{t}\mathrm{a}\mathrm{n}}^{-1}\left(\frac{X_{Lr}-X_{Cr}}{R_o}\right)$$

(18)

where I_in,max and I_in,min are same current values of Equations (9) and (12), respectively, and I_o is the load current. Using Equations (16) and (17), the switch currents of Q1 and Q2 can be described according to the input inductance, as shown in Figure 6a. Their operating conditions are the symmetric duty ratio, resonant switching frequency, and rated load. The Q2 has a worse ZVS condition compared with the Q1, because the I_Q2(t₂) is a subtraction of I_in(t₂) and I_r(t₂). Therefore, the input inductance should be designed with the ZVS condition of Q2, which can be derived through Equation (17), as follows:

$$L_{in}\le \frac{\left(V_{s}D{T}_s/2\right)-V_s{t}_{dt}}{\left[\frac{V_s}{R(1-D{)}^2}\right]+I_{r,max}\sin \left({\omega }_r{t}_{dt}-\phi \right)}$$

(19)

Figure 6b,c show the proper input inductance to guarantee the ZVS condition, depending on the duty ratio and switching frequency, respectively. The small duty ratio of Q2 reduces the current ripple, which requires the smaller input inductance to achieve the ZVS condition. The high switching frequency operation reduces the input current ripple. However, the resonant current is phase-shifted with inductive resonant impedance, which can compensate the input current ripple reduction. From the analysis of input and resonant currents, the input inductance can be designed at the worst ZVS condition.

3.3. Analysis of RMS Current

The resonant impedance is designed to obtain the desired resonant frequency and voltage gain. The input inductance is designed to guarantee the ZVS condition for Q1 and Q2. From the designed passive components, the RMS value of input and resonant currents can be derived as follows:

$$I_{in,rms}=\sqrt{I_{in,\min }{}^2+I_{in,\min }\left(\frac{V_s}{L_{in}}\right)D{T}_s+\frac{1}{3}{\left(\frac{V_s}{L_{in}}\right)}^2{\left(D{T}_s\right)}^2}$$

(20)

$$I_{r,rms}=\frac{2\pi I_o}{\sqrt{2}[1-\cos (2\pi D\left)\right]}$$

(21)

Figure 7 shows the RMS value of input and resonant currents under rated and light load conditions according to the duty ratio and switching frequency. The RMS value of input current decreases with the high switching frequency and small duty ratio. Also, the RMS value of resonant current decreases with the large duty ratio. The proposed converter utilizes the hybrid control method using asymmetric PWM and FM. The increase in switching frequency is effective for obtaining the buck voltage gain and reduce the input RMS current at the fixed duty ratio. The increase in the duty ratio of Q2 is effective for obtaining the boost voltage gain and reducing the resonant RMS current at the fixed switching frequency. Figure 8 shows the simple control strategy of proposed converter, which is appropriate for obtaining the small input and resonant currents.

3.4. Design Example

The impedance of the LC series resonant tank can be designed with the maximum switching frequency and minimum voltage gain. The voltage gain can be rearranged with the normalized frequency and quality factor as follows:

$$G_{FM}\left(f_n\right)=\frac{j{f}_n}{-Q{f}_n^2+Q+j{f}_n}$$

(22)

where f_n is the normalized frequency (f_s/f_r) and Q is the quality factor. The desired voltage gain range is unity to three. Also, the resonant frequency and maximum switching frequency are 100 kHz and 150 kHz. The quality factor is obtained through Equation (22). Using the resonant frequency and Q, the L_r and C_r can be determined as 66.8 μH and 37.8 nF. The input inductance can be designed to obtain the ZVS condition. The maximum switching frequency and minimum duty ratio are the worst ZVS conditions, as shown in Figure 6. The input inductance is determined as 5 μH, which can guarantee the ZVS condition for the entire operating condition.

Table 2. Designed MLPE specifications.

Parameter	Value
Maximum Power	500 W
Input Voltage Range	15–60 V
Voltage Gain	1–3
Output Voltage	<80 V
Duty Control Range	0.5–0.75
Resonant Frequency (Fr)	100 kHz
Maximum Switching Frequency	150 kHz
Resonant Inductance	66.8 μH
Resonant Capacitance	37.8 nF
Input Inductance	5 μH

shows the designed MLPE specifications.

4. Experimental Verification

The validation of the designed power converter’s performance is aptly substantiated through the integration of meticulous experimental investigations. Figure 9 provides a comprehensive insight into the experimental setup orchestrated for the current-fed series resonant converter. This setup encompasses essential elements such as the photovoltaic (PV) simulator, the electric load unit, and the oscilloscope, all collectively constituting a robust platform for real-world testing and data collection.

Delving deeper into the empirical analysis, Figure 10 showcases the experimental outcomes during steady-state operations, delineated across varying operating frequencies. A noteworthy highlight of the proposed converter is its seamless realization of zero-voltage switching operations, a pivotal feat that attests to its advanced design and efficiency. Moreover, the dynamic behavior of the DC link voltage in the current-fed structure emerges as a significant performance enabler. This voltage augmentation effectively translates into an amplified effective input voltage, contributing to enhanced energy conversion efficiency.

Figure 11 presents an illuminating glimpse into the experimental results associated with different duty cycle scenarios. Evidently, the voltage gain exhibits a perceptible variation, mirroring the fluctuations in the DC link voltage. Notably, the converter adeptly maintains its zero-voltage switching prowess across all operating conditions, reinforcing its robustness and adaptability.

Furthermore, Figure 12 adds a layer of insight by showcasing the step load response of the meticulously designed converter. This response not only serves as a litmus test for the converter’s stability but also underscores its resilience in the face of dynamic load variations.

From Figure 13, the power conversion efficiency is dissected with respect to varying power ratings and input voltages, offering a holistic view of the converter’s prowess across different operational scenarios.

In summation, the culmination of empirical evidence underscores the prowess of the designed power converter. The multifaceted experimentation not only establishes its zero-voltage switching capabilities, stability under dynamic load conditions, and dynamic voltage gain performance but also offers insights into its efficiency under diverse power ratings and input voltages. This empirical validation thus encapsulates the converter’s versatility, reliability, and efficiency, positioning it as a promising contender in the realm of contemporary power electronics.

5. Discussion

5.1. Advantage of Proposed Converter

The current-fed series resonant converter presents a distinct set of advantages that set it apart from various conventional boost converters. These differentiators stem from its unique topology and operational characteristics, enabling it to excel in specific applications and scenarios.

Firstly, one of the prominent advantages of the current-fed series resonant converter lies in its inherent ability to achieve zero-voltage switching (ZVS) across a wide range of operating conditions. This characteristic minimizes switching losses, leading to improved overall efficiency. In contrast, many conventional boost converters face challenges related to switching losses, especially at high-frequency operations or under variable load conditions. The ZVS operation not only enhances efficiency but also contributes to reduced electromagnetic interference, making the current-fed series resonant converter well-suited for applications demanding low noise and high efficiency.

Secondly, the current-fed series resonant converter demonstrates exceptional voltage boosting capabilities, making it highly suitable for applications requiring high voltage gains. This is particularly advantageous in scenarios like renewable energy systems, where stepping up the voltage output of energy sources like solar panels is crucial for efficient power conversion. Unlike some conventional boost converters, the inherent characteristics of the series resonant topology allow for seamless voltage amplification, resulting in improved energy utilization and minimized losses.

Furthermore, the current-fed series resonant converter’s soft-switching operation enhances its reliability and extends the lifespan of components. The reduced stress on switching devices contributes to decreased wear and tear, ultimately leading to longer operating lifetimes. In contrast, certain conventional boost converters might experience higher stress on their components due to hard switching, potentially compromising their longevity and requiring more frequent maintenance.

In conclusion, the current-fed series resonant converter’s advantages in zero-voltage switching, high voltage gain capabilities, and improved component longevity position it as a compelling alternative to conventional boost converters. Its suitability for applications demanding efficiency, high voltage conversion ratios, and reduced wear on components makes it a valuable option in various industries, particularly in areas like renewable energy, where these advantages are highly sought after.

5.2. Operating Scenarios of Proposed Converter

Using the current-fed series resonant converter as a DC optimizer opens up several promising scenarios where its unique capabilities can be leveraged to enhance power management and conversion efficiency in various applications.

1.

Photovoltaic (PV) Systems Enhancement:

In photovoltaic (PV) systems, where solar energy is harvested and converted into electrical power, the current-fed series resonant converter can serve as a valuable DC optimizer. By interfacing between the PV array and the inverter, this converter can efficiently regulate the DC voltage output from the solar panels. Its inherent ability to achieve zero-voltage switching (ZVS) ensures minimal switching losses, leading to improved overall system efficiency. Additionally, its high voltage gain capabilities can enable effective voltage matching between the PV array and the inverter’s input requirements, optimizing power transfer. This not only enhances the energy harvesting efficiency of the PV system but also reduces the stress on components, prolonging their lifespans. The converter’s adaptability to varying insolation levels and load conditions further solidifies its role as an efficient DC optimizer for PV systems.

2.

Battery Charging and Energy Storage Systems:

Current-fed series resonant converters can play a pivotal role in battery charging and energy storage applications. In scenarios where energy from renewable sources, such as solar or wind, needs to be efficiently stored in batteries, these converters can optimize the charging process. By regulating the voltage and current levels fed to the battery, the converter ensures efficient energy transfer while minimizing losses through its ZVS operation. Moreover, its soft-switching characteristics reduce stress on both the converter and the battery, contributing to prolonged battery life. The converter’s high voltage gain capabilities facilitate the stepping up of low-voltage energy sources to match the battery voltage, ensuring optimal energy storage without excessive losses. This role as a DC optimizer contributes to the seamless integration of renewable energy sources into the grid and enhances the reliability of energy storage systems.

3.

Electric Vehicle Charging Infrastructure:

In the realm of electric vehicle (EV) charging, the current-fed series resonant converter can serve as a DC optimizer to enhance the efficiency and flexibility of EV charging stations. These converters can be employed to efficiently convert the grid AC voltage to the required high DC voltage for fast-charging EVs. Their ZVS operation minimizes switching losses, enabling high-efficiency power conversion. Additionally, high voltage gain capabilities can facilitate the transformation of the available grid voltage to the desired charging voltage, accommodating different EV models and battery capacities. As EV charging infrastructure continues to expand, the use of current-fed series resonant converters as DC optimizers can contribute to faster and more efficient charging while minimizing stress on the grid.

In essence, deploying the current-fed series resonant converter as a DC optimizer offers transformative possibilities across various applications. Its ZVS operation, high voltage gain capabilities, and soft-switching characteristics make it a valuable tool for enhancing power management, improving efficiency, and extending the lifespan of components in diverse scenarios, ranging from renewable energy systems to battery storage and electric vehicle charging infrastructure.

6. Conclusions

The paper introduces a novel approach by suggesting the utilization of the current-fed series resonant converter in the realm of module-level power electronics applications. The primary objective is to leverage the unique attributes of this converter topology for enhanced performance within such contexts. The operational intricacies of the proposed converter are meticulously examined, with a particular focus on how variations in duty ratio and switching frequency influence its behavior.

Central to this study is the elucidation of the design methodology formulated for the suggested converter. This methodology is explicitly developed to harness two critical advantages: achieving wide voltage gains and enabling zero-voltage switching capabilities. By accomplishing these twin objectives, the proposed converter becomes a potential solution for scenarios where voltage augmentation and efficient power management are paramount.

The innovation introduced in this work extends to the control domain as well. A hybrid control algorithm is devised, effectively amalgamating asymmetric pulse-width modulation (PWM) with frequency modulation (FM). This hybrid strategy presents a distinctive advantage—the ability to operate with minimal root mean square (RMS) current levels. Such an outcome is invaluable, as it not only enhances the overall system efficiency but also contributes to a reduction in power losses and heat generation.

The culmination of this research effort rests in the empirical validation of the proposed converter’s performance. Through a series of meticulously designed experiments, the practical efficacy of the converter is established. These results offer concrete evidence of the viability of the suggested approach, demonstrating its potential to meet the demands of module-level power electronics applications. This multifaceted exploration thus contributes to the expanding landscape of power electronics by introducing a novel converter topology, an innovative design methodology, and a sophisticated hybrid control algorithm—all substantiated by empirical validation—that collectively foster a new avenue of possibilities for efficient and high-performance power electronics systems.