Resonant Converter with Soft Switching and Wide Voltage Operation

Abstract

A new DC/DC resonant converter with wide output voltage range operation is presented and studied to have the benefits of low switching losses on active devices and low voltage stresses on power diodes. To overcome serious reverse recovery losses of power diodes on a conventional full-bridge pulse-width modulation converter, the resonant converter is adopted to reduce the switching loss and increase the circuit efficiency. To extend the output voltage range in conventional half-bridge or full-bridge resonant converters, the secondary sides of two diode rectifiers are connected in series to have wide output voltage operation. The proposed converter can be either operated at one-resonant-converter mode for low voltage range or two-resonant-converter mode for high voltage range. Thus, the voltage rating of power diodes is decreased. Experiments with the design example are given to show the circuit performance and validate the theoretical discussion and analysis.

1. Introduction

Power saving and power loss issues have become more and more important with respect to reducing the effect of global warming. Low power loss means less power demand is needed from the utility company. Modern power converters have demanded the development of high power density and low power loss for modern consumer and industry products. Soft switching techniques or wide-band gap power devices, such as SiC or GaN power switches, have been widely developed to lessen the switching losses. SiC and GaN have low switching loss characteristics and power converters and can operate at high frequency and high power density, however, the wide-band gap power switches are much more expensive than the conventional MOSFETs. For zero voltage switching techniques, a quasi-resonant (QR) approach has been applied in flyback [1,2] to lessen the switching loss. The circuit efficiency of QR flyback with synchronous rectifier can be higher than 90%. Active clamp techniques [3] have been applied in buck, boost, flyack, and forward converters to lessen the switching losses for low-power applications. More components are needed in the circuit, and these additional components introduce more power loss. For medium-power products, resonant converters [4,5,6,7] and full-bridge pulse width modulation converters [8,9,10,11] were developed to reduce switching losses. Full-bridge converters lose soft switching operation at low output power and the circuit efficiency will decrease at low load conditions. The resonant converters have soft switching characteristics on power semiconductors over the full load range so that the efficiency of the resonant converter at low load is better than phase-shift full-bridge converters. The limited voltage variation range is the main drawback of the resonant converters. If the voltage operation range is increased, then the low inductor ratio and low quality factor should be used. However, the circulating current and conduction loss on the primary side will increase and the circuit efficiency decreases.

Wide output voltage power converters are demanded for battery chargers for hybrid electric vehicles or outdoor LED lighting systems. In electric vehicle and hybrid electric vehicle battery charging systems, the single-phase power factor corrector and isolated DC converter are used in type I and II charging systems, and the three-phase power factor corrector and full-bridge DC converter are adopted in type III charging system. The output voltage of the high-voltage battery stack in PHEV and EV varies widely with battery capacity, such as 200–420 V. In outdoor LED lighting systems, the DC output voltage is variable for different series and parallel combinations of LED strings’ applications, such as 120–360 V. The DC-DC converters with wide output voltage variation have been developed for battery chargers. To achieve wide output voltage capability, the full-bridge DC converters with phase-shift pulse width modulation were presented and discussed in [12,13] to have high circuit efficiency and low switching losses. The voltage range is limited to less than 2:1 (v_o,max = 2v_o,min) or 3:1 (v_o,max = 3v_o,_min) due to low converter efficiency in low duty cycle or low output voltage cases. To realize a wide output voltage range, the basic circuit topologies are two-stage DC-DC converters (boost or buck converter + full-bridge converter) to implement 4:1 (v_o,max = 4v_o,_min) voltage variation. However, the circuit efficiency is reduced in these circuit topologies. The hybrid resonant converter with half-bridge or full-bridge operation has been studied in [14,15,16] to increase the voltage operation range, such as a 4:1 voltage range, v_in,max = 4v_in,min. The main problems are that a transient operation interval will be introduced between the half-bridge resonant circuit and the full-bridge resonant circuit operation, and one power switch is always conducting under the half-bridge resonant circuit to decrease circuit efficiency. The voltage rating of power diodes on conventional full-bridge or center-tapped diode rectifiers equals the output voltage or two times the output voltage, respectively. The rectifier diodes have serious voltage stress problems for high-voltage output conditions.

A hybrid resonant circuit with an input-parallel output-series structure is presented and studied to accomplish the improvements of the low-voltage rating on power diodes, wide voltage operation range, and wide soft switching range. By using the frequency modulation approach, the proposed converter is controlled under inductive impedance. The soft switching operation of power devices is achieved. The switching losses on power semiconductors are reduced. To overcome the high-voltage rating problem on rectifier diodes, two full-bridge diode rectifiers are employed on the proposed circuit. For high-voltage output, two full-bridge resonant converters are all operated and the secondary-side rectified voltages are series-connected. The voltage rating of each diode is V_o,H/2 instead of V_o,H in conventional DC-DC converters. On the other hand, only one full-bridge resonant converter is operated under the low-voltage output case. Only one diode rectifier is connected to the output load and the other diode rectifier is bypassed. The voltage rating of the power diodes is V_o,L. The drawback of the high-voltage rating on rectifier diodes is improved in the presented circuit. The theoretical analysis, the principle of operation, and the design procedure are provided in this paper. Finally, a 2 kW laboratory circuit was built and tested. Experimental results are used to confirm the theoretical analysis and demonstrate the helpfulness of the presented hybrid resonant converter for wide output voltage capability.

2. Circuit Structure and Operating Principle

The studied converter with wide output voltage operation is illustrated in Figure 1a. Two resonant converters with primary-parallel secondary-series connection are adopted to accomplish the wide voltage operation. V_in is the input voltage. V_o is the output voltage. In each converter, the primary side is a full-bridge resonant structure and the secondary side is the full-bridge diode rectifier. Four power switches are used in the full-bridge resonant structure and four rectifier diodes are employed in the full-bridge rectifier structure. C_r1 and L_r1 are naturally resonant in circuit 1 and C_r2 and L_r2 are resonant in circuit 2. For high-voltage output, two converters are operated and V_o is the summation of the rectified voltages of two full-bridge diode rectifiers shown in Figure 1a. For low-voltage output (Figure 1b), only converter 1 (S₁~S₄) is controlled and converter 2 (S₅~S₈) is off. The load voltage V_o equals the rectified voltage of the full-bridge rectifier by D₁~D₄ and the diodes D₅~D₈ are forward-biased to bypass the secondary side current of transformer T₁. The wide output voltage is achieved in the studied circuit. The frequency control method is adopted to control active devices and adjust the load voltage. The power semiconductors are operated with soft switching characteristics due to the resonant tank having an inductive impedance characteristic.

2.1. High-Voltage Output

The resonant converter is controlled by the frequency modulation. If f_s (the switching period) > or < f_r (series resonant frequency), there are four or six operating modes in every switching period. For high output voltage range, two converters are controlled to increase the load voltage due to the series connection of two full-bridge diode rectifiers at the load side, as shown in Figure 1a. Figure 2a (Figure 2b) illustrates the voltage and current waveforms of the studied converter at high-voltage output under f_s > f_r (f_s < f_r) condition. Figure 2c–h demonstrate six equivalent mode circuits at high-voltage output operation and f_s < f_r (Figure 2b). If f_s > f_r (Figure 2a), only modes 1, 3, 4, and 6 are operated in a switching cycle.

Mode 1 [t₀, t₁]: At time t < t₀, S₁~S₈ are off and i_Lr1 and i_Lr2 are negative. C_S1, C_S4, C_S5, and C_S8 are discharged and C_S2, C_S3, C_S6, and C_S7 are charged. At t₀, v_CS1 = v_CS4 = v_CS5 = v_CS8 = 0. Since i_Lr1(t₀) and i_Lr2(t₀) are all negative value, D_S1, D_S4, D_S5, and D_S8 are conducting. S₁, S₄, S₅, and S₈ turn on after t₀ under zero voltage. Due to D₁, D₄, D₅, and D₈ are conducting, v_Lm1 = v_Lm2 = nV_o/2 where n is the turns ratio of T₁ and T₂. C_r1 and L_r1 are naturally resonant in converter 1 and C_r2 and L_r2 are resonant in converter 2 with v_ab = v_cd = V_in and v_Lm1 = v_Lm2 = nV_o/2. The series resonant frequency in mode 1 is $f_r=1/2\pi \sqrt{L_{r1}{C}_{r1}}$. If f_s < f_r, then i_D1, i_D4, i_D5, and i_D8 will be decreased to zero current before S₁, S₄, S₅, and S₈ turn off.

Mode 2 [t₁, t₂]: Owing to f_s < f_r, i_Lm1(t₁) = i_Lr1(t₁) and i_Lm2(t₁) = i_Lr2(t₁) so that i_D1 = i_D4 = i_D5 = i_D8 = 0. No power is delivered to the load resistor. (L_m1, L_r1, and C_r1) and (L_m2, L_r2, and C_r2) are resonant with input voltage v_ab = v_cd = V_in in converters 1 and 2, respectively.

Mode 3 [t₂, t₃]:S₁, S₄, S₅ and S₈ turn off at t₂. C_S2, C_S3, C_S6, and C_S7 are discharged in mode 2 due to i_Lr1(t₂) > 0 and i_Lr2(t₂) > 0. Since the secondary side currents of T₁ and T₂ are negative, the diodes D₂, D₃, D₆, and D₇ conduct. If the energy stored on (L_r1 and L_m1) and (L_r2 and L_m2) at time t₂ is large enough, then the voltages of C_S2, C_S3, C_S6 and C_S7 are decreased to zero at t = t₃. The resonant currents are calculated in Equation (1):

$$i_{Lr1}(t_2)=i_{Lr2}(t_2)\ge i_{Lm1}(t_2)=i_{Lm2}(t_2)=n{V}_o/(4L_m{f}_s)$$

(1)

where L_m = L_m1 = L_m2. The soft switching condition of S₂, S₃, S₆, and S₇ are given in Equation (2):

$$i_{Lm1}(t_2)=i_{Lm2}(t_2)\ge V_{in}\sqrt{4C_e/(L_m+L_r)}$$

(2)

where C_e = C_S1 = … = C_S8 and L_r = L_r1 = L_r2. If the dead time t_d between each switch is given, then the maximum magnetizing inductances are calculated in Equation (3):

$$L_{m1}=L_{m2}\le \frac{n{V}_o{t}_d}{8V_{in}{f}_s{C}_e}$$

(3)

Mode 4 [t₃, t₄]: The voltages of C_S2, C_S3, C_S6, and C_S7 are reduced to zero at t₃. D_S2, D_S3, D_S6, and D_S7 are forward biased owing to i_Lr1(t₃) and i_Lr2(t₃) are positive value. S₂, S₃, S₆, and S₇ turn on after t₃ with zero voltage condition. In mode 4, D₂, D₃, D₆, and D₇ are conducting so that v_Lm1 = v_Lm2 = −nV_o/2. (C_r1 and L_r1) and (C_r2 and L_r2) are naturally resonant in converters 1 and 2, respectively, with input voltage v_ab = v_cd = −V_in and load voltage v_Lm1 = v_Lm2 = −nV_o/2. If f_s < f_r, then i_D2, i_D3, i_D6, and i_D7 will be decreased to zero current before S₂, S₃, S₆, and S₇ are turned off.

Mode 5 [t₄, t₅]: Since f_s < f_r, i_Lm1 = i_Lr1 and i_Lm2 = i_Lr2 at time t₄ and i_D2 = i_D3 = i_D6 = i_D7 = 0. No power is transferred to the load resistor. (L_m1, L_r1 and C_r1) and (L_m2, L_r2 and C_r2) are naturally resonant in converters 1 and 2, respectively, with input voltage v_ab = v_cd = −V_in.

Mode 6 [t₅, T_s + t₀]: At t₅, S₂, S₃, S₆, and S₇ turn off. Due to i_Lr1(t₅) and i_Lr2(t₅) are negative value, C_S1, C_S4, C_S5, and C_S8 are discharged. At time T_s + t₀, the voltages of C_S1, C_S4, C_S5, and C_S8 are decreased to zero voltage and the studied circuit goes to the next switching cycle operation.

2.2. Low-Voltage Output

For low-voltage output, only S₁~S₄ are controlled and S₅~S₈ are off. The secondary side voltage of T₁ is connected to the output load and D₅~D₈ are forward biased, as shown in Figure 1b. Figure 3a,b give the current and voltage waveforms of the studied circuit for low-voltage output under f_s > f_r and f_s < f_r conditions, respectively. If f_s > f_r (Figure 3a), only modes 1, 3, 4, and 6 are operated in a switching cycle. If f_s < f_r (Figure 3b), there are six equivalent mode circuits (Figure 3c–h) at low-voltage output operation. The operation principle at low-voltage output at f_s < f_r is presented below.

Mode 1 [t₀, t₁]:S₁~S₄ are off and i_Lr1 < 0 before time t₀. C_S1 and C_S4 discharge by the current i_Lr1. At time t₀, C_S1 and C_S4 discharge to zero voltage and D_S1 and D_S4 conduct. The drain-to-source voltages v_S1,d = v_S4,d = 0 and S₁ and S₄ turn on after t₀ under zero voltage condition. In mode 1, the load current flows through D₁, C_o, R_o, D₅~D₈ and D₄. The magnetizing inductor voltage v_Lm1 = nV_o. C_r1 and L_r1 are naturally resonant with v_ab = V_in and v_Lm1 = nV_o. If f_s < f_r, then i_D1, i_D4, and i_D5~i_D8 will be reduced to zero before S₁ and S₄ turn off.

Mode 2 [t₁, t₂]: Due to f_s < f_r, i_Lm1 = i_Lr1 at time t₁ so that D₁~D₈ are off on the output side. L_r1, C_r1, and L_m1 are naturally resonant with V_ab = V_in. If f_s ≈ f_r, the peak-to-peak magnetizing current Δi_Lm1 during one-half of switching period approximates:

$$\mathsf{\Delta }{i}_{Lm1}(t)\approx \frac{n{V}_o}{2L_m{f}_s}$$

(4)

Mode 3 [t₂, t₃]: At time t₂, S₁ and S₄ turn off. Owing to i_Lr1 > 0, the voltages of C_S2 and C_S3 are decreased. If the current i_Lr1(t₂) ≈ Δi_Lm1/2 is large enough, then the voltages of C_S2 and C_S3 will be decreased to zero at t₃.

Mode 4 [t₃, t₄]: At t₃, v_CS2 = v_CS3 = 0 and D_S2 and D_S3 conduct. S₂ and S₃ turn on after t₃ under zero voltage. The load current flows through D₃, C_o, R_o, D₅~D₈, and D₂ so that v_Lm1 = −nV_o. In mode 4, C_r1 and L_r1 are naturally resonant with v_ab = −V_in and v_Lm1 = −nV_o in mode 4. If f_s < f_r, then i_D2, i_D3, and i_D5~i_D8 decrease to zero current before S₂ and S₃ turn off.

Mode 5 [t₄, t₅]: At t₄, i_Lm1 = i_Lr1 so that D₁~D₈ are reverse biased. L_r1, C_r1, and L_m1 are resonant with v_ab = −V_in. Mode 5 ends at t₅ when S₂ and S₃ turn off.

Mode 6 [t₅, T_s + t₀]: At t₅, S₂ and S₃ turn off. Owing to i_Lr1 < 0, C_S1 and C_S4 are discharged by i_Lr1. At time T_s + t₀, v_CS1 = v_CS4 = 0.

3. Circuit Characteristics and Design Guidelines

The load voltage control is based on the frequency modulation against line and load regulations. To ignore the harmonic components, the fundamental harmonic frequency approach is used to estimate the AC voltage gain of the studied circuit. The wide voltage operation is accomplished by selecting the series connections of two full-bridge diode rectifiers for high-voltage output (Figure 1a) or only one diode rectifier for low-voltage operation (Figure 1b). Since S₁~S₈ have the T_s/2 turn-on time, the AC voltages v_ab and v_cd are square voltage waveforms with voltage values ±V_in. The fundamental root-mean-square (rms) voltages V_ab,rms and V_cd,rms are calculated in Equation (5):

$$V_{ab,rms}=V_{cd,rms}=2\sqrt{2}{V}_{in}/\mathsf{\pi }$$

(5)

For high-voltage output, two full-bridge diode rectifiers are series-connected and connect to the output load. The magnetizing voltages v_Lm1 = v_Lm2 = ±nV_o/2. For low-voltage output, only one full-bridge connects to the output load and the magnetizing voltage v_Lm1 = ±nV_o and v_Lm2 = 0. The fundamental rms magnetizing voltages V_Lm1,rms and V_Lm2,rms are calculated as:

$$V_{Lm1,rms}=\{\begin{array}{c}\sqrt{2}n{V}_{o,H}/\mathsf{\pi },\mathrm{for}\mathrm{high}\mathrm{voltage}\mathrm{output}\\ 2\sqrt{2}n{V}_{o,L}/\mathsf{\pi },\mathrm{for}\mathrm{low}\mathrm{voltage}\mathrm{output}\end{array}$$

(6)

$$V_{Lm2,rms}=\{\begin{array}{c}\sqrt{2}n{V}_{o,H}/\mathsf{\pi },\mathrm{for}\mathrm{high}\mathrm{voltage}\mathrm{output}\\ 0,\mathrm{for}\mathrm{low}\mathrm{voltage}\mathrm{output}\end{array}$$

(7)

where V_o,L and V_o,H denote the output voltage V_o at the low and high voltage ranges, respectively. From the given DC load resistor R_o, the equivalent primary side resistors R_ac1 and R_ac2 of T₁ and T₂ are calculated in Equations (8) and (9):

$$R_{ac1}=\{\begin{array}{c}\frac{4n^2{R}_o}{{\mathsf{\pi }}^2},\mathrm{for}\mathrm{high}\mathrm{voltage}\mathrm{output}\\ \frac{8n^2{R}_o}{{\mathsf{\pi }}^2},\mathrm{for}\mathrm{low}\mathrm{voltage}\mathrm{output}\end{array}$$

(8)

$$R_{ac2}=\frac{4n^2{R}_o}{{\mathsf{\pi }}^2},\mathrm{for}\mathrm{high}\mathrm{voltage}\mathrm{output}$$

(9)

Based on the resonant circuit, the transfer function of the studied circuit is calculated in Equation (10):

$$|G|=\frac{V_{Lm1,rms}}{V_{ab,rms}}=\frac{1}{\sqrt{{[1+\frac{1}{l_n}\frac{f_n^2-1}{f_n^2}]}^2+x^2{(\frac{f_n^2-1}{f_n})}^2}}=\{\begin{array}{l}\frac{n{V}_{o,H}}{2V_{in}},\mathrm{for}\mathrm{high}\mathrm{voltage}\mathrm{output}\\ \frac{n{V}_{o,L}}{V_{in}},\mathrm{for}\mathrm{low}\mathrm{voltage}\mathrm{output}\end{array}$$

(10)

where f_n = f_s/f_r is the frequency ratio, l_n = L_m1/L_r1 is the inductor ratio, and $x=\sqrt{L_{r1}/C_{r1}}/R_{ac1}$ is the quality factor. The output voltage at high and low output voltage range can be re-written in Equations (11) and (12):

$$V_{o,H}=\frac{2V_{in}}{n\sqrt{{[1+\frac{1}{l_n}\frac{f_n^2-1}{f_n^2}]}^2+x^2{(\frac{f_n^2-1}{f_n})}^2}}$$

(11)

$$V_{o,L}=\frac{V_{in}}{n\sqrt{{[1+\frac{1}{l_n}\frac{f_n^2-1}{f_n^2}]}^2+x^2{(\frac{f_n^2-1}{f_n})}^2}}$$

(12)

Based on the output voltage range, S₁~S₈ are regulated with variable frequency control to accomplish the wide voltage range output capability. Due to the resonant circuit always being controlled at the inductive impedance load, S₁~S₈ will be operated under the soft switching condition.

To confirm the theoretical analysis, the design guideline of a laboratory circuit is provided. The electrical specifications of the prototype circuit are V_in = 380 V and V_o = 110 V~440 V (4:1 voltage ratio). The rated output power is 1000 W (200 W) for low (high)-voltage output and f_r = 100 kHz. To accomplish wide voltage output operation (4:1), two full-bridge resonant converters with input-parallel output-series connection are employed under the high output voltage range V_o,H = 220 V~440 V. However, only converter 1 is operated for low-voltage output V_o,L = 110 V~220 V. The voltage gain of the studied circuit with wide output voltage operation V_o = 110 V~440 V under the V_in = 380 V condition is provided in Figure 4. The transient voltage (V_o,tran) between two resonant converters’ operation and one resonant converter operation is 220 V with ±5 V voltage tolerance using a Schmitt trigger comparator, as shown in Figure 5. Since the voltage gain of the studied circuit in the low-voltage output range is two times the voltage gain in the high-voltage output range in Equation (10), two resonant converters’ operation for high-voltage output (Figure 1a) and one resonant converter operation for low-voltage output (Figure 1b) have the same design procedure due to V_o,H = 2V_o,L in Equations (11) and (12). Only one resonant converter is designed for the low-voltage output case to simplify the design guideline. For the low-voltage output case, the minimum voltage gain G_min is equal to unity at 110 V output voltage. The turns ratio n of T₁ can be obtained from Equation (13):

$$n=\frac{G_{\min }{V}_{in}}{V_{o,L,\min }}=\frac{1\times 380\mathrm{V}}{110\mathrm{V}}\approx 3.455$$

(13)

TDK (Tokyo Denki Kagaku, Tokyo, Japan) EE-55 magnetic cores with flux density ΔB = 0.4 T and A_e = 3.54 cm² are adopted to implement transformers T₁ and T₂. The minimum switching frequency in the studied circuit is assumed at 60 kHz under V_o,tran output. The minimum primary winding turns are obtained in Equation (14):

$$N_{p1,\min }\ge \frac{n{V}_{o,tran}}{2f_{s,\min }\mathsf{\Delta }B{A}_e}=\frac{3.455\times 220\mathrm{V}}{2\times 60000\mathrm{H}z\times 0.4\mathrm{T}\times 3.54\times {10}^{-4}{\mathrm{m}}^2}\approx 45$$

(14)

In the prototype circuit, the actual transformer winding turns of T₁ and T₂ are N_p1 = N_p2 = 45 and N_s1 = N_s2 = 13. The equivalent primary side resistance R_ac1 under P_o,max = 1000 W and V_o,L,min = 110 V is calculated in Equation (15):

$$R_{ac1}=\frac{8n^2}{{\mathsf{\pi }}^2}{R}_{o,rated}=\frac{8\times {(45/13)}^2}{{\mathsf{\pi }}^2}\times \frac{{(110\mathrm{V})}^2}{1000\mathrm{W}}\approx 117.5\mathrm{\Omega }$$

(15)

In the prototype circuit, the selected parameters l_n = 5 and x = 0.2. The resonant components L_r1, L_r2, L_m1, L_m2, C_r1, and C_r2 are calculated as:

$$L_{r1}=L_{r2}=\frac{x{R}_{ac1}}{2\mathsf{\pi }{f}_r}=\frac{0.2\times 117.5\mathrm{\Omega }}{2\mathsf{\pi }\times 100\times {10}^3\mathrm{Hz}}\approx 37.4\mathsf{\mu }\mathrm{H}$$

(16)

$$L_{m1}=L_{m2}=l_n{L}_{r1}=37.4\times 5\mathsf{\mu }\mathrm{H}=187\mathsf{\mu }\mathrm{H}$$

(17)

$$C_{r1}=C_{r2}=\frac{1}{4{\mathsf{\pi }}^2{L}_{r1}{f}_r^2}=\frac{1}{4{\mathsf{\pi }}^2\times 37.4\times {10}^{-6}\mathrm{F}\times {(100\times {10}^3\mathrm{Hz})}^2}\approx 68\mathrm{nF}$$

(18)

The voltage ratings of D₁~D₈ are equal to V_o,max/2 = 220 V. The voltage ratings of S₁~S₈ are V_in = 380 V. Power switches SiHG20N50C (500 V/ 11A) are employed for S₁~S₈ and SBR20A300CTFP (300 V/20 A) are used for D₁~D₈. The selected output capacitance C_o = 720 μF/450 V. The frequency modulation controller UCC25600 is adopted to generate the gating signals of S₁~S₈.

4. Experimental Verification

A laboratory prototype (Figure 6) was constructed and tested. The circuit parameters are calculated from the previous section. The test waveforms are demonstrated and shown to confirm the theoretical discussion and effectiveness of the studied circuit with a wide output voltage range. Figure 7 and Figure 8 provide the experimental results under low-voltage range with only one resonant converter operation. Figure 7 demonstrates the experimental waveforms for V_o = 110 V and P_o = 1 kW. The experimental waveforms of v_S1,g~v_S4,g are given in Figure 7a. The switching frequency of S₁~S₄ is about 100 kHz. The leg voltage v_ab and the resonant component waveforms v_Cr1 and i_Lr1 are provided in Figure 7b. v_Cr1 and i_Lr1 are almost sinusoidal waveforms due to f_s ≈ f_r. The inductor current i_Lr1 is lagging to the measured voltage v_ab. The input impedance of the resonant tank by C_r1, L_r1, L_m1, and R_ac1 is operated at the inductive load. Figure 7c,d provide the measured diode currents. It is obvious that D₁~D₄ turn off under zero current. There is a slight current imbalance on D₅ and D₇ due to the unequal PCB layout length of D₅ and D₇. Figure 7e provides the test results of V_in, V_o, and I_o. The output voltage V_o is controlled at 110 V output and the load current is 9.1 A. Figure 7f shows the experimental waveforms v_S1,g, v_S1,d, and i_S1. It is observed that the drain voltage v_S1,d is decreased to zero before S₁ turns on. The zero-voltage turn-on switching of switch S₁ is achieved in Figure 7f. S₂~S₄ have the same turn-on characteristic as S₁. It can be expected that the zero-voltage turn-on switching of S₂~S₄ is also implemented.

Figure 8 provides the measured results for V_o = 215 V and P_o = 1 kW. The experimental waveforms v_S1,g~ v_S4,g are given in Figure 8a and the switching frequency is about 57 kHz. Comparing the test results in Figure 7a and Figure 8a, the switching frequency under V_o = 215 V is lower than the switching frequency under the V_o = 110 V case. Figure 8b illustrates the measured signals of v_ab, v_Cr1, and i_Lr1. i_Lr1 is a quasi-sinusoidal waveform due to f_s < f_r under V_o = 215 V. The inductor current i_Lr1 is lagging the measured voltage v_ab. The input impedance of the resonant tank by C_r1, L_r1, L_m1, and R_ac1 is operated at the inductive load. The measured diode currents on the output side are provided in Figure 8c,d. It can be observed that D₁~D₄ turns off under zero current. The measured waveforms of V_in, V_o, and I_o are provided in Figure 8e. Figure 8f shows the experimental waveforms v_S1,g, v_S1,d, and i_S1. It is clear that v_S1,d is decreased to zero before S₁ turns on. The zero-voltage turn-on switching of S₁ is achieved in Figure 8f for V_o = 215 V condition.

Figure 9 and Figure 10 give the measured waveforms for the high-voltage output range. Two full-bridge resonant converters with primary-parallel secondary-series connection are controlled in the presented converter to provide high-voltage output. Figure 9 provides the experimental results at V_o = 225 V and P_o = 2000 W. Figure 10 gives the experimental waveforms at 440 V output voltage and 2 kW output power. Figure 9a,b provide the waveforms of S₁~S₈ and the switching frequency is about 106 kHz. Figure 9c provides the capacitor voltages and inductor currents of two full-bridge resonant converters. The voltage and current waveforms on two full-bridge circuits are balanced well. The measured diode currents on the secondary sides are demonstrated in Figure 9d,e and the diode currents i_D1~i_D8 of two full-bridge diode rectifiers are also balanced well. Figure 9f provides the measured results of V_in, V_o, and I_o. The measured voltage V_o = 225 V and I_o = 8.8 A. The experimental waveforms v_S1,g, v_S1,d, and i_S1 are given in Figure 9g. Before S₁ is turned on, v_S1,d is decreased to zero. The soft switching turn-on of S₁ is realized for V_o = 225 V case. The switch signals S₁~S₄ and S₅~S₈ are demonstrated in Figure 10a,b under V_o = 440 V and P_o = 2 kW conditions. The switching frequency is about 59 kHz. The experimental results of v_Cr1, v_Cr2, i_Lr1, and i_Lr2 are shown in Figure 10c. It is clear that capacitor voltages v_Cr1 and v_Cr2 and inductor currents i_Lr1 and i_Lr2, i_Lr1, and i_Lr are well balanced. The diode currents i_D1~i_D4 and i_D5~i_D8 are provided in Figure 10d,e, respectively. It can be observed that i_D1~i_D4 and i_D5~i_D8 are well balanced. The measured waveforms of V_in, V_o, and I_o are provided in Figure 10f under V_o = 440 V and I_o = 4.6 A. The experimental waveforms v_S1,g, v_S1,d, and i_S1 are given in Figure 10g. Before S₁ is turned on, the drain voltage v_S1,d has been decreased to zero. The soft switching turn-on of S₁ is achieved for the V_o = 400 V case.

Figure 11 shows the measured efficiencies of the proposed converter under different output voltages and rated power. The test efficiencies of the presented circuit are 94.4% at V_o = 110 V under P_o = 1 kW (low output voltage range), 90.1% at V_o = 215 V under P_o = 1 kW (low output voltage range), 96.7% at V_o = 225 V under P_o = 2 kW (high output voltage range) and 90.6% at V_o = 440 V under P_o = 1 kW (high output voltage range). From the test results, it can be observed that the circuit efficiency in the low-voltage case V_o = 110 V (V_o = 225 V) is better than the high-voltage case V_o = 215 V (V_o = 440 V) for low output voltage range (high output voltage range). The switching period at V_o = 215 V is greater than the switching period at V_o = 110 V. Due to the magnetizing current loss, which depended on the output voltage and the switching period, the estimated power loss from the magnetizing current at V_o = 215 V output is greater than the V_o = 110 V case. Therefore, the measured efficiency of the studied converter at V_o = 110 V is better than the V_o = 215 V output. Similarly, the measured circuit efficiency at V_o = 225 V is better than the V_o = 440 V output. Comparing the circuits shown in Figure 1, it can be observed that one full-bridge resonant converter and eight rectifier diodes are operated for the low-voltage output case, and two full-bridge resonant converters and eight rectifier diodes are operated for the high-voltage output case. It is clear that there are more conduction losses on diodes D₅~D₈ for the low-voltage output. The power losses on these diodes are equal to 2V_fI_o, where V_f is the voltage drop on D₅~D₈. The percentage between the power loss on diodes D₅~D₈ and the rated power is about 2V_f/V_o. Since V_o >> V_f, it can be expected that this extra power loss on diodes D₅~D₈ will not result in serious problems. Due to the V_f = 0.9 V for diodes SBR20A300CTFP, the calculated power loss percentage on diodes D₅~D₈ is about 1.6% in the 110 V output case and 0.8% in the 215 V output case.

5. Conclusions

A new resonant circuit with a series-parallel structure is presented in this paper to achieve the benefits of low switching loss, wide output voltage capability, and low-voltage rating on rectifier diodes. In the proposed converter, two diode rectifiers with series-connection are employed on the secondary side to reduce the voltage rating on the rectifier diodes for high output voltage range operation. The voltage rating of rectifier diodes is V_o,max/2 instead of the V_o,max voltage rating in conventional resonant converters. Since the proposed resonant tank is operated under the inductive load condition, the soft switching operation can be implemented in the studied converter. Finally, a 2 kW laboratory circuit was constructed and experiments are demonstrated to confirm the usefulness of the presented hybrid resonant converter for wide voltage output capability.