1. Introduction
In terms of reducing the loss of transmission, building integrated photovoltaic (BIPV) systems reduce costs and increase efficiency as compared to a centralized PV plant [
1], however they require high reliability, low cost, and high voltage gain. Traditional inverters are buck-type converters which can no longer satisfy BIPV requirements. Additionally, they are vulnerable to electromagnetic interference because their bridge, which contains series switches, can easily short-circuit with strong current of shoot-through. In order to increase the voltage gain, a traditional solution is to cascade a voltage pumping structure with only a few passive elements [
2,
3]. However, it results in low efficiency. Owing to their high efficiency, single-state inverters have been widely used which usually combine a transformer or passive elements into traditional inverters to achieve a higher voltage gain [
2,
4,
5]. However, adding a transformer results in a bulky volume and low efficiency. Moreover, this solution has a narrow window of voltage adjustment since the transformer only has a fixed voltage ratio. Furthermore, they work under the threat of a shoot-through which leads to the need for controlling strategies. To provide an effective method to solve the shoot-through problem and increase output voltage systematically, the authors in [
6] proposed an
network named a Z-source inverter (ZSI), which performs a boost function and prevents components being destroyed when working during a shoot-through. However, applying a Z-source into a half-bridge inverter leads to the problem of an imbalance at the midpoint voltage of the input capacitors. Targeting this issue, authors in [
7] proposed a novel Z-source half-bridge converter which also realizes buck-boost function. However, with a fixed structure, ZSI or quasi-Z-source inverter (qZSI) cannot realize a high enough voltage gain, which is a requirement in BIPV. Magnetic coupling is an effective method to improve the performance and efficiency. Many magnetically coupled impedance networks have been explored, such as T-source [
8,
9], Trans-Z-source [
10],
-Z-source [
11], flipped-
-Z-source [
12], and A-source [
13]. Their different winding placements lead to different advantages and applications. Combining their advantages, a Y-source impedance network (also known as a Y-impedance-network) is proposed which has higher voltage gain and a smaller size with less passive components [
14]. However, there is a severe problem: the existence of pulsed input currents. Addressing this issue, quasi-Y-source inverters with an additional inductor are proposed in [
15,
16,
17], and they also have additional voltage gain. However, they can only vary voltage gain by adjusting the turns ratio and duty cycle which cannot meet special industrial requirements. Under the premise of solving the pulsed input currents problem, an improved quasi-Y-source converter was proposed with higher voltage gain [
18]. However, the additional switch leads to difficultly regarding control.
In order to increase voltage gain and provide more degrees of adjustment to vary the voltage, the additional inductor in [
15] is replaced by a general step-up cell. Combining the advantages of half-bridge converters and a Y-impedance network, we propose a novel family of high step-up Y-impedance-network half-bridge converters, coupled with a general step-up cell for additional voltage adjustment to solve the above problems.
The rest of this paper is organized as follows. Detailed descriptions and analyses of the proposed converters are given in
Section 2. In
Section 3, the proposed converter and conventional Z-source half-bridge converter are compared to demonstrate the unique features of the proposed solution. In
Section 4 and
Section 5, the simulation and experimental studies are presented. Finally, a conclusion is drawn in
Section 6.
2. Operating Principle and Analysis
The structure of the proposed topologies are depicted in
Figure 1 and the step-up cells are illustrated in
Figure 2, in which the step-up cell is a general cell, which can be a single inductor, a switched inductor, a quasi Z-source network, or a switched-coupled inductor [
19]. The cell endows the proposed converters with an additional voltage adjustment function.
Since the operating principles of the proposed converters are the same as the conventional ones with the condition of
, but not
, the latter is analyzed in this study. Furthermore, different step-up cells perform the same boost function, so a switched inductor Y-impedance-network half-bridge converter is analyzed as a typical example. In detail, a Y-impedance-network and a switched inductor are integrated into a conventional half-bridge inverter to form a new topology with three operating modes, as shown in
Figure 3.
Denote as the beginning of one period, as the turning point from Mode 1 to Mode 2, as the time from Mode 2 to Mode 3, and as the end of the period. The operating process in one period is analyzed in detail in the following, and the output voltage is deduced in each mode.
(1) Mode 1: .
As shown in
Figure 3a, the proposed converter works in a shoot-through state. Diodes
and
are reverse biased simultaneously, capacitors
discharges energy to the coupled inductor, and capacitors
and the input voltage source discharge energy to the switched inductor cell (
,
) and the load
R. Voltages across inductors
,
, coupled inductor
, and output voltage can be deduced as
and
where
and
are turns ratios of the three-winding coupled inductor.
(2) Mode 2: .
As shown in
Figure 3b, switch
is on and
is off. Diodes
and
are reverse biased while
and
conduct at time
. The input power
begins to transfer energy to the Y-impedance-network and recharge
, and the energy of
and
is delivered to
,
, and the load. The output voltage is the same as (
3). Furthermore, the voltages across inductors
,
, and the coupled inductor
are deduced as
and
(3) Mode 3: .
As shown in
Figure 3c, switch
is off and
is on. Similar to Mode 2,
and
are reverse biased while
and
conduct at time
. The voltage of inductors
,
, and the coupled inductor
are same as (
4) and (
5), respectively. The energy that capacitor
and the switched inductor cell (
,
) release to the resistive load and output voltage can be deduced as
In terms of voltage-second property in
and
, one can obtain
and
By substitutig (
1) and (
4) into (
7), we can deduce that
where
and
are duty cycles of switches
and
, respectively.
Substitute (
2) and (
5) into (
8), and we can deduce that
One can obtain the voltages of
and
, i.e.,
and
, via the solution of (
9) and (
10). Similarly, in terms of ampere second property in capacitor
, one can obtain
where
is the current flowing through
.
Denote the errors of
and
as
and
, respectively. As shown in
Figure 3, we can obtain that
,
and
. Therefore, it can be deduced that
on the basis of
, and (
11) can be derived as
where
and
R is the resistive load.
Thus, the output voltage
can be deduced as
where
.
According to (
13), it is noted that positive output voltage is equal to negative only if
. By adjusting the duty ratio of the switches and the winding ratio of the coupled inductors, we can obtain asymmetric and symmetric voltages, and positive and negative peak output voltages. When
, the proposed converter performs as a buck converter; when
, it functions as a boost converter. Therefore, it acts as a buck-boost converter.
In the same way, the output voltages of the inductor Y-impedance-network, quasi-Z-source, and switched-coupled-inductor Y-impedance-network are summarized in
Table 1, in which
and
.
The key waveforms of the proposed converter are depicted in
Figure 4, where
and
stand for the driving voltages of switches
and
, respectively;
,
, and
are the currents across diodes
,
, and
, respectively;
and
are the currents across inductors
and
, respectively;
and
are the voltages of capacitors
,
, respectively; and
is the output voltage. As shown in
Figure 4, it is obvious that the output voltage of the proposed converter exceeds the limited output voltages of the traditional half-bridge converter which are
and
.
4. Simulation Verifications
To verify the feasibility and validity of the proposed converter, PSIM software was applied for the simulation.
The preassigned parameters are as listed as follows: input voltage
V, resistive load
, permitted fluctuation range
= 1%,
= 10%, and period
= 100
s. The high harmonic frequency of the capacitance and inductance are approximately equal to the switching frequency of the converter. Therefore,
,
and
can be deduced as
and
Thus, capacitors and inductors are designed as
and
Then, based on the analysis in
Section 2, the parameters can be expressed as
Therefore, the parameters of the converter are chosen as follows: H, :: = 5:1:3, = = 47 F, = = 470 F, and H.
Simulations are shown in
Figure 6, and the output voltage of the proposed converter
60 V, which is consistent with the theoretical analyses, is shown in
Figure 4. Under the same conditions, as documented in [
7], the voltage gain of the conventional converter can be calculated as 0.735 and the output voltage
7.35 V, which coincides with the simulations in
Figure 6b. As shown in
Figure 6, by adjusting the turns ratio and duty cycle, the voltage gain of the proposed converter can be eight times higher than that of the conventional.
5. Experimental Verifications
In this section, the experimental environment and physical prototype are established, which are shown in
Table 2 and
Figure 7, to verify the theoretical analyses. Experimental parameters are shown in
Table 3.
The experimental results are shown in
Figure 8, which contains gate-source voltages of switches
,
, the output voltage
. It is noted that the experimental results agree well with the theoretical analyses and simulation, demonstrating the functionality and feasibility of the proposed converter.
In order to demonstrate the practicality and implementability of the proposed converter comprehensively, key waveforms of the proposed converter with different turns ratios and duty ratios are shown in
Figure 9. Therein, with small inductance of the coupled inductor, the proposed converter operates in discontinuous conduction mode (DCM), as shown in
Figure 9a,b. When
, the negative and positive output voltages are asymmetric, as shown in
Figure 9c,d, which agrees well with above theoretically analyses.