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Electronics 2013, 2(1), 94112; doi:10.3390/electronics2010094
Published: 8 March 2013
Abstract
: A new auxiliary circuit that can be implemented in DCDC and ACDC ZVSPWM converters is proposed in the paper. The circuit is for ZVSPWM converters used in applications where highfrequency operation is needed and the load current is higher than that of typical ZVSPWM converters. In the paper, the operation of a new ZVSPWM converter is described, its steadystate operation is analyzed, and a procedure for its design is derived and then demonstrated. The feasibility of the new converter is confirmed by experimental results obtained from a prototype.1. Introduction
Many techniques that use an active auxiliary circuit to help the main switch of a singleswitch pulsewidth modulated (PWM) converter turn on with zerovoltage switching (ZVS) have been proposed [1,2,3,4,5,6,7,8,9,10,11,12,13,14,15,16,17,18,19,20,21,22,23,24,25,26,27]. These techniques reduce switching losses in the main power switch, reduce reverserecovery losses in the main power diode, and reduce EMI in the converter. The auxiliary circuit is typically placed parallel to the main switch (Figure 1), and is activated just before the main converter switch is to be turned on.
The circuit gradually diverts current away from the main power diode to eliminate diode reverse recovery current after it is activated. It then discharges the capacitance across the main switch so that the switch can be turned on with ZVS. Finally, the circuit is deactivated from the main power circuit shortly after the main switch is turned on, so that the converter operates as a conventional PWM converter for the remainder of the switching cycle. The auxiliary circuit components have lower ratings than those in the main power circuit, as the circuit is active for only a small portion of the switching cycle. This allows a device that can turn on with fewer switching losses than the main switch to be used as the auxiliary switch.
Previously proposed ZVSPWM converters, however, have at least one of the following drawbacks:
The auxiliary switch is turned off while it is conducting current, which generates switching losses and EMI that offset the benefits of the auxiliary circuit [1,5,7,10,13,14,22,23].
The auxiliary circuit causes the main switch or boost diode to operate with higher peak current stress and more circulating current, which increases conduction losses and results in the need for a higher currentrated device for the main switch [2,3,6,8,11,12,15,21,22,23].
The auxiliary circuit components have high peak voltage stresses (at least twice the output voltage) and/or current stresses [2,3,6,8,12].
Energy from the output, which contributes to circulating current and losses, must be placed into the auxiliary circuit to trigger a resonant process [16,17].
It is standard practice in industry to implement a PWM converter with several MOSFETs in parallel to reduce the onstate resistance of the main power switch and thus its conduction losses. Examples of such an implementation are shown in Figure 2a,b for a DCDC PWM boost converter and a threephase ACDC converter respectively. Although using a single IGBT as the boost switch may be cheaper, IGBTs cannot operate with switching frequencies as high as those that MOSFETs can so that the size of the magnetic and filtering components (and thus the converter size) cannot be made as small. Small converter size is necessary for industrial applications such as telecom power converters that are part of power systems that are placed in cabinets where space is a major issue. An auxiliary circuit can be used to reduce switching losses in a paralleled MOSFET converter that operates with higher power and current, but the abovementioned drawbacks become worse than what they are for lower power converters.
For example, the turnoff losses of the auxiliary switch in a converter with a nonresonant auxiliary circuit (i.e., Figure 1a) are considerable. If resonant (Figure 1b and dual Figure 1c) approaches are used to ensure the soft turnoff of the auxiliary switch, then other problems arise, as can be seen from the auxiliary inductor waveforms shown in Figure 3. The negative part of the waveform for the Figure 1b circulates in the main switches and increases their peak stresses and conduction losses. The current waveform for the Figure 1c converter (I_{Lr1}) has an extremely high peak, at least double the input current, which makes it difficult to find an appropriate device for the auxiliary switch to carry this current.
A new auxiliary circuit for ZVSPWM converters that are implemented with paralleled MOSFETs for higher current applications is proposed in the paper. The circuit is shown in Figure 4. Although almost all previously proposed auxiliary circuits contain only a single active switch because of cost (it is difficult to justify a twoswitch circuit in a converter with a single MOSFET as the power switch), the proposed auxiliary circuit can be justified on the following grounds:
Its performance is superior to all other singleswitch auxiliary circuits for higher current applications because its switches can be turned off softly and it can operate with greater flexibility than singleswitch resonant and dual auxiliary circuits. Resonant and dual auxiliary circuits have issues related to the timing of the operation of the auxiliary switch relative to that of the main power switch(es) as the time window of opportunity to turn the auxiliary switch softly varies considerably from light load to heavy load. In other words, ZVSPWM converters with singleswitch auxiliary circuits like the ones shown in Figure 1 are not suitable for higher current applications and should not be used for these applications.
Cost is less of an issue and performance is the key criterion in applications where multiple MOSFETs are used. If the cost of multiple MOSFETs to improve performance can be justified for the power switch, then it can be justified in the auxiliary circuit.
Twoswitch auxiliary circuits for ZCSPWM IGBT converters are commonly used in high current applications and there is a vast literature about them [19]. Most multiswitch auxiliary circuits for ZCSPWM converters that have been proposed have been for threephase bucktype rectifiers and threephase current source inverters. In the case of a threephase rectifier, as shown in Figure 5a, the auxiliary circuit can be placed either across the dc link inductor (Position A) or across the output of the bridge (Position B). Several multiswitch auxiliary circuits are shown in Figure 5b,c. Given that multiswitch auxiliary circuits are widely used in higher power ZCSPWM applications to improve performance, the use of such circuits in higher power ZVSPWM applications where paralleled MOSFETs are used can be justified for the same reason.
In the paper, the operation of the new converter is described, its steadystate operation is analyzed, and a procedure for its design is derived and then demonstrated with an example. The feasibility of the new converter is confirmed by experimental results obtained from a prototype converter.
2. Modes of Operation
The proposed converter in Figure 4 has an auxiliary circuit that consists of two switches, S_{aux1} and S_{aux2}, three diodes, and a resonant tank made of capacitor C_{r} and inductor L_{r}. The basic operating principles of the proposed circuit are as follows: Auxiliary switch S_{aux1} is turned on just before the main power switch S is to be turned on, thus diverting current away from the main power diode D. Once current has been completely diverted away from D, the output capacitances of the switch begin to discharge and the voltage across it eventually falls to zero. The main power switch can be turned on with ZVS as soon as the capacitance is fully discharged. Due to the C_{r}L_{r} resonant tank, the current in the auxiliary circuit naturally falls to zero, thus allowing S_{aux1} to turn off with ZCS.
Sometime during the switching cycle, while the main power switch is conducting the input current, auxiliary switch S_{aux2} is turned on. This action results in the voltage across C_{r} flipping polarity so that it is negative instead of positive. When the main power switch is turned off, the input current completely discharges C_{r} so that there is no voltage across it when the auxiliary circuit is reactivated sometime during the next switching cycle. Equivalent circuit diagrams of the modes of operation that the converter goes through during a switching cycle are shown in Figure 6, and typical converter waveforms are shown in Figure 7. To save on space, switches S_{1}, S_{2}, and S_{3} are shown in Figure 6 as a single switch, S_{123}.
The converter’s modes of operation are as follows:
Mode 0 (t < t_{0}): All converter switches are off during this mode and current is flowing through the main power diode D.
Mode 1 (t_{0} < t < t_{1}): At t = t_{0}, switch S_{aux1} is turned on and current begins to be transferred away from diode D to the auxiliary circuit. This current transfer is gradual due to the presence of inductor L_{r} in the auxiliary circuit, so that charge is removed at a sufficiently slow rate to allow diode D to recover; this helps minimize reverse recovery current. The equations that represent the auxiliary circuit inductor current I_{Lr} and the auxiliary circuit capacitor voltage V_{Cr} in this mode are:
It should be noted that current can flow through the output capacitor of S_{aux2} after S_{aux1} is turned on. In order to minimize a sudden increase in current through this capacitor that can cause voltage spikes to appear, a saturable reactor or “spikekiller” inductor (L_{s}) should be placed in series with S_{aux2}.
Mode 2 (t_{1} < t < t_{2}): At t = t_{1}, current stops flowing through the main power diode D and the net capacitance across S_{123} begins to be discharged through L_{r} and C_{r}. The current in the auxiliary circuit is the sum of the input current and the current due to the discharging of the capacitances across S_{123}. The equations that describe the auxiliary circuit inductor current I_{Lr}, the voltage across S_{123}, V_{Cs}, and the auxiliary circuit capacitor voltage V_{Cr} in this mode are:
During this mode, the auxiliary circuit inductor current I_{Lr}, reaches its peak when Vc_{s} − Vc_{r} = 0 and it is equal to the peak current of S_{aux1} so that
Mode 3 (t_{2} < t < t_{3}): At t = t_{2}, the capacitance across the main power switches is completely discharged and current begins to flow through the body diodes of the devices; this allows the switches to be turned on with ZVS. The equations that describe the auxiliary circuit inductor current I_{Lr} and the auxiliary circuit capacitor voltage V_{Cr} in this mode are:
Mode 4 (t_{3} < t < t_{4}): At t = t_{3}, the current that was flowing in the body diodes of the main power switches in the previous mode reverses direction and begins to flow through the switches. The modal equations of this mode are the same as those of the previous mode except that the direction of the current through the main power switches is different.
Mode 5 (t_{4} < t < t_{5}): Current stops flowing in the auxiliary circuit at t = t_{4} due to the resonant interaction between L_{r} and C_{r}. Switch S_{aux1} can be turned off softly with zerocurrent switching (ZCS) sometime soon afterwards. The converter then operates like a standard PWM boost converter. The voltage across C_{r} remains fixed until S_{aux2} is turned on later in the switching cycle.
Mode 6 (t_{5} < t < t_{6}): At t = t_{5}, auxiliary switch S_{aux2} is turned on, sometime before the main power switches are turned off. As a result, capacitor C_{r} begins to discharge through L_{r}, S_{aux2} and D_{2}, and the voltage that was across it at the start of the mode changes polarity. At the end of this mode, the current in C_{r} and L_{r} is zero so that S_{aux2} can be turned off with ZCS. The equations that define this mode are:
Mode 7 (t_{6} < t < t_{7}): The output capacitance of S_{aux1} needs to be charged after this switch has been turned off so that current continues to flow through L_{r} and C_{r}. The length of this mode is negligible compared to the length of the other modes given that the output capacitance of S_{aux1} is much smaller than C_{r}, but the voltage across C_{r} can be changed during this mode. The voltage across S_{aux1} during this mode can be expressed as
It should be mentioned that if the output capacitance of S_{aux1} is charged to less than the output voltage V_{o} during this mode, then it would be charged up to V_{o} during Mode 9 when the main power switches turn off.
Mode 8 (t_{7} < t < t_{8}): During this mode, the main power switches are still on and current in the input inductor rises.
Mode 9 (t_{8} < t < t_{9}): At t = t_{8}, the main power switches are all turned off. The voltage of the net capacitance across the main power switches is
As a result, the auxiliary circuit capacitor C_{r} begins to be discharged as the net capacitance across the main switches continues to be charged; the energy stored in C_{r} is transferred to the output during this mode. The mode ends at t = t_{9} and the converter enters Mode 0, where it remains until S_{aux1} is turned on.
3. SteadyState Characteristics
The modal equations that are derived in the previous section of the paper can be used to generate steadystate characteristic curves that can be used to see the effect of certain key parameters on the operation of the auxiliary circuit. These key parameters include the values of auxiliary circuit components L_{r} and C_{r} and the net capacitance across the main power switches, C_{S}. Examples of such graphs are shown in Figure 8. Each graph has been generated by keeping certain parameters constant, then varying other parameters to see the effect of doing so.
Figure 8a is a graph of V_{cr} vs. L_{r} for different values of C_{S} with C_{r} = 50 nF. This graph shows that V_{cr} increases as either C_{S} or L_{r} is increased. The first characteristic can be explained by noting that increasing C_{S} increases the amount of energy that is discharged into the auxiliary circuit and is stored in C_{r} after the main power diode stops conducting. More energy in C_{r} results in higher values of V_{cr}. On the other hand, according to (4) and (12), higher values of L_{r} increase the time duration between t_{0} and t_{2}; therefore, more energy is transferred to C_{r}, which leads to higher values of V_{cr}.
Figure 8b is a graph of characteristic curves of I_{Lr} vs. L_{r} for different values of C_{S} with C_{r} = 50 nF. This graph shows that when C_{S} increases, more energy is stored in C_{r}, which results in higher peak values for I_{Lr}. Moreover, when L_{r} increases, it extends the resonant cycle and reduces the peak value of I_{Lr}. The average value of the resonant current is related to C_{S} and load current and is independent of length of the resonant cycle and the peak of the resonant current.
Figure 8c shows a graph of characteristic curves of ZVS time values vs. L_{r} for different values of C_{S} with C_{r} = 50 nF. These time values are when the net capacitance across the main power switches is completely discharged after S_{aux1} is turned on and is measured from the turnon instant of this switch. The graph shows that the ZVS times increase as L_{r} or C_{r} increases. Increasing L_{r} increases the time needed for current to be transferred away from the main power diode and it also increases the resonant cycle of the auxiliary circuit. On the other hand, by increasing C_{S}, the amount of stored energy in this capacitor increases and, therefore, it takes more time for it to be discharged.
Figure 8d shows a graph of characteristic curves of V_{aux1} vs. L_{r} for different values of C_{r} when C_{S} = 2 nF. It can be seen that increasing L_{r} increases the maximum voltage across S_{aux1}. Before D_{1} goes off after Mode 7 and V_{aux1} becomes constant, V_{aux1} is
Equation (26) shows that V_{aux1} is increased by increasing L_{r}. Also, for the same amount of energy transferred to C_{r}, increasing C_{r} reduces the voltage across it and thus reduces V_{aux1} as well.
4. Design Procedure and Example
Steadystate characteristic curves like the ones shown in Figure 8 can be used to develop a procedure that can be used to select key component values. Such a procedure is developed in this section of the paper. The procedure shown here will not consider the design of the main boost power circuit as the design of a standard PWM boost converter is wellknown—this includes the selection of the number and the type of device to be used for the main power switches. Moreover, it will be assumed that the net output capacitance across the paralleled main power devices is sufficient to slow down the rise in voltage across them after turnoff so that additional external capacitance is not needed.
4.1. Selection of Auxiliary Circuit Inductor L_{r}
The minimum value of L_{r} is determined by the inductor’s ability to limit the reverse recovery current of the main power boost diode. The reverse recovery current of this diode can be significantly reduced if the transition of current away from the diode to the auxiliary circuit is made to be gradual. The rate of current transfer is dependent on the value of L_{r} so that the larger the value of L_{r} is, the less recovery current there will be.
According to [20], an approximate rule of thumb that can be used for the determination of a minimum value of L_{r} is to make the current transfer time to be at least three times the reverse recovery time of the diode, t_{rr}. This can be expressed as
It should be noted that the voltage across C_{r} is zero at the time that S_{aux1} is turned on so that V_{Lr} is equal to the output voltage V_{o} at this moment. As current is transferred to the auxiliary circuit, V_{cr} changes as does V_{Lr} so that it is no longer equal to V_{o}. It is assumed in (27) that C_{r} is sufficiently large so that the change in V_{cr} and thus in V_{Lr} is small during the current transfer time.
Another thing to note about the value of L_{r} is that it cannot be too large. Very large values of L_{r} can result in increased peak voltage stresses in S_{aux1} and can increase the time required to discharge the net capacitances across the paralleled main switches, according to the graphs in Figure 8. As a result, the value of L_{r} should be close to the determined minimum value.
4.2. Selection of Auxiliary Circuit Capacitor C_{r}
If the value of C_{r} is too small, then the peak voltage stress of diode D_{3} increases which is equal to V_{o} + V_{Cr,P} where
If the value of C_{r} is too large, however, then the length of time needed to change the polarity of C_{r} after S_{aux2} may become excessive and thus place a limit on the acceptable range of duty cycle D that the main switches can operate with. A tradeoff must be considered when selecting a value for C_{r} and the extent of this tradeoff can be determined from steadystate characteristic curves like the ones shown in Figure 8.
4.3. Selection of Auxiliary Switches S_{aux1} and S_{aux2}
The maximum voltage across S_{aux1} is shown in Figure 8d and this switch should handle maximum drainsource voltage equal to 500 V. Maximum voltage across S_{aux2} is the output voltage V_{o}. The peak current of these switches can be found from graphs of steadystate characteristic curves like the one shown for I_{Lr} in Figure 8b.
4.4. Design Example
The following example is given to clarify the design procedure. The converter is to be designed according to the following specifications: Input voltage V_{in} = 75 Vdc, output voltage V_{o} = 375 Vdc, maximum output power P_{o,max} = 700 W, and switching frequency f_{s} = 100 kHz. The devices that are to be used in the main power circuit are two paralleled IRFP460 MOSFETs for the main power switch and a 15ETX06 device for the main power diode. It will be assumed that the efficiency of the converter is about 90%, that C_{S} = 2 nF, and the reverse recovery time of the main boost diode is t_{rr} = 75 ns and that the maximum voltage across S_{aux1} is not be more than 450 V so that a 500 V MOSFET device can be used for this switch.
The steps of the procedure are as follows:
 (1)

Based on abovementioned characteristics of the converter, the current of the boost inductor when S_{aux1} turns on is
 (2)

The value of L_{r} can be fixed by using (27)
Referring to Figure 8a, in order to keep the value of V_{cr} less than 200 V, the value of L_{r} should not be more than 8.2 μH. The maximum reverse voltage across D_{3} is V_{o} + V_{cr}, which is
 (3)

Referring to Figure 8c, the ZVS time with the selected C_{r} and L_{r} is 450 ns. This is the time window during which the main power switches can be turned on with ZVS.
 (4)

Referring to Figure 8b, with the selected value for L_{r} and output capacitance of the main switches which is 2 nF, the peak current in L_{r} is 15 A. This means that the auxiliary switches should be able to handle this peak current.
 (5)

Figure 8d shows that if the L_{r} is 8.2 μH, C_{r} should be less than 45 nF to keep the maximum voltage across S_{aux1} about 450 V, which allows MOSFETS with maximum drainsource voltage equal to 500 V to be used as S_{aux1}. It should be noted that maximum voltage of S_{aux2} is equal to output voltage.
5. Variations of the Auxiliary Circuit
The basic structure of the new auxiliary circuit can be modified in several ways, either to improve performance or to reduce cost. Several of these modified circuits are shown in Figure 9.
The circuit shown in Figure 9a is an auxiliary circuit that has a different resonant inductance when auxiliary switch S_{aux2} is conducting current than when S_{aux1} is conducting current. Having the resonant inductance be different under these two sets of circumstances allows the resonant inductance to be tailored to achieve the best performance for each set of circumstances.
The circuits shown in Figure 9b,c are auxiliary circuits with transformers in them. The presence of a transformer in the auxiliary circuit helps reduce circulating current losses after S_{aux2} has been turned on. The transformer allows energy to be transferred from the auxiliary circuit to the load instead of it just being trapped in the auxiliary circuit, where is contributes to losses. Moreover, the presence of a transformer in the auxiliary circuit makes the design of this circuit more flexible as it provides an additional degree of freedom.
The circuit shown in Figure 9d is a simplified version of the basic auxiliary circuit. The circuit has been simplified by Figure 4. What this does is to make current continuously flow in the auxiliary circuit inductor. Since this can happen, auxiliary circuit conduction losses may increase. This circuit variation can be considered for lower currents if it is desired to save on the cost of two diodes.
Figure 10 shows a graph of ZVS time interval vs. auxiliary circuit inductor value for the basic auxiliary and for the circuit in Figure 9c, implemented with an auxiliary circuit transformer turns ratio of n = N_{2}/N_{1} = 5. It can be seen that implementing the auxiliary circuit with a transformer can result in extending the amount of time that is available for the main converter switches to turn on with ZVS. This allows for greater flexibility in the design and the performance of the converter.
6. Experimental Results
An experimental proofofconcept prototype of the proposed converter was built to confirm its feasibility. The converter was built according to the same specifications as in the design example with input voltage V_{in} = 70 V, output V_{o} = 375 V, maximum output power P_{o,max} = 700 W and switching frequency f_{sw} = 100 kHz. The main power boost circuit was implemented as described in the design example. IRFP840 MOSFETs were used for the two auxiliary switches and 15ETX06 diodes for diodes D_{1}, D_{2}, D_{3}. The values of L_{r} and C_{r} were L_{r} = 8.2 μH and C_{r} = 44 nF.
Typical experimental waveforms are shown in Figure 11. Figure 11a,b shows the current waveform of L_{r}, I_{Lr}, and the gating signals of the two auxiliary switches. Since the positive part of I_{Lr} and the negative part of I_{Lr} represent the currents through S_{aux1} and S_{aux2} respectively, it can be seen that both switches can be turned off softly with ZCS. Figure 11c shows the gating signal and the drain source voltage of a main power switch. It can be seen that the switch turns on with ZVS, as the voltage across the switch is zero before it is turned on. Figure 11d shows the auxiliary inductor current and capacitor voltage waveforms. It can be seen that whatever energy is placed in C_{r} is removed before the auxiliary circuit is reactivated.
Figure 12 shows a graph of converter efficiency vs. load for the cases of the prototype with and without the basic auxiliary circuit. It can be seen that the converter efficiency dips sharply when the converter is operating with a heavy load while this does not happen when the converter has the basic auxiliary circuit. The comparison was limited to 700 W as the efficiency of the hardswitched converter becomes very poor past this point, while this is not the case for the ZVS converter. The reason for the sharp fall off in efficiency is the fact that the converter was operated with a high input current—higher than what is normally considered for a boost converter [15]. High current boost applications can include boost converters for solar power systems and telecom systems.
For the low power range, the efficiency of the hard switching circuit is better because there is relatively little current in the circuit so that turnon switching losses, which depend on the product of switch voltage and current during switch turnon, are low. Moreover, the hardswitching converter does not have any losses that are caused by an auxiliary circuit, which consumes some energy. When the load is low, the energy saved by the auxiliary circuit is less than its consumption. For the high power range, the turnon losses for the hardswitching converter become very high and are greater than the power consumed by the auxiliary circuit of the softswitching converter, which has ZVS, so that the softswitching converter is more efficient.
7. Conclusion
A new auxiliary circuit for ZVSPWM converters that are implemented with paralleled MOSFETs for higher current applications was proposed in the paper. The auxiliary circuit has two auxiliary switches to overcome the drawbacks of typical standard singleswitch auxiliary circuits that are limited to lower current applications.
In the paper, the operation of a ZVSPWM boost converter implemented with the new auxiliary circuit was described, its steadystate operation was analyzed, and a procedure for its design was derived and then demonstrated with an example. The feasibility of the new converter was confirmed by experimental results obtained from a prototype converter.
It should be noted that the proposed converter is being proposed for higher current applications where ZVSPWM converters are not typically used. For typical ZVSPWM converter applications, simpler, cheaper, and more conventional approaches are more suitable.
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