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A new auxiliary circuit that can be implemented in DC-DC and AC-DC ZVS-PWM converters is proposed in the paper. The circuit is for ZVS-PWM converters used in applications where high-frequency operation is needed and the load current is higher than that of typical ZVS-PWM converters. In the paper, the operation of a new ZVS-PWM converter is described, its steady-state operation is analyzed, and a procedure for its design is derived and then demonstrated. The feasibility of the new converter is confirmed by experimental results obtained from a prototype.

Many techniques that use an active auxiliary circuit to help the main switch of a single-switch pulse-width modulated (PWM) converter turn on with zero-voltage switching (ZVS) have been proposed [

The circuit gradually diverts current away from the main power diode to eliminate diode reverse recovery current after it is activated. It then discharges the capacitance across the main switch so that the switch can be turned on with ZVS. Finally, the circuit is deactivated from the main power circuit shortly after the main switch is turned on, so that the converter operates as a conventional PWM converter for the remainder of the switching cycle. The auxiliary circuit components have lower ratings than those in the main power circuit, as the circuit is active for only a small portion of the switching cycle. This allows a device that can turn on with fewer switching losses than the main switch to be used as the auxiliary switch.

Zero-voltage switching (ZVS)-pulse-width modulated (PWM) boost converters with auxiliary circuits. (

Previously proposed ZVS-PWM converters, however, have at least one of the following drawbacks:

The auxiliary switch is turned off while it is conducting current, which generates switching losses and EMI that offset the benefits of the auxiliary circuit [

The auxiliary circuit causes the main switch or boost diode to operate with higher peak current stress and more circulating current, which increases conduction losses and results in the need for a higher current-rated device for the main switch [

The auxiliary circuit components have high peak voltage stresses (at least twice the output voltage) and/or current stresses [

Energy from the output, which contributes to circulating current and losses, must be placed into the auxiliary circuit to trigger a resonant process [

It is standard practice in industry to implement a PWM converter with several MOSFETs in parallel to reduce the on-state resistance of the main power switch and thus its conduction losses. Examples of such an implementation are shown in

Converters with paralleled MOSFETs. (

For example, the turn-off losses of the auxiliary switch in a converter with a non-resonant auxiliary circuit (_{Lr1}) has an extremely high peak, at least double the input current, which makes it difficult to find an appropriate device for the auxiliary switch to carry this current.

Typical auxiliary inductor current waveforms for ZVS-PWM boost converters operating with input voltage _{in} = 100 V, output voltage _{o} = 400 V, output power _{o} = 2 kW, and switching frequency _{sw} = 100 kHz. Scale:

A new auxiliary circuit for ZVS-PWM converters that are implemented with paralleled MOSFETs for higher current applications is proposed in the paper. The circuit is shown in

Its performance is superior to all other single-switch auxiliary circuits for higher current applications because its switches can be turned off softly and it can operate with greater flexibility than single-switch resonant and dual auxiliary circuits. Resonant and dual auxiliary circuits have issues related to the timing of the operation of the auxiliary switch relative to that of the main power switch(es) as the time window of opportunity to turn the auxiliary switch softly varies considerably from light load to heavy load. In other words, ZVS-PWM converters with single-switch auxiliary circuits like the ones shown in

Cost is less of an issue and performance is the key criterion in applications where multiple MOSFETs are used. If the cost of multiple MOSFETs to improve performance can be justified for the power switch, then it can be justified in the auxiliary circuit.

Two-switch auxiliary circuits for ZCS-PWM IGBT converters are commonly used in high current applications and there is a vast literature about them [

Proposed DC-DC boost converter.

(

In the paper, the operation of the new converter is described, its steady-state operation is analyzed, and a procedure for its design is derived and then demonstrated with an example. The feasibility of the new converter is confirmed by experimental results obtained from a prototype converter.

The proposed converter in _{aux1} and S_{aux2}, three diodes, and a resonant tank made of capacitor C_{r} and inductor L_{r}. The basic operating principles of the proposed circuit are as follows: Auxiliary switch S_{aux1} is turned on just before the main power switch S is to be turned on, thus diverting current away from the main power diode D. Once current has been completely diverted away from D, the output capacitances of the switch begin to discharge and the voltage across it eventually falls to zero. The main power switch can be turned on with ZVS as soon as the capacitance is fully discharged. Due to the C_{r}-L_{r} resonant tank, the current in the auxiliary circuit naturally falls to zero, thus allowing S_{aux1} to turn off with ZCS.

Sometime during the switching cycle, while the main power switch is conducting the input current, auxiliary switch S_{aux2} is turned on. This action results in the voltage across C_{r} flipping polarity so that it is negative instead of positive. When the main power switch is turned off, the input current completely discharges C_{r} so that there is no voltage across it when the auxiliary circuit is reactivated sometime during the next switching cycle. Equivalent circuit diagrams of the modes of operation that the converter goes through during a switching cycle are shown in _{1}, S_{2}, and S_{3} are shown in _{123}.

Modes of operation. (

Typical waveforms.

The converter’s modes of operation are as follows:

_{0}): All converter switches are off during this mode and current is flowing through the main power diode D.

_{0} _{1}): At _{0}, switch S_{aux1} is turned on and current begins to be transferred away from diode D to the auxiliary circuit. This current transfer is gradual due to the presence of inductor L_{r} in the auxiliary circuit, so that charge is removed at a sufficiently slow rate to allow diode D to recover; this helps minimize reverse recovery current. The equations that represent the auxiliary circuit inductor current _{Lr} and the auxiliary circuit capacitor voltage _{Cr} in this mode are:
_{Lr} and _{cr} at the beginning of this mode are zero.

It should be noted that current can flow through the output capacitor of S_{aux2} after S_{aux1} is turned on. In order to minimize a sudden increase in current through this capacitor that can cause voltage spikes to appear, a saturable reactor or “spike-killer” inductor (L_{s}) should be placed in series with S_{aux2}.

_{1} _{2}): At _{1}, current stops flowing through the main power diode D and the net capacitance across S_{123} begins to be discharged through L_{r} and C_{r}. The current in the auxiliary circuit is the sum of the input current and the current due to the discharging of the capacitances across S_{123}. The equations that describe the auxiliary circuit inductor current _{Lr}, the voltage across S_{123}, _{Cs}, and the auxiliary circuit capacitor voltage _{Cr} in this mode are:

During this mode, the auxiliary circuit inductor current _{Lr}, reaches its peak when _{s} − _{r} = 0 and it is equal to the peak current of S_{aux1} so that

_{2} < t < t_{3}_{2}_{Lr} and the auxiliary circuit capacitor voltage _{Cr} in this mode are:

_{3} _{4}): At _{3}, the current that was flowing in the body diodes of the main power switches in the previous mode reverses direction and begins to flow through the switches. The modal equations of this mode are the same as those of the previous mode except that the direction of the current through the main power switches is different.

_{4} _{5}): Current stops flowing in the auxiliary circuit at _{4} due to the resonant interaction between L_{r} and C_{r}. Switch S_{aux1} can be turned off softly with zero-current switching (ZCS) sometime soon afterwards. The converter then operates like a standard PWM boost converter. The voltage across C_{r} remains fixed until S_{aux2} is turned on later in the switching cycle.

_{5} _{6}): At _{5}, auxiliary switch S_{aux2} is turned on, sometime before the main power switches are turned off. As a result, capacitor C_{r} begins to discharge through L_{r}, S_{aux2} and D_{2}, and the voltage that was across it at the start of the mode changes polarity. At the end of this mode, the current in C_{r} and L_{r} is zero so that S_{aux2} can be turned off with ZCS. The equations that define this mode are:

_{6} _{7}): The output capacitance of S_{aux1} needs to be charged after this switch has been turned off so that current continues to flow through L_{r} and C_{r}. The length of this mode is negligible compared to the length of the other modes given that the output capacitance of S_{aux1} is much smaller than C_{r}, but the voltage across C_{r} can be changed during this mode. The voltage across S_{aux1} during this mode can be expressed as

It should be mentioned that if the output capacitance of S_{aux1} is charged to less than the output voltage _{o} during this mode, then it would be charged up to _{o} during Mode 9 when the main power switches turn off.

_{7} _{8}): During this mode, the main power switches are still on and current in the input inductor rises.

_{8} _{9}): At _{8}, the main power switches are all turned off. The voltage of the net capacitance across the main power switches is

As a result, the auxiliary circuit capacitor C_{r} begins to be discharged as the net capacitance across the main switches continues to be charged; the energy stored in C_{r} is transferred to the output during this mode. The mode ends at _{9} and the converter enters Mode 0, where it remains until S_{aux1} is turned on.

The modal equations that are derived in the previous section of the paper can be used to generate steady-state characteristic curves that can be used to see the effect of certain key parameters on the operation of the auxiliary circuit. These key parameters include the values of auxiliary circuit components L_{r} and C_{r} and the net capacitance across the main power switches, C_{S}. Examples of such graphs are shown in

Characteristic curves. (_{cr} _{r} for different values of _{S} with _{r} = 50 nF; (_{Lr} _{r} for different values of _{S} with _{r} = 50 nF; (_{Lr} _{r} for different values of _{S} with _{r} = 50 nF; (_{aux1} _{r} for different values of _{r} with _{S} = 2 nF.

_{cr} _{r} for different values of _{S} with _{r} = 50 nF. This graph shows that _{cr} increases as either _{S} or _{r} is increased. The first characteristic can be explained by noting that increasing _{S} increases the amount of energy that is discharged into the auxiliary circuit and is stored in _{r} after the main power diode stops conducting. More energy in _{r} results in higher values of _{cr}. On the other hand, according to (4) and (12), higher values of _{r} increase the time duration between _{0} and _{2}; therefore, more energy is transferred to _{r}, which leads to higher values of _{cr}.

_{Lr} _{r} for different values of _{S} with _{r} = 50 nF. This graph shows that when _{S} increases, more energy is stored in _{r}, which results in higher peak values for _{Lr}. Moreover, when _{r} increases, it extends the resonant cycle and reduces the peak value of _{Lr}. The average value of the resonant current is related to _{S} and load current and is independent of length of the resonant cycle and the peak of the resonant current.

_{r} for different values of _{S} with _{r} = 50 nF. These time values are when the net capacitance across the main power switches is completely discharged after S_{aux1} is turned on and is measured from the turn-on instant of this switch. The graph shows that the ZVS times increase as _{r} or _{r} increases. Increasing _{r} increases the time needed for current to be transferred away from the main power diode and it also increases the resonant cycle of the auxiliary circuit. On the other hand, by increasing _{S}, the amount of stored energy in this capacitor increases and, therefore, it takes more time for it to be discharged.

_{aux1} _{r} for different values of _{r} when _{S} = 2 nF. It can be seen that increasing _{r} increases the maximum voltage across S_{aux1}. Before D_{1} goes off after Mode 7 and _{aux1} becomes constant, _{aux1} is

Equation (26) shows that _{aux1} is increased by increasing _{r}. Also, for the same amount of energy transferred to _{r}, increasing _{r} reduces the voltage across it and thus reduces _{aux1} as well.

Steady-state characteristic curves like the ones shown in

The minimum value of _{r} is determined by the inductor’s ability to limit the reverse recovery current of the main power boost diode. The reverse recovery current of this diode can be significantly reduced if the transition of current away from the diode to the auxiliary circuit is made to be gradual. The rate of current transfer is dependent on the value of _{r} so that the larger the value of _{r} is, the less recovery current there will be.

According to [_{r} is to make the current transfer time to be at least three times the reverse recovery time of the diode, _{rr}. This can be expressed as
_{Lr} is the voltage across boost inductor when S_{aux1} is turned on, and _{D} is the amount of current that passes through boost diode and that should be diverted into the auxiliary circuit.

It should be noted that the voltage across _{r} is zero at the time that S_{aux1} is turned on so that _{Lr} is equal to the output voltage _{o} at this moment. As current is transferred to the auxiliary circuit, _{cr} changes as does _{Lr} so that it is no longer equal to _{o}. It is assumed in (27) that _{r} is sufficiently large so that the change in _{cr} and thus in _{Lr} is small during the current transfer time.

Another thing to note about the value of _{r} is that it cannot be too large. Very large values of _{r} can result in increased peak voltage stresses in S_{aux1} and can increase the time required to discharge the net capacitances across the paralleled main switches, according to the graphs in _{r} should be close to the determined minimum value.

If the value of _{r} is too small, then the peak voltage stress of diode D_{3} increases which is equal to _{o} + _{Cr,P} where

If the value of _{r} is too large, however, then the length of time needed to change the polarity of _{r} after S_{aux2} may become excessive and thus place a limit on the acceptable range of duty cycle D that the main switches can operate with. A trade-off must be considered when selecting a value for _{r} and the extent of this trade-off can be determined from steady-state characteristic curves like the ones shown in

The maximum voltage across S_{aux1} is shown in _{aux2} is the output voltage _{o}. The peak current of these switches can be found from graphs of steady-state characteristic curves like the one shown for _{Lr} in

The following example is given to clarify the design procedure. The converter is to be designed according to the following specifications: Input voltage _{in} = 75 Vdc, output voltage _{o} = 375 Vdc, maximum output power _{o,max} = 700 W, and switching frequency _{s} = 100 kHz. The devices that are to be used in the main power circuit are two paralleled IRFP460 MOSFETs for the main power switch and a 15ETX06 device for the main power diode. It will be assumed that the efficiency of the converter is about 90%, that _{S} = 2 nF, and the reverse recovery time of the main boost diode is _{rr} = 75 ns and that the maximum voltage across S_{aux1} is not be more than 450 V so that a 500 V MOSFET device can be used for this switch.

The steps of the procedure are as follows:

Based on above-mentioned characteristics of the converter, the current of the boost inductor when S_{aux1} turns on is

The value of _{r} can be fixed by using (27)

Referring to _{cr} less than 200 V, the value of _{r} should not be more than 8.2 μH. The maximum reverse voltage across D_{3} is _{o} + _{cr}, which is

Referring to _{r} and _{r} is 450 ns. This is the time window during which the main power switches can be turned on with ZVS.

Referring to _{r} and output capacitance of the main switches which is 2 nF, the peak current in _{r} is 15 A. This means that the auxiliary switches should be able to handle this peak current.

_{r} is 8.2 μH, _{r} should be less than 45 nF to keep the maximum voltage across S_{aux1} about 450 V, which allows MOSFETS with maximum drain-source voltage equal to 500 V to be used as S_{aux1}. It should be noted that maximum voltage of S_{aux2} is equal to output voltage.

The basic structure of the new auxiliary circuit can be modified in several ways, either to improve performance or to reduce cost. Several of these modified circuits are shown in

Modified structures of new aux circuit.

The circuit shown in _{aux2} is conducting current than when S_{aux1} is conducting current. Having the resonant inductance be different under these two sets of circumstances allows the resonant inductance to be tailored to achieve the best performance for each set of circumstances.

The circuits shown in _{aux2} has been turned on. The transformer allows energy to be transferred from the auxiliary circuit to the load instead of it just being trapped in the auxiliary circuit, where is contributes to losses. Moreover, the presence of a transformer in the auxiliary circuit makes the design of this circuit more flexible as it provides an additional degree of freedom.

The circuit shown in

_{2}/_{1} = 5. It can be seen that implementing the auxiliary circuit with a transformer can result in extending the amount of time that is available for the main converter switches to turn on with ZVS. This allows for greater flexibility in the design and the performance of the converter.

Graph of characteristic curves of ZVS time values _{r} = 50 nF and

An experimental proof-of-concept prototype of the proposed converter was built to confirm its feasibility. The converter was built according to the same specifications as in the design example with input voltage _{in} = 70 V, output _{o} = 375 V, maximum output power _{o,max} = 700 W and switching frequency _{sw} = 100 kHz. The main power boost circuit was implemented as described in the design example. IRFP840 MOSFETs were used for the two auxiliary switches and 15ETX06 diodes for diodes D_{1}, D_{2}, D_{3}. The values of _{r} and _{r} were _{r} = 8.2 μH and _{r} = 44 nF.

Typical experimental waveforms are shown in _{r}, _{Lr}, and the gating signals of the two auxiliary switches. Since the positive part of _{Lr} and the negative part of _{Lr} represent the currents through S_{aux1} and S_{aux2} respectively, it can be seen that both switches can be turned off softly with ZCS. _{r} is removed before the auxiliary circuit is reactivated.

Experimental waveforms. (_{GS} of S_{aux1} (_{r} resonant inductor (_{r} (_{GS} of S_{aux2} (_{GS} of main switch (_{r} (_{DS} of main switch (_{r} (_{r} (

For the low power range, the efficiency of the hard switching circuit is better because there is relatively little current in the circuit so that turn-on switching losses, which depend on the product of switch voltage and current during switch turn-on, are low. Moreover, the hard-switching converter does not have any losses that are caused by an auxiliary circuit, which consumes some energy. When the load is low, the energy saved by the auxiliary circuit is less than its consumption. For the high power range, the turn-on losses for the hard-switching converter become very high and are greater than the power consumed by the auxiliary circuit of the soft-switching converter, which has ZVS, so that the soft-switching converter is more efficient.

Efficiency

A new auxiliary circuit for ZVS-PWM converters that are implemented with paralleled MOSFETs for higher current applications was proposed in the paper. The auxiliary circuit has two auxiliary switches to overcome the drawbacks of typical standard single-switch auxiliary circuits that are limited to lower current applications.

In the paper, the operation of a ZVS-PWM boost converter implemented with the new auxiliary circuit was described, its steady-state operation was analyzed, and a procedure for its design was derived and then demonstrated with an example. The feasibility of the new converter was confirmed by experimental results obtained from a prototype converter.

It should be noted that the proposed converter is being proposed for higher current applications where ZVS-PWM converters are not typically used. For typical ZVS-PWM converter applications, simpler, cheaper, and more conventional approaches are more suitable.