Wide Voltage Resonant Converter Using a Variable Winding Turns Ratio

Abstract

This paper presents a inductor–inductor–capacitor (LLC) resonant converter with variable winding turns to achieve wide voltage operation (100–400 V) and realize soft switching operation over the entire load range. Resonant converters have been developed for consumer power units in computers, power servers, medical equipment, and adaptors due to the advantages of less switching loss and better circuit efficiency. The main disadvantages of the LLC resonant converter are narrow voltage range operation owing to wide switching frequency variation and limited voltage gain. For computer power supplies with hold-up time function, electric vehicle battery chargers, and for power conversion in solar panels, wide input voltage or wide output voltage operation capability is normally demanded for powered electronics. To meet these requirements, the variable winding turns are used in the presented circuit to achieve high- or low-voltage gain when V_in is at low- or high-voltage, respectively. Therefore, the wide voltage operation capability can be implemented in the presented resonant circuit. The variable winding turns are controlled by an alternating current (AC) power switch with two back-to-back metal-oxide-semiconductor field-effect transistors (MOSFETs). A 500-W prototype is implemented and test results are presented to confirm the converter performance.

1. Introduction

Renewable energy systems with electronic power techniques [1,2,3,4] have been widely researched to lessen the demands for fossil fuels and reduce the effects of global warming. Form the various renewable energy sources, solar and wind power are more attractive out of the clean energy sources. The main problem with solar and wind power is the non-constant output voltage from wind turbine generators and photovoltaic (PV) solar panels. The output voltage of a wind turbine generator is related to wind speed and the output voltage of a PV solar panel is related to solar intensity. High-efficiency direct current (DC) power converters [5,6,7,8] have been proposed for power units for computers, battery dischargers and chargers, telecommunication systems, and server systems. Out of the various soft switching techniques, the active clamp technique [9,10], asymmetric pulse width modulation (PWM) [11,12], and phase-shift PWM converters [13,14] are the most attractive circuit topologies with the lowest switching losses. The PWM scheme is widely used to control these circuit topologies with constant output voltage. However, the turn-on time of the PWM scheme is dependent on the input voltage. PWM converters have a low (high) duty cycle at high (low) voltage input. However, the low duty cycle results in a high root mean square current and high conduction loss. Thus, the PWM converters have low (high) circuit efficiency at high (low) input voltage conditions. This drawback limits PWM converters in terms of wide voltage operation. Pulse frequency modulation (PFM), or variable frequency control, has been adopted in resonant converters [15]. In many resonant circuit topologies, the inductor–inductor–capacitor (LLC) resonant converter is a more attractive circuit topology for use in high-input voltage inputs. The main advantages of LLC resonant converters [16,17,18] are low switching loss in power devices, high efficiency, and low electromagnetic interference (EMI). The main drawback of LLC resonant converters is the limited voltage gain. This means it is difficult to operate the resonant converter in wide voltage operation. To realize wide voltage operation, two-stage converters and series–parallel connection for full-bridge or half-bridge converters [19,20,21,22] have been proposed for battery charger, solar power, and wind power applications. The two-stage converters have the drawback of low efficiency, while the series- or parallel-connected converters have more circuit components, which increase the conduction loss and reduce the converter reliability. In [23,24], two full-bridge LLC converters with primary-parallel–secondary-series connection were proposed to achieve wide input or output voltage operation. However, eight power switches and eight diodes are used in this circuit topology. The conduction losses in power semiconductors and the cost are increased. An interleaved LLC resonant converter has been studied and implemented in [25] to realize wide input voltage operation and ripple-free input current. However, this circuit topology is a two-stage circuit structure, meaning that the circuit efficiency is low in full load conditions, when both the boost converter and LLC converter are operated. In [26], a half-bridge LLC resonant converter with two sets of center-tapped rectifiers was proposed to extend the hold-up time problem for personal computers. However, four winding turns are used on the secondary side in this circuit topology, and the copper losses on magnetic windings are increased.

An LLC resonant converter with variable winding turns is proposed and investigated to achieve soft switching operation on active switches over the entire load range and for wide voltage range operation (V_in = 100–400 V). The variable winding turns controlled by an alternating current (AC) switch are adopted to achieve different voltage gains based on low- or high-input voltage ranges. During the low-input voltage range (from V_in = 100 V to 200 V), the converter with more secondary turns is selected to achieve high-voltage gain for the LLC converter. During the input voltage range (V_in = 200–400 V), the converter with fewer secondary turns is operated to achieve low-voltage gain for the LLC converter. Therefore, the proposed converter can be operated under wide input voltage variation, such as for solar power or switching mode power supplies without power factor correction, under the universal AC input voltage. The LLC resonant circuit with pulse-frequency modulation is used to control active devices. Due to the inductive impedance load of the LLC resonant tank, the converter can realize soft switching operation for both active switches and rectifier diodes. The voltage double rectifier topology is adopted on the output side to limit the voltage rating of the secondary diodes at V_o instead of 2V_o in center-tapped rectifiers. The rectifier diodes with a low-voltage rating have low-voltage drop. Comparing the center-tapped rectifier structure, the voltage double rectifier structure has less conduction losses for rectifier diodes and secondary windings. Comparing the conventional two-stage power converters, parallel-series connected converters, and the other resonant converters in [19,20,21,22,23,24,25,26], the presented LLC resonant converter has a simple circuit structure, a simple control algorithm, and requires fewer power components to achieve wide voltage operation. The description of the presented circuit schematic and operation principle is presented in Section 2. The circuit analysis and design procedures of the presented resonant converter are provided in Section 3. Experiments are demonstrated and discussed to confirm the effectiveness of the presented converter in Section 4. Finally, the conclusions are discussed in Section 5.

2. Proposed Converter and Operation Principle

Figure 1a shows the conventional LLC converter. The half-bridge leg and center-tapped rectifier are used on the primary and secondary sides, respectively, for high-voltage input and low-voltage output applications. According to the center-tapped rectifier topology, the secondary diodes have 2V_o voltage stress. Figure 1b shows the circuit topology with the voltage double rectifier for medium- or high-voltage output applications. The advantages of this circuit topology are the low-voltage rating of the rectifier diodes (V_D1,rating = V_D1,rating = V_o) and fewer secondary winding turns (n_s instead of 2n_s) in the center-tapped rectifier structure. Therefore, the voltage double rectifier structure has less copper loss on the secondary windings.

In order to improve the drawback of the limit voltage range in conventional LLC converters, Figure 2a shows the circuit configuration of the presented LLC converter with variable winding turns to achieve wide input voltage operation (100–400 V). The input voltage of the proposed LLC converter can be from the output of the solar panels, wind turbine DC generators, or battery banks. The output voltages of battery banks and large solar panels vary widely. A half-bridge LLC resonant tank is used on the primary side for soft switching operation in active devices S₁ and S₂ over the entire load range. The voltage rating of the active devices equals the maximum input voltage. Two sets of voltage double rectifiers and one AC switch S_ac implemented with two metal-oxide-semiconductor field-effect transistors (MOSFETs) back-to-back connection are used on the output-side. If the proposed converter is operated under low-voltage input V_in,L (100–200 V), then S_ac is turned on (Figure 2b). Due to S_ac is conducting, the 2n_s secondary winding turns are connected to load voltage. One observes that D₃ and D₄ are off. The transformer turns-ratio under low-input voltage operation is n_p/(2n_s). The proposed converter has DC voltage gain V_o/V_in,L = G(f)(2n_s)/n_p, where V_in,L and G (f) are low-input voltage range and the voltage transfer function of LLC converter. When the circuit is worked under high-voltage range V_in,H (200–400 V), S_ac is controlled at off state (Figure 2c). The n_s winding turns are connected to load voltage and the diodes D₁ and D₂ are off. Therefore, the transformer turns-ratio for high-voltage operation is equal to n_p/n_s. The direct current voltage gain V_o/V_in,H of the LLC converter is G(f)n_s/n_p, where V_in,H is high-input voltage range. From the previous discussion, one can observe that V_o/V_in,L = G(f)(2n_s)/n_p = 2V_o/V_in,H. It is concluded that V_in,H = 2V_in,L, such as V_in,H = 200–400 V and V_in,L = 100–200 V. Therefore, the presented resonant converter has high (low) DC voltage gain operated at low (high) input voltage range with low (high) transformer turns-ratio. The PWM signals of S₁ and S₂ are controlled with pulse-frequency modulation scheme shown in Figure 3a and Figure 4a and the AC switch S_ac is controlled by the Schmitt voltage comparator. The reference voltage of the Schmitt voltage comparator is 200 V and the hysteresis voltage band is 10 V.

2.1. Low-Voltage Operation V_in,L = V_in,min–2V_in,min (S_ac: on, D₃, D₄: off)

If the converter is operated at the low-voltage input from V_in,min to 2V_in,min, then S_ac is turned on and D₃ and D₄ are turned off. The proposed converter has a DC voltage gain of V_o/V_in,L = G(f)(2n_s)/n_p. According to the switching states of S₁, S₂, D₁, and D₂, the converter has six operating modes for each switching cycle. The PWM waveforms are demonstrated in Figure 3a. Figure 3b–g gives the equivalent circuits of six operating modes under f_r1 (series resonant frequency) > f_sw (switching frequency).

Mode 1 [t₀ ~ t₁]: This mode begins at t₀ if v_CS1 = 0 V and v_CS2 = V_in. Owing to i_Lr(t₀) < 0 and i_Lr > i_Lm, the body diode D_S1 conducts and D₁ is forward-biased. Switch S₁ turns on after t₀ to achieve zero voltage switching. The magnetizing voltage v_Lm = n_pV_Co1/(2n_s) = n_pV_o/(4n_s) and the magnetizing current i_Lm increase in this mode. In mode 1, the series resonant frequency is $f_{r1}=1/2\pi \sqrt{L_r{C}_r}$. If f_r1 (series resonant frequency) < f_sw (switching frequency), then the circuit goes to mode 3. Otherwise, the circuit goes to mode 2 at time t₁.

Mode 2 [t₁ ~ t₂]: At t₁, i_D1 = 0 and D₁ turns off without the reverse recovery current. In mode 2, the resonant frequency is $f_{r2}=1/2\pi \sqrt{(L_r+L_m)C_r}$. It is clear that the resonant frequency f_r2 in mode 2 is less than the frequency f_r1 in mode 1. The peak-to-peak current of L_m is approximately obtained as

$$\Delta i_{Lm}\approx \frac{n_p{V}_o}{8n_s{L}_m{f}_{sw}}$$

(1)

Thus, the magnetizing current at time t₂ is given as

$$i_{Lm}(t_2)=\frac{\Delta i_{Lm}}{2}\approx \frac{n_p{V}_o}{16n_s{L}_m{f}_{sw}}$$

(2)

Mode 3 [t₂ ~ t₃]: This mode starts at t₂ when switch S₁ is turned off. Owing to i_Lr(t₂) > 0 and i_Lr < i_Lm, C_S2 (C_S1) is discharged (charged) from V_in (0 V) and D₂ is forward-biased. If the inductor energy E_Lr+Lm at time t₂ is larger than the capacitor energy E_CS1+CS2, then v_Cs2 will be decreased to zero voltage at time t₃. The zero voltage switching condition of S₂ is expressed as

$$i_{Lm}(t_2)\ge V_{in}\sqrt{2C_{oss}/(L_m+L_r)}$$

(3)

where C_oss = C_S1 = C_S2. The dead time (t_d) between S₁ and S₂ must be greater than the discharge time of C_S2 to ensure the zero voltage turn-on of S₂. Thus, the maximum value of L_m can be calculated as

$$L_m\le \frac{n_p{V}_o{t}_d}{32V_{in}{f}_s{n}_s{C}_{oss}}$$

(4)

Mode 4 [t₃ ~ t₄]: At time t₃, v_CS2 = 0 V and v_CS1 = V_in. Owing to i_Lr(t₃) > 0 and i_Lr < i_Lm, the body diode D_S2 conducts and D₂ is forward-biased. S₂ can turn on after t₃ with soft switching operation. In mode 4, v_Lm = −n_pV_Co2/(2n_s) = −n_pV_o/(4n_s) and i_Lm decreases. The resonant frequency in mode 4 is f_r1. If f_sw < f_r1, then i_D2 = 0 at time t₄.

Mode 5 [t₄ ~ t₅]: At t₄, i_D2 = 0 and D₂ turns off without the reverse recovery current. L_r, L_m, and C_r are naturally resonant with the frequency f_r2.

Mode 6 [t₅ ~ T_sw+t₀]: At t₅, S₂ is turned off. Owing to i_Lr(t₅) < 0 and i_Lr > i_Lm, C_S1 is discharged and D₁ is forward-biased. The soft switching turn-on condition of S₁ is expressed as

$$|i_{Lm}(t_5)|=i_{Lm}(t_2)\ge V_{in}\sqrt{2C_{oss}/(L_m+L_r)}$$

(5)

At time t₀ + T_sw, v_CS1 = 0 V.

2.2. High-Voltage Operation V_in,H = 2V_in,min–4V_in,min (S_ac: off, D₁, D₂: off)

If the converter is worked at the high-voltage input from 2V_in,min to 4V_in,min, then S_ac turns off, and D₁ and D₂ are also off. The DC voltage gain becomes V_o/V_in,H = G(f)n_s/n_p. According to the on–off states of S₁, S₂, D₃, and D₄, the LLC converter has six operating modes for each switching period. Figure 4 gives the PWM waveforms and equivalent circuits under high-input voltage operation.

Mode 1 [t₀ ~ t₁]: At time t₀, v_CS1 = 0 V. Since i_Lr > i_Lm and i_Lr(t₀) < 0, D_S1 is forward-biased and D₃ conducts. Thus, the zero voltage turn-on of S₁ can be realized after t₀. Due to D₃ being forward-biased, one can obtain v_Lm = n_pV_Co1/n_s = n_pV_o/(2n_s) and i_Lm increases. The resonant frequency in mode 1 is $f_{r1}=1/2\pi \sqrt{L_r{C}_r}$.

Mode 2 [t₁ ~ t₂]: At time t₁, i_D3 = 0 and D₃ turns off at t₁ without the reverse recovery current. The resonant frequency in mode 2 is $f_{r2}=1/2\pi \sqrt{(L_r+L_m)C_r}$. The peak-to-peak ripple current ΔL_m and the magnetizing current at time t₂ are approximately calculated as

$$\Delta i_{Lm}\approx \frac{n_p{V}_o}{4n_s{L}_m{f}_{sw}}$$

(6)

$$i_{Lm}(t_2)=\frac{\Delta i_{Lm}}{2}\approx \frac{n_p{V}_o}{8n_s{L}_m{f}_{sw}}$$

(7)

Mode 3 [t₂ ~ t₃]: At t₂, S₁ turns off. Due to i_Lr < i_Lm and i_Lr(t₂) > 0, D₄ conducts and C_S2 is discharged from V_in. If the inductor energy on L_r and L_m is larger than the capacitor energy on C_S1 and C_S2, then v_CS2 is decreased to zero voltage at t₃. The zero voltage turn-on condition of S₂ is expressed as

$$i_{Lm}(t_2)=\frac{n_p{V}_o}{8n_s{L}_m{f}_{sw}}\ge V_{in}\sqrt{2C_{oss}/(L_m+L_r)}$$

(8)

If the dead time (t_d) between S₁ and S₂ is given, the maximum value of L_m under the high-input voltage range is obtained as

$$L_m\le \frac{n_p{V}_o{t}_d}{16V_{in}{f}_s{n}_s{C}_{oss}}$$

(9)

Mode 4 [t₃ ~ t₄]: At t₃, v_CS2 = 0. Owing to i_Lm > i_Lr and i_Lr(t₃) > 0, D₄ conducts and D_S2 is forward-biased. Therefore, the zero voltage turn-on of S₂ can be achieved. Due to D₄ conducting, one can obtain v_Lm = -n_pV_Co2/n_s = -n_pV_o/(2n_s) and i_Lm decreases in this mode. The resonant frequency in this mode is f_r1.

Mode 5 [t₄ ~ t₅]: At time t₄, i_D4 = 0 and D₄ turns off without the reverse recovery current. The resonant frequency in mode 5 equals f_r2.

Mode 6 [t₅ ~ T_sw+t₀]: At time t₅, S₂ turns off. Since i_Lm < i_Lr and i_Lr(t₅) < 0, D₃ is forward-biased and C_S1 is discharged. The zero voltage turn-on condition of S₁ is expressed as

$$|i_{Lm}(t_5)|=i_{Lm}(t_2)=\frac{n_p{V}_o}{8n_s{L}_m{f}_{sw}}\ge V_{in}\sqrt{2C_{oss}/(L_m+L_r)}$$

(10)

At t₀ + T_sw, v_CS1 = 0.

3. Circuit Analysis and Design Procedure

The presented LLC converter is operated with pulse frequency modulation (PFM). Power switches have a constant duty cycle (d = 0.5) with variable frequency. The switching frequency is dependent on V_in and P_o. The fundamental harmonic frequency approach is used to analyze the circuit characteristics. The variable winding turns controlled by the AC switch S_ac are used on the secondary side to adjust the voltage gain under low- or high-voltage input. Based on the fundamental harmonic frequency analysis, the AC equivalent circuit of the resonant tank is given in Figure 5. R_e is the equivalent primary side resistor of transformer. Since the PWM signals of S₁ and S₂ are square waves with duty cycle d = 0.5, the input voltage of the LLC circuit becomes a square voltage wave measuring 0 V (if S₂ is on) or V_in (if S₁ is on). The fundamental root mean square (rms) input voltage is calculated as $\sqrt{2}{V}_{in}/\pi$. Since S_ac is turned on at low-voltage inputs and off at high-voltage inputs, the secondary winding turns 2n_s (n_s) are connected to the output load under low (high) input voltage ranges. The primary side magnetizing voltage is a square wave with a voltage value of v_Lm = ±n_pV_o/(4n_s) if S_ac is conducting or v_Lm = ±n_pV_o/(2n_s) if S_ac is off. The fundamental rms voltage V_Lm,rms is calculated as $n_p{V}_o/(\sqrt{2}\pi n_s)$ for low-input voltage ranges or $\sqrt{2}{n}_p{V}_o/(\pi n_s)$ for high-input voltage operation. The DC load resistor R_o is reflected to the primary side of the transformer and the AC equivalent resistor R_e can be obtained as $R_e={(n_p/n_s)}^2{R}_o/(2{\pi }^2)$ for low-voltage ranges or $R_e=2{(n_p/n_s)}^2{R}_o/{\pi }^2$ for high-voltage ranges. The voltage transfer function in Figure 5 is calculated as

$$|G|=1/\sqrt{X^2{(\frac{F_n^2-1}{F_n})}^2+{[1+\frac{1}{L_n}\frac{F_n^2-1}{F_n^2}]}^2}=\{\begin{array}{ll}\frac{n_p{V}_o}{2n_s{V}_{in}},& \mathrm{if}{S}_{ac}\mathrm{is}\mathrm{on}\\ \frac{n_p{V}_o}{n_s{V}_{in}},& \mathrm{if}{S}_{ac}\mathrm{is}\mathrm{off}\end{array}$$

(11)

Here, $X=\sqrt{L_r/C_r}/R_e$, F_n = f_sw/f_r and L_n = L_m/L_r. From Equation (11), the output voltage is obtained as

$$V_o==\{\begin{array}{l}\frac{2n_s{V}_{in}}{n_p\sqrt{X^2{(\frac{F_n^2-1}{F_n})}^2+{[1+\frac{1}{L_n}\frac{F_n^2-1}{F_n^2}]}^2}},\mathrm{if}{S}_{ac}\mathrm{is}\mathrm{on}\\ \frac{n_s{V}_{in}}{n_p\sqrt{X^2{(\frac{F_n^2-1}{F_n})}^2+{[1+\frac{1}{L_n}\frac{F_n^2-1}{F_n^2}]}^2}},\mathrm{if}{S}_{ac}\mathrm{is}\mathrm{off}\end{array}$$

(12)

Power switches S₁ and S₂ are controlled with pulse frequency modulation and the LLC converter is operated over wide voltage ranges.

The design procedures of prototypes are provided to confirm the feasibility of the presented LLC converter. The input voltage range V_in is from 100 to 400 V, the output voltage V_o is 48 V, the full power P_o,rated is 500 W, and the series resonant frequency f_r1 is 100 kHz. The inductance ratio L_n is assumed as 7. When V_in = 100–200 V, the proposed converter is operated at low-voltage ranges. If V_in = 200–400 V, then the proposed LLC converter is worked under high-voltage ranges. Since the proposed converter has the same resonant tank under high and low-voltage ranges, the prototype circuit can be designed for either high- or low-input voltage ranges. For high-voltage range operation, the minimum voltage gain of the presented LLC converter under V_in,max = 400 V is assumed at unity. Then, the turns ratio n_p/n_s is expressed in Equation (13) as

$$\frac{n_p}{n_s}=\frac{G{V}_{in,\max }}{V_o}=\frac{1\times 400}{48}\approx 8.33$$

(13)

The magnetic core TDK EE-55 with ΔB = 0.4 T and A_e = 354 mm² is adopted to implement transformer T. It is assumed that the minimum switching frequency is 50 kHz under V_in = 200 V conditions. Therefore, the minimum primary winding turns are calculated as

$$n_{p,\min }\ge \frac{(n_p/n_s)V_o}{4f_{s,\min }\Delta B{A}_e}=\frac{(8.333)\times 48}{4\times 50000\times 0.4\times 354\times {10}^{-6}}\approx 14.123$$

(14)

In the laboratory prototype, we selected n_p = 16 and n_s = 2. Thus, the actual DC gains are obtained in Equations (15) and (16) as

$$G_{\max ,H}=\frac{V_o{n}_p}{V_{in,\min }{n}_s}=\frac{48\times 16}{200\times 2}\approx 1.92$$

(15)

$$G_{\min ,H}=\frac{V_o{n}_p}{V_{in,\max }{n}_s}=\frac{48\times 16}{400\times 2}\approx 0.96$$

(16)

Similarly, the DC voltage gains of the presented resonant converter for low-voltage ranges are obtained as

$$G_{\max ,L}=\frac{V_o{n}_p}{2V_{in,\min }{n}_s}=\frac{48\times 16}{2\times 100\times 2}\approx 1.92$$

(17)

$$G_{\min ,L}=\frac{V_o{n}_p}{2V_{in,\max }{n}_s}=\frac{48\times 16}{2\times 200\times 2}\approx 0.96$$

(18)

From the calculated winding turns n_p and n_s, the equivalent resistance R_e at high-voltage ranges and the rated power are calculated as

$$R_e=\frac{2{(n_p/n_s)}^2{R}_o}{{\pi }^2}=\frac{2\times {(16/2)}^2\times ({48}^2/500)}{{3.14159}^2}\approx 60\Omega$$

(19)

The quality factor X is selected as 0.2. The resonant inductor L_r is calculated as

$$L_r=\frac{x{R}_e}{2\pi f_{r1}}=\frac{0.2\times 60}{2\pi \times 100000}\approx 19\mathsf{\mu }\mathrm{H}$$

(20)

In the proposed converter, the actual resonant inductance L_r is 20 μH. Then, the resonant capacitor is calculated as

$$C_r=\frac{1}{4{\pi }^2{L}_r{f}_{r1}^2}=\frac{1}{4{\pi }^2\times 20\times {10}^{-6}\times {(100000)}^2}\approx 127\mathrm{nF}$$

(21)

Since the selected L_n = L_m/L_r = 7, the magnetizing inductance L_m is obtained as

$$L_m=L_n{L}_r=7\times 20=140\mathsf{\mu }\mathrm{H}$$

(22)

MBR40100PT diodes with a 100 V/40 A rating are used for D₁–D₄. The selected output capacitance C_o1 = C_o2 = 540 μF/100 V. The Schmitt voltage comparator with a reference voltage of 200 V is used to control switch S_ac. Switch S_ac is implemented by two IRF1405ZPBF metal-oxide-semiconductor field-effect transistors (MOSFETs) with a 55 V/118 A rating. The voltage stress of S₁ and S₂ is V_in,max = 400 V. In the prototype, STW48N60M2 MOSFETs with a 600 V/26 A rating are used for S₁ and S₂. The UCC25600 PWM integrated circuit is adopted to realize pulse frequency modulation. The voltage TL431 regulator and the PC 817 optocoupler are adopted on the low-voltage side to achieve load voltage regulation and signal isolation, respectively. Figure 6 provides the circuit diagram of the prototype with the adopted control scheme.

4. Experimental Results

The measured waveforms of the prototype circuit are given to confirm the effectiveness of the presented LLC resonant converter with wide voltage operation. The experimental waveforms of v_S1,g, v_S2,g, v_Cr, and i_Lr for different input voltages 100 V, 190 V, 210 V, and 400 V are given in Figure 7. For V_in = 100 V and 190 V, the presented converter is operated at low-voltage ranges. The S_ac conducts and 2n_s winding turns are connected to the output load. The converter operated at V_in = 100 V has high-voltage gain compared to the converter operated at V_in = 190 V. One can observe that the switching frequency of the converter at V_in = 190 V is greater than the switching frequency at V_in = 100 V. For V_in = 210 V and 400 V conditions, the presented LLC converter is operated at high-voltage ranges. The S_ac is turned off and n_s instead of 2n_s winding turns are connected to the output load. Similarly, the proposed converter operated at V_in = 210 V has a lower switching frequency compared to the converter operated at V_in = 400 V. The experimental waveforms of i_D1–i_D4 for different input voltage conditions and the full power are given in Figure 8. For V_in = 100 V and 190 V conditions, the switch S_ac is turned on and D₃ and D₄ are reverse-biased. Therefore, i_D3 = i_D4 = 0, as shown in Figure 8a,b. On the other hand, S_ac is turned off, while D₁ and D₂ are also off for V_in = 210 V and 400 V conditions. Therefore, i_D1 = i_D2 = 0, as shown in Figure 8c,d. Diodes D₁ and D₂ turn off without the reverse recovery current at V_in = 100 V (Figure 8a) because f_sw at V_in = 100 V is about 45 kHz < f_r1 = 100 kHz. Similarly, D₃ and D₄ also turn off without the reverse recovery current at V_in = 210 V (Figure 8c) because f_sw is about 50 kHz < f_r1 = 100 kHz. The experimental waveforms of switch S₁ at 20% power and different input voltage conditions are given in Figure 9. The experimental waveforms of S₁ at 100% power and different input voltage conditions are illustrated in Figure 10. From the experimental waveforms in Figure 9; Figure 10, it is clear that S₁ is turned when zero voltage switches from 20% power to 100% power and over the entire voltage range.

The measured input voltage and the PWM signal of the AC switch S_ac are provided in Figure 11. The hysteresis voltage band and reference voltage of the voltage comparator are 10 V and 200 V, respectively. First, the input voltage is increased from 100 V to 400 V. When V_in is greater than 205 V, the PWM signal of S_ac is changed from the “on” state (high-voltage) to “off” state (low-voltage). Second, the input voltage is decreased from 400 V. When V_in is less than 195 V, S_ac is turned on (high-voltage). Figure 12 gives the experimental switching frequency of the proposed resonant converter. For low-voltage ranges and the same load condition, the voltage gain of the resonant converter under V_in = 100 V is higher than V_in = 190 V. One can see that the measured switching frequency at 100 V input voltage is lower than 190 V input voltage. The experimental circuit efficiencies of the proposed converter at 100% load are 87.3% (100 V input), 91.2% (190 V input), 89.5% (210 V input), and 92.8% (400 V input). Since the turns ratio n_p/n_s is used under high-voltage ranges (V_in = 200–400 V) instead of n_p/(2n_s) under low-voltage ranges (V_in = 100–200 V), the converter has a lower magnetizing current under high-voltage ranges compared to the low-voltage range operation. The converter has lower conduction loss on power switches at high-input voltage ranges. Therefore, the experimental circuit efficiency at 400 V input is better than the 190 V input, and the efficiency with the 210 V input is better than the 100 V input.

5. Conclusions

The half-bridge LLC resonant converter with variable winding turns is studied and implemented in this paper to accomplish two objectives: (1) wide, soft switching ranges over the entire input voltage and output road; and (2) wide voltage operation. The general half-bridge resonant converter is adopted on the high-voltage side to a achieve series resonant circuit and realize the soft switching operation for powered semiconductors. Two sets of winding turns are adopted on the low-voltage side to accomplish wide voltage operation capability. Based on the circuit analysis and test results, the proposed LLC converter can achieve zero voltage switching with wide input voltage (100 V to 400 V) and wide load (20% to 100% rated power) ranges. Compared to the conventional LLC resonant converters with wide voltage operation, the proposed converter has lower component counts, a simple control scheme, and less voltage stress on rectifier diodes. The studied LLC converter can be used for power units in computers and power servers with large hold-up times, PV power converters, fuel cell converters, and battery chargers and dischargers with wide voltage operation. Experimental waveforms from a 500 W prototype are given and presented to confirm the usefulness of the presented circuit topology. The future works will investigate the new wide voltage converters with low cost and high power density advantages for industry power applications and renewable energy power conversion.