Power Compression and Phase Analysis of GaN HEMT for Microwave Receiver Protection

Abstract

This paper reports a high-performance microwave receiver protector (RP) based on a single gallium nitride (GaN) high electron mobility transistor (HEMT) at an operation frequency of 30 to 3000 MHz. The HEMT-based RP exhibits multi features: high power compression, constant output power, tunable threshold power level, and insensitivity to frequency variation. With a low drain voltage (V_ds) of 3 V, constant output power of 9.9 dBm was acquired for input power over its threshold power of 3.2 dBm. Power compression of 13.3 dB was achieved at the input power of P_in = 20 dBm. In addition, adjustable threshold power level P_th could be obtained by merely tuning drain voltage. Transducer gain measurement results were employed to explain the occurrence of output power saturation. Relatively higher P_th was linked to wider gate voltage swing which extended the linear region of the P_out-P_in characteristic. In addition, the GaN HEMT’s power compression capability shows great immunity to frequency variation, which is promising for protecting sensitive receiver components at both low and high frequencies. Finally, the phase shift of the GaN HEMT RP at high input power was measured and analyzed by the nonlinear behaviors of input capacitance C_gs.

1. Introduction

Receiver protectors (RPs) are widely used to provide protection to radio frequency (RF) and microwave receivers and components, such as low noise amplifiers (LNAs) [1,2] and analog-to-digital converters (ADCs) [3,4]. An RP allows input power below a certain value to pass through ideally without loss, and attenuating input signal strength when it exceeds the threshold. A number of device technologies have been developed to achieve RF and microwave RPs, including Schottky barrier diodes (SBDs) [5,6,7], p-i-n diodes [3,8,9], and transistors [10,11,12]. Employing a steep-mesa technology, a gallium nitride (GaN) SBD based RP demonstrated a low on-resistance (R_ON) and a power compression of 3.3 dB with a corresponding input power of 20 dBm at 2 GHz [7]. A low insertion loss of 0.3 dB has also been reported for a diamond diode based RP at an operation frequency of 1 GHz [9]. However, for a diode-based RP, typically a tradeoff often has to be made between C_off and R_on [7,8,9]. Although diode-based RPs have made steady progress, they still face some important challenges, such as insufficient power compression, inadequacy in tuning their threshold power level (the input power with 1-dB gain compression), and sensitivity to frequency change. Some attempts, including reverse-biased configurations, stacked antiparallel configurations, etc. have been enacted to mitigate the issues [3,13]; however, most of the efforts required complex circuits to work with. Transistors have been regarded as another viable option for RF RPs. The basic principle of transistor-based RPs is to operate the transistors in their saturation region to limit the output power [10,14,15]. Owing to their saturated output power, RPs formed by transistors possess the advantages of constant output power and outstanding power compression capability.

AlGaN/GaN-based high electron mobility transistors (HEMTs) with embedded 2-dimension electron gas (2DEG) have been widely used in power amplifiers (PAs) [16,17], RF switches [18,19] and LNAs with extremely short recovery time [20,21], due to superior properties such as high power density at high frequencies, high breakdown voltages and large carrier density and mobility [22,23,24].

In this paper, we demonstrated a microwave RP based on a single GaN HEMT on Silicon carbide (SiC) substrate. The HEMT receiver protector offers a certain gain for the small input signal and attenuates input signal strength beyond the threshold power into a constant output power. The threshold power could be well adjusted by varying the bias. It is also demonstrated that the power reduction capability was hardly sensitive to working frequency. Moreover, the phase shift—a critical issue when an RP was applied in high performance phased-array receiver front end [25]—was investigated and analyzed by the small RF signal measurements. The results paved a solid path for a single GaN HEMT to form an RP with high power compression, threshold power adjustability, and frequency stability.

2. Materials and Methods

The GaN-based HEMT used in this study was grown and fabricated on a SiC substrate as shown in Figure 1a. The epitaxial structure includes an Al_0.25Ga_0.75N barrier layer, a GaN layer buffer layer, and an AlN nucleation layer. The fabrication process of the HEMT started with mesa isolation, which was completed in Inductive Coupled Plasma Reactive Ion Etching (ICP-RIE) and wet etching in 5% TMMA at 50 °C. Then, the ohmic contact was formed by depositing Ti/Al/Ni/Au (20/150/50/80 nm), followed by annealing at 850 °C for 45 s. Finally, a Ni/Au (50/150 nm) Schottky metal was evaporated as the gate and interconnection. The gate terminal was fabricated into a rectangular shape with its length and width of 1.2 µm and 100 µm, respectively, and the distances between gate to source and to drain were 1.5 μm and 2 μm (L_g = 1.2 μm, L_gs = 1.5 μm, L_gd = 2 μm, W_g = 100 μm).

The setup shown in Figure 1b was used for power compression measurements of the GaN HEMT RP. To investigate the power limiting performance of the GaN HEMT at various frequencies, an on-wafer GaN HEMT measurement was conducted without being constrained by the matching network. The input RF signal which was generated by the signal generator (Rohde & Schwarz SMA100B) was injected into the gate using a bias-tee. The output RF signal was extracted from the drain terminal using a spectrum analyzer. To protect the spectrum analyzer from being overdriven, a 30 dB attenuator was used. Power-dependent measurements at various DC bias voltages and different frequencies were carried out with the aid of a computer based controller. The phase shift derived from the reference value at P_in = −20 dBm was measured by a Vector Network Analyzer (VNA) (Keysight ENA5080A).

3. Results and Discussion

3.1. DC Characteristic

Figure 2a shows the transfer characteristics of the GaN HEMT, with a peak DC transconductance (g_m) of 197 mS/mm obtained at V_gs/V_ds = −1/3 V. The threshold voltage V_th was determined to be −1.8 V, extracted from the linear extrapolation of the I_ds-V_gs curve. Figure 2b depicts the output characteristics of the GaN HEMT, showing saturated drain current density (I_dsat) of 667 mA/mm at V_gs = 4 V. The slight drop of the drain current during relatively high drain voltage with V_gs >1 V is due to the self-heating effect (SHE) [26]. Under the field strength of 0.5 MV/cm, the gate leakage current density of this GaN HEMT was measured to be 1.95 × 10⁻³ mA/mm, which is much smaller than the typical gate breakdown standard of 1 mA/mm [27,28], indicating a V_gd breakdown voltage higher than 100 V.

3.2. High Power Compression of the GaN HEMT RP

Figure 3a presents representative output power (P_out) as a function of input power. When the input power was low, the receiver protector offered a fixed gain of 2.9 dB. The results show that P_out increased linearly as injected power was enhanced from −20 dBm to 3.2 dBm, with a fixed gain of 2.9 dB rather than insertion loss typically observed in a diode-based RP. With an input power of 3.2 dBm, a gain reduction of 1 dB was observed. When the input power was further enhanced, the output power began to saturate and was clamped at 9.9 dBm as shown in Figure 3a. The input power level could only reach 21 dBm due to the short length of the L_gd (2 microns), and it could be further improved by some advanced processing technologies, such as a metal-insulator-semiconductor (MIS) HEMT with a gate dielectric [29] or structures with surface passivation [30].

In the shaded area of Figure 3b, the power compression as a function of input power could be well fitted by a linear relationship with a slope of 1 dB per 1dB, which indicated a constant output power. The improvement of the power compression of this GaN HEMT RP was at the expense of DC power consumption.

The GaN HEMT RP in this work presented a better power limiting capability than diode-based RPs reported in the literature [7,31,32]. As shown in Figure 3c, when the injection power was 17 dB higher than individual threshold power levels, the GaN HEMT RP had 2~4 dB more power compression than those diode-base RPs. It was attributed to the saturated output power of the GaN HEMT, while the diode-based RP cannot sufficiently attenuate the input power due to its non-zero on-resistance (R_on) at high input power [33]. Moreover, there was a tradeoff between high power compression and circuit complexity for a diode based RP [34]. The high power compression, as shown in Figure 3b, was challenging to realize for a single-stage diode based RP [3,33]. Therefore, increasing the number of stages was often used to achieve a constant output power at a large input signal, which would increase the size and complexity of the diode based RP circuit. Whereas, for the GaN HEMT-based RP, the circuit complexity only came from the bias circuit.

3.3. Tunable Threshold of the GaN HEMT RP

Figure 4a shows the relationship between the output power and the input power of the GaN HEMT RP at V_gs = −1 V (working frequency of 3 GHz), with three different drain voltages (3/5/10 V). When the P_in was below 3.5 dBm, uniform output power P_out was acquired for all the bias conditions. However, when the P_in was beyond 7 dBm, apparent divergence was spotted as V_ds increased from 3 V to 5 V and 10 V. In addition to enhanced output power with increasing V_ds, the threshold power level P_th was also enhanced. Threshold power levels P_th of 3.97/6.97/7.97 dBm were extracted from V_ds = 3/5/10 V, respectively, indicating that a higher drain bias may extend the linear region of an RP and postpone the occurrence of power saturation.

To investigate the dependence of the threshold power level P_th on the drain voltage V_ds, the transducer gain, which was measured at a low input power of −20 dBm, was utilized. Figure 4b depicts transducer gain as a function of V_gs at the frequency of 3 GHz. The gate voltage swing (GVS) [35] for a 9 dB transducer gain drop was determined to be 2.9 V, 4.4 V, and 5.2 V, for V_ds = 3/5/10 V, respectively.

A relative wider GVS means the transistor could keep the transducer gain high for an ample range of input power, as the input power was applied at the gate-source terminals. Thus, with a wide GVS, the transistor could allow a broad range of input power through without causing output power to be compressed, as shown in an extended linear region of the P_out-P_in graph [Figure 4a]. As a result, a larger threshold power level P_th could be obtained by merely increasing drain-source bias, and P_th can be easily electronically-tuned by V_ds, which contributes to the application of the HEMT based RP in a dynamic environment and the precise protection of sensitive devices [13,36]. It should also be noted that reducing the L_g of a GaN HEMT would enhance the non-uniformity of g_m, which could enable the lowering of P_th to protect more power-sensitive devices [37,38]. However, for a p-i-n diode RP, the threshold power level was typically governed by the thickness of the i-layer, and can hardly be adjusted once the device was manufactured [39].

3.4. Frequency Independent HEMT RP

Figure 5a reports the input and output power characteristics of the GaN HEMT based RP at various working frequencies. The measured data show that at a given V_ds, the GaN HEMT based RP exhibited typically the same power compression capability and threshold power level over a wide frequency range from 30 MHz to 3 GHz. For example, at a V_ds of 3 V, the maximum output power and P_th were extracted to be 9.86 $\pm$0.3 dBm and 2.81 $\pm$ 0.4 dBm, respectively, showing little divergence among various operation frequencies from 30 MHz to 3 GHz. Whereas, P_th of a p-i-n based RP dropped rapidly as shifting to low operation frequencies due to the transit time of carriers across the i-region is shorter than the signal period time at a low frequency, e.g., P_th was reduced from 20 dBm at 3 GHz to 9 dBm at 30 MHz for a silicon p-i-n diode [31].

Unlike p-i-n based RPs, this GaN HEMT based RP can maintain a good frequency response within its cut-off frequency (f_t) of 11.28 GHz. The frequency-insensitive property could be attributed to the uniform transducer gain of the device over a wide operation frequency, as shown in Figure 5b, which shows the transducer gain as a function of frequency for this GaN HEMT. In the range of 30–3000 MHz, an almost steady transducer gain was obtained for either a small RF signal (P_in = −20 dBm) or relatively large RF power (P_in = 5 dBm). As a result, almost identical output power could be expected in the linear region and saturation region of the device. Therefore, constant output power could be achieved at both small and large input power, resulting in a good broadband property without matching circuits. It should be noted that adding a matching network in a real application will not affect the characteristics of the GaN HEMT RP, such as high power compression and tunable threshold power.

The frequency response of this HEMT RP is related to the current cut-off frequency (f_t) and maximum oscillation frequency (f_max). It is noted that the device used in this study featured a gate length of 1.2 μm and held an f_t of 11.28 GHz and an f_max of 22.35 GHz, respectively. It is expected that the frequency steadiness property could be extended to an even higher frequency on shrinking the gate length dimension [40,41,42]. Based on its frequency-insensitive property, GaN HEMT could be used to protect sensitive components such as mixers in transmitters over a large frequency range. In addition, it paved a path to improving the frequency response without using multiple devices or auxiliary circuits [3].

3.5. Phase Shift Analysis of the HEMT RP

Figure 6a illustrates the phase shift of GaN HEMT based RP with different V_ds. The output phase was typically unchanged when the device was fed with a small input power lower than 0 dBm. However, as P_in increased, the phase dropped dramatically. Phase shift over 1 degree occurred at the P_in of 6.2/10.7/12.7 dBm for V_ds = 3/5/10 V respectively.

A small RF signal equivalent circuit model of this GaN HEMT is shown in Figure 6b [43,44]. This model consisted of an input resistance (R_i), an input (C_gs), output (C_ds) and feedback (C_gd) capacitors, and a voltage controlled current source (g_m·V_gsiexp(-jωτ)) where τ was transconductance delay. The RF power source is modeled by a voltage source and an internal impedance Z_s = R_s. The output load is modeled by an admittance Y_L = G_L.

The feedback capacitor C_gd could be ignored due to the fact that it had small value. Therefore, the phase of output voltage could be computed as Equation (1):

$$\angle {\mathrm{V}}_{\mathrm{out}}=\arctan (-{\mathsf{\omega }\mathrm{C}}_{\mathrm{gs}}\left({\mathrm{R}}_{\mathrm{s}}{+\mathrm{R}}_{\mathrm{i}}\right))+\arctan \left(-\frac{{\mathsf{\omega }\mathrm{C}}_{\mathrm{ds}}}{{\mathrm{g}}_{\mathrm{ds}}{+\mathrm{G}}_{\mathrm{L}}}\right)-\mathsf{\omega }\mathsf{\tau }.$$

(1)

Since C_ds (at fF level) is much smaller than the input capacitor C_gs, the phase of output voltage could be further simplified as Equation (2):

$$\angle {\mathrm{V}}_{\mathrm{out}}=\arctan (-{\mathsf{\omega }\mathrm{C}}_{\mathrm{gs}}\left({\mathrm{R}}_{\mathrm{s}}{+\mathrm{R}}_{\mathrm{i}}\right))-\mathsf{\omega }\mathsf{\tau }.$$

(2)

Thus, the phase of output voltage could be regarded to be merely determined by the C_gs. C_gs could be extracted by the small RF signal equivalent circuit model shown in the dashed box in Figure 6b [44].

Figure 6c shows C_gs as a function of V_gs. For a given V_ds, C_gs was unchanged around V_gs = −1 V, which corresponds to a relatively low input power. However, on increasing the input power, which could be interpreted as increasing the V_gs, a dramatically enhanced C_gs was observed. The increase of C_gs would cause a decrease in the phase of output voltage according to Equation (2), resulting in a significant drop in the phase shift at large inputs as shown in Figure 6a.

When V_ds was increased from 3 V to 5 V and 10 V, the corresponding transition voltages extracted from the linear extrapolation of C_gs-V_gs curve were also increased from 1.2 V to 2.6 V and 6.2 V, respectively. The lower transition voltage at a relatively smaller drain voltage leads to the early onset of C_gs increasing, which causes the early significant output phase drop at a relatively small V_ds when a large power is injected into the gate of the device.

4. Conclusions

High performance microwave RP based on a single GaN HEMT is demonstrated, with features of high power compression, tunable threshold power level, and a frequency-insensitive property. Although additional power was dissipated, constant output power was achieved at large input power owing to the saturated output power of the HEMT RP, indicating a competitive advantage over diode-based RPs. In addition, adjustable P_th was presented and P_th could be easily tuned by adjusting drain voltage. The tunable P_th was well modeled and explained by transducer gain measurement results. The higher P_th was attributed to the wider gate voltage swing, which extended the linear region of the P_out-P_in characteristics. Both P_th and the power compressions are frequency-independent, indicating a good broadband property without matching circuits, which means the GaN HEMT RP could be used to protect ADCs and sensitive receiver components at both low and high frequencies. The phase shift of the GaN HEMT at high input power was measured and analyzed. The occurrence of the phase shift was mainly caused by the nonlinearity of the input capacitor C_gs at a large signal input. It was found that a higher drain voltage would hinder the onset of the phase shift and reduce the extent of the phase shift.