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*Energies*
**2017**,
*10*(4),
434;
doi:10.3390/en10040434

Article

A High-Efficiency Isolated-Type Three-Port Bidirectional DC/DC Converter for Photovoltaic Systems

Department of Electronic Engineering, National Kaohsiung First University of Science and Technology, Kaohsiung 824, Taiwan

*

Correspondence: Tel.: +886-7-6011000 (ext. 2511)

Academic Editor:
Andrés G. Muñoz

Received: 3 February 2017 / Accepted: 22 March 2017 / Published: 25 March 2017

## Abstract

**:**

This paper proposes a novel high-efficiency isolated three-port bidirectional DC/DC device for photovoltaic (PV) systems. The device contains a high step-up converter for PV modules to supply power to the DC bus, and a bidirectional charge/discharge control circuit for the battery with an improved boost-flyback converter. When the PV modules supply sufficient energy, their output can be stepped up and energy supply to the DC bus and charging of the battery can be achieved simultaneously. However, when the energy supplied is insufficient, the battery provides energy to the DC bus. When the proposed converter is operated in the step-down mode, the DC-blocking capacitor on the high-voltage side is used to reduce the voltage on the transformer and achieve high step-down performance. Moreover, to improve the overall efficiency of the system, the energy stored in the leakage inductance is recycled and supplied to the DC-blocking capacitor during operation in the step-up mode. Finally, to verify the feasibility and practicability of the proposed devices, a 500 W three-port bidirectional DC/DC devices was implemented. The highest efficiencies achieved for operation in different modes were as follows: high step-up mode for the PV modules, 95.2%; battery step-up mode, 94.2%; and step-down mode, 97.6%.

Keywords:

isolated type; three-port bidirectional converter; photovoltaic systems## 1. Introduction

In recent years, greenhouse gas emissions have increased with advances in technology, leading to global warming and climate change. The energy technology industry continues to be heavily dependent on fossil fuels and is unable to handle this stringent environment [1,2]. Therefore, the use of green energy has become the core of major economic strategies and is a key focus in world politics, especially because reducing greenhouse gas emissions and conserving energy are the main global concerns at present.

In general, because of the large difference between the output voltage of green energy apparatus and the operating voltage of DC bus, a high step-up converter is always required for connecting these two stages. A conventional boost converter should operate with an extremely high duty ratio to provide high step-up voltage. In particular, its maximum step-up ratio is limited by the parasitic elements of the circuit components. The existence of parasitic components [3] and the reverse recovery time of diode [4] reduce the converter efficiency. Consequently, the use of conventional boost converters in the field of green energy is limited. Recently, to achieve a high step-up ratio, the use of a series boost converter, the voltage-lift technique, a coupled inductor, and a cascade circuit structure has been proposed [5,6,7,8,9,10,11]. When the switch of the boost converter is open, a large current flows through the power components, and therefore, power components with high current stress are required, which implies high cost and conduction energy loss. The conventional flyback converter and forward converter have also been proposed for achieving high step-up voltages. However, because the leakage inductance of the transformer in these ones can lead to high reverse voltage across the switch, a passive snubber circuit or an active clamp is required to suppress such a reverse voltage. A passive snubber circuit has a resistance that depletes the excess energy of the leakage inductance, thereby reducing the circuit efficiency. By contrast, an active clamp results in many components being added to the circuit, which not only increases cost but also renders inverter control difficult. In [12], a topology was proposed for solving the leakage inductance problem; however, the remaining disadvantages—large circuit size and high cost—persist.

An application block diagram of a typical green energy system is shown in Figure 1. Achieving low cost, a small circuit size, and high efficiency will always be R&D objectives. Furthermore, energy storage systems are a major research subject related to the use of green energy. In the green energy system, because of the large difference between the battery voltage and the DC bus voltage, a high step-up converter is required to transfer energy from the battery to the DC bus. Recently, many studies have conducted research on bidirectional converters [13,14,15,16,17,18,19,20,21,22,23,24,25]. They have attempted to develop a topology that can provide both step-up and step-down operations; which has the potential to reduce the number of components, cost, and size of the system. The commonly used non-isolated bidirectional devices, including the buck/boost bidirectional devices [13], bidirectional Cuk devices [14], and bidirectional single-ended primary inductance devices (SEPIC) [15], have few components, low cost, and high controllability; moreover, they are easy to design. However, all of them have low efficiency and can become unstable when operating at a high voltage conversion rate [3]. Furthermore, these non-isolated circuits are susceptible to interference and cannot be used to achieve electricity isolation, factors that considerably limit their use. While the commonly used isolated bidirectional converters are full-bridge bidirectional converters [17,26,27], half-bridge bidirectional converters [19], and bidirectional flyback converters [18]. Both full-bridge and half-bridge bidirectional converters can perform soft switching by controlling the phase shift, resulting in high efficiency, but their transformer design is such that achieving inductance–inductance–capacitance (IIC) resonance is difficult. Furthermore, bidirectional flyback converters have the same advantages as non-isolated bidirectional converters: few components, low cost, and ease of design. When its switch is OFF, the leakage inductor of the transformer and the parasitic capacitor in the switch produce resonance, resulting in a voltage spike at the switch; therefore, a switch with high voltage stress is required. The aforementioned isolated bidirectional converters always have a large transformer because a high turns ratio is necessary for operation at a high voltage conversion rate. Recently, many modified bidirectional converters have been proposed. For example, Reference [21] shows a bidirectional converter that can recycle the energy of the leakage inductor of the transformer and perform synchronous rectification; thus, the converter shows high efficiency while meeting the requirement of electricity isolation. However, the converter topology involves two inductors, and the transformer is considered an ideal transformer despite having a low turns ratio; moreover, the energy release problem associated with the magnetizing inductor leads to low efficiency for high-power operations. Reference [22] proposed that the battery, which is the auxiliary power source, should be charged and used for providing energy to the DC bus during circuit operation in the step-up mode. The leakage inductance of the coupled inductor can recycle the energy and release it to the battery auxiliary power circuit and DC bus on the high-voltage side. However, the circuit control procedure is complicated, and the battery auxiliary power cannot supply energy to the DC bus.

Although a bidirectional converter provides both step-up and step-down voltages, when it is used in a photovoltaic (PV) system, energy generated by the PV system is required to pass through two converter stages for charging the battery, resulting in a reduction in the overall efficiency of the system. To overcome this problem, a three-port DC/DC converter topology has been proposed [23,24,25,26,27,28,29,30,31,32,33,34] for the PV system. When the energy supplied by the PV modules is sufficient, supply of energy to the DC bus and charging of the battery are simultaneously performed. However, in the absence of energy supply from the PV modules, the battery can provide energy to the DC bus. The three-port DC/DC converter can operate in different modes, thereby reducing the cost and size of the system [26,27,28,29,30,31,32,33]. However, it has a larger size, higher cost, and lower efficiency compared with the converter proposed in the current study. In general, a three-port DC/DC converter is not involved in the charging of the battery by the DC bus. This implies that the converter lacks the step-down mode, similar to bidirectional converters. If a topology could combine the advantages of both bidirectional converters and the three-port converter, the size and cost of the system could be further reduced.

For the reasons mentioned above, in this paper we propose a novel isolated three-port bidirectional DC/DC converter whose block diagram is shown in Figure 2. When the energy supplied by the PV modules is sufficient, the converter can not only step up the output of the PV modules, but also provide energy to the DC bus and charge the battery simultaneously. However, when the energy supplied is insufficient, the converter can step up the output of the battery and provide energy to the DC bus. Any excess energy in the DC bus is used to charge the battery, implying that the proposed topology can perform both high step-up and high step-down functions. The topology proposed is isolated, and the DC -blocking capacitor on the high-voltage side can reduce the voltage on the transformer, facilitating the operation of the system in the high step-down mode. An improved boost-flyback converter is used on the low-voltage side of the PV modules to step up the output of the PV modules for charging the battery and providing energy to the DC bus. The auxiliary inductor on the battery side can modify the output step-up voltage and provide a continuous current for battery charging, which can extend the battery lifetime. Additional switches and diodes are added to prevent the energy generated by the PV system from directly flowing to the battery and to control the step-up process for providing energy to the DC bus in the absence of energy supply from the PV system. The proposed topology consists of three operational modes, which reduce the cost and size of the system considerably. Furthermore, the converter can perform synchronous rectification and recycle energy from the leakage inductor simultaneously, and these functions increase the overall efficiency of the system considerably.

## 2. Operating Principles of the Main Circuit

The detailed scheme of our proposed device is shown in Figure 3 and briefly discussed in this section. The DC-blocking capacitor Cm on the high-voltage side facilitates the operation of the system in the high step-down mode by reducing the voltage on the transformer. It can also recycle the energy of the leakage inductor on the secondary side of the transformer and increase the efficiency of the system. An improved boost-flyback converter is used on the low-voltage side of the PV modules to step up the output of the PV modules for charging the battery and providing energy to the DC bus. The auxiliary inductor Lb on the battery side can modify the stepped-up output voltage for providing a continuous current for charging the battery. Switch S2 and diode D2 not only prevent the energy generated by the PV system from directly charging the battery, but also control the step-up process for providing energy to the DC bus and charging the battery when the energy supplied by the PV modules is sufficient. In the absence of energy supply from the PV modules, the battery output is stepped up and energy is provided to the DC bus. Any excess energy in the DC bus is used to charge the battery.

The proposed topology was studied and analyzed by considering three operation stages. In Stage 1, the output of PV modules is stepped up for charging the battery and providing energy to the DC bus when the energy supplied by the PV modules is sufficient. In Stage 2, the output of the battery is stepped up to provide energy to the DC bus when the energy supplied by the PV modules is insufficient. In Stage 3, the output of the DC bus is stepped down and the battery is charged when there is excess energy in the DC bus. To simplify the analysis of the converter, the following assumptions were made:

- (1)
- Capacitors Ci and Co have very high capacitance and can be considered as current sources.
- (2)
- Switches S1, S2, S3, and S4 and diodes D1 and D2 are considered as ideal components.
- (3)
- The magnetizing inductances of the transformer are larger than the leakage inductances of the transformer.

#### 2.1. Stage 1: Stepping up the PV Output for Providing Energy to The DC Bus and Charging the Battery

#### 2.2. Stage 2: Stepping up the Battery Output

#### 2.3. Stage 3: Stepping Down the Output of the DC Bus

## 3. Steady-State Analysis

This section focuses on the mathematical derivation of the voltage conversion ratio and the voltage stress of components. The following assumptions were made to simplify the analysis:

- (1)
- Capacitors Ci and Co have very high capacitance and can be considered as current sources;
- (2)
- Switches S1, S2, S3, and S4 and diodes D1 and D2 are considered as ideal components;
- (3)
- The leakage inductances of the transformer are smaller than the magnetizing inductances, and therefore, they can be neglected;
- (4)
- The system operates in the continuous conduction mode.

#### 3.1. Voltage Conversion Ratio

When the energy supplied by the PV modules is sufficient, switch S1 is turned ON and the voltage difference across the magnetizing inductor is equal to the input voltage of the PV modules. In the absence of energy supplied by the PV modules, switches S1 and S2 are turned ON and the voltage across the magnetizing inductor is equal to the battery voltage. Because the input voltage of the PV modules and the battery voltage are almost the same in a three-port converter, both step-up from the PV modules to the DC bus and that from the battery to the DC bus have the same step-up ratio. To simplify the derivation of the voltage conversion ratio, first, we neglect part of the battery circuit.

(1) Step-up voltage conversion ratio G

_{Vo}When switch S1 is turned on:
where it is assumed that N = $\frac{{\mathrm{N}}_{2}}{{\mathrm{N}}_{1}}$ and the total turn-on time T

$${\mathrm{V}}_{\mathrm{Lm}}=\mathrm{Lm}\frac{{\mathrm{d}}_{\mathrm{iL}}}{{\mathrm{d}}_{\mathrm{t}}}={\mathrm{V}}_{\mathrm{i}}$$

$$\Delta {\mathrm{i}}_{\mathrm{Lm}}{}^{+}=\frac{{\mathrm{V}}_{\mathrm{i}}}{{\mathrm{L}}_{\mathrm{m}}}\mathrm{DTs}$$

$${\mathrm{V}}_{\mathrm{Cm}}={\mathrm{V}}_{\mathrm{i}}\text{}\frac{{\mathrm{N}}_{2}}{{\mathrm{N}}_{1}}$$

_{on}= DTs.When switch S1 is turned OFF:

$${\mathrm{V}}_{\mathrm{Lm}}=\mathrm{Lm}\frac{{\mathrm{d}}_{\mathrm{iL}}}{{\mathrm{d}}_{\mathrm{t}}}=(\mathrm{Vo}-{\mathrm{V}}_{\mathrm{Cm}})\frac{{\mathrm{N}}_{2}}{{\mathrm{N}}_{1}}$$

Substituting (3) into (4) gives:

$${\mathrm{V}}_{\mathrm{Lm}}=\mathrm{Lm}\frac{{\mathrm{d}}_{\mathrm{iL}}}{{\mathrm{d}}_{\mathrm{t}}}=\mathrm{Vo}\frac{{\mathrm{N}}_{1}}{{\mathrm{N}}_{2}}-{\mathrm{V}}_{\mathrm{i}}$$

$$\Delta {\mathrm{i}}_{\mathrm{L}}{}^{-}=\frac{{\mathrm{V}}_{\mathrm{O}}\text{}\frac{{\mathrm{N}}_{1}}{{\mathrm{N}}_{2}}-\mathrm{Vi}}{{\mathrm{L}}_{\mathrm{m}}}(1-\mathrm{D})\mathrm{Ts}$$

The total turn-off time T

_{off}is given by (1 − D)Ts.According to the principle of voltage-second balance:

$$\Delta {\mathrm{i}}_{\mathrm{L}}{}^{+}=\Delta {\mathrm{i}}_{\mathrm{L}}{}^{-}$$

$${\mathrm{G}}_{\mathrm{Vo}}=\frac{{\mathrm{V}}_{\mathrm{O}}}{{\mathrm{V}}_{\mathrm{i}}}=\frac{{\mathrm{N}}_{2}}{{\mathrm{N}}_{1}}\frac{1}{1-\mathrm{D}}=\frac{\mathrm{N}}{1-\mathrm{D}}$$

Assuming that all the energy of the magnetizing inductors is transferred to the battery and that the maximum current is equal to the current of the magnetizing inductors, we can design the battery charging current:

$${\mathrm{i}}_{\mathrm{Lb}\left(\mathrm{max}\right)}=\frac{{\mathrm{V}}_{\mathrm{i}}}{{\mathrm{L}}_{\mathrm{m}}}\mathrm{DTs}=\frac{{\mathrm{V}}_{\mathrm{b}}}{{\mathrm{L}}_{\mathrm{b}}}\mathrm{DTs}$$

(2) Step-down voltage conversion ratio G

_{Vb}For deriving the step-down voltage conversion ratio of the DC bus, which charges the battery, switch S2 and diode D2 should first be turned OFF, and subsequently, switch S4 is turned ON and switch S3 is turned OFF:

V

_{Lm}= (Vo − V_{Cm})
$${\mathrm{V}}_{1}=-{\mathrm{V}}_{\mathrm{Lm}}\frac{{\mathrm{N}}_{1}}{{\mathrm{N}}_{2}}=-(\mathrm{Vo}-{\mathrm{V}}_{\mathrm{Cm}})\frac{{\mathrm{N}}_{1}}{{\mathrm{N}}_{2}}$$

$${\mathrm{V}}_{\mathrm{Lb}}={\mathrm{V}}_{\mathrm{Ci}}-{\mathrm{V}}_{1}-{\mathrm{V}}_{\mathrm{B}}=\mathrm{Lb}\frac{{\mathrm{d}}_{\mathrm{iL}}}{{\mathrm{d}}_{\mathrm{t}}}$$

$$\Delta {\mathrm{i}}_{\mathrm{Lm}}{}^{+}=\frac{{\mathrm{V}}_{\mathrm{Lb}}}{{\mathrm{L}}_{\mathrm{b}}}\mathrm{DTs}$$

When switch S4 is turned OFF and switch S3 is turned ON, we have:

$${\mathrm{V}}_{1}={\mathrm{V}}_{\mathrm{Lm}}\frac{{\mathrm{N}}_{1}}{{\mathrm{N}}_{2}}={\mathrm{V}}_{\mathrm{Ci}}.$$

Substituting (11) into (14) gives:

$${\mathrm{V}}_{\mathrm{Lm}}=\frac{{\mathrm{V}}_{\mathrm{O}}}{2}$$

$${\mathrm{V}}_{\mathrm{Lb}}={\mathrm{V}}_{\mathrm{B}}=\mathrm{Lb}\frac{{\mathrm{d}}_{\mathrm{iL}}}{{\mathrm{d}}_{\mathrm{t}}}$$

$$\Delta {\mathrm{i}}_{\mathrm{L}}{}^{-}=\frac{{\mathrm{V}}_{\mathrm{Lb}}}{{\mathrm{L}}_{\mathrm{b}}}(1-\mathrm{D})\mathrm{Ts}.$$

According to the principle of voltage-second balance:

$$\Delta {\mathrm{i}}_{\mathrm{L}}{}^{+}=\Delta {\mathrm{i}}_{\mathrm{L}}{}^{-}$$

$${\mathrm{G}}_{\mathrm{Vb}}=\frac{{\mathrm{V}}_{\mathrm{b}}}{{\mathrm{V}}_{\mathrm{O}}}=\frac{{\mathrm{N}}_{1}}{{\mathrm{N}}_{2}}\mathrm{D}=\frac{\mathrm{D}}{\mathrm{N}}.$$

#### 3.2. Voltage Stress of Components

The voltage stress of each component can be derived by analyzing its operation in the ON and OFF states. The voltage stress of switch S1 in the ON state can be determined using Figure 5a:

V

_{i}− V_{Lm}= 0On the basis of voltage-second balance, V

_{Lm}DT_{S}+ V_{Lm}(1 − D)T_{S}= 0. Therefore, we can derive the voltage stress of switch S1 to be:
$${\mathrm{V}}_{\mathrm{S}1}=\frac{1}{1-\mathrm{D}}\mathrm{Vi}$$

Because diode D2 is considered ideal, the voltage of switch S2 is equal to the difference between the battery voltage and the voltage across the input capacitor:

V

_{S2}= V_{B}− V_{Ci}In the step-down mode, when switch S4 is turned ON, switch S3 is turned OFF, as shown in Figure 9e, and the voltage across S3 is equal to the output voltage:

V

_{S3}= VoIn the step-down mode, when switch S3 is turned ON, switch S4 is turned OFF, as shown in Figure 9b, and the voltage across S4 is equal to the output voltage:

V

_{S4}= VoWhen the entire energy stored in Lb is released and switch S1 is turned ON, diode D1 is turned OFF, and the voltage difference across the diode equals the battery voltage:

V

_{D1}= V_{B}Diode D2 has the same voltage stress as switch S2 does:

V

_{D2}= V_{B}− V_{Ci}When switch S3 is turned ON in the step-down mode, the voltage stress on the input capacitor Ci is:

$${\mathrm{V}}_{\mathrm{Ci}}=\frac{{\mathrm{V}}_{\mathrm{O}}}{2}\frac{{\mathrm{N}}_{1}}{{\mathrm{N}}_{2}}$$

The voltage stress of output capacitor Co is equal to Vo:

V

_{Co}= Vo.The voltage stress of voltage-doubling capacitor Cm can be expressed as:

V

_{Cm}= Vo/2## 4. Experimental Results and Analysis

In this section, a comparison between the proposed converter and two groups of converters is presented. First, we compare the proposed converter with the bidirectional converters presented in [21,22]. As shown in Table 4, the converters presented in [21,22] can be used only as traditional bidirectional converters, whereas the proposed three-port bidirectional converter can operate in three stages and perform different functions. Apart from regular bidirectional step-up/step-down operations, the proposed converter can be used in a PV system as a three-port converter. Although the converter presented in [21] does not require a diode and has a lower turns ratio of the transformer, it has one more inductor, resulting in a larger system size. Furthermore, the efficiency of the converter presented in [21] decreases during high-power operations, and the maximum power at which the converter can be operated is 200 W. Although the converter presented in [22] can operate at 1 kW, its efficiency does not reach even 90% for operation above 500 W. In addition, controlling the switches is considerably complicated.

Table 5 presents a comparison between the proposed converter and the converters proposed in [24,25]; all of them are three-port converters. In the table, the highest conversion efficiency is the calculated efficiency of energy conversion from the PV modules to the DC bus, and the proposed converter is superior in conversion efficiency. In addition, the proposed converter has a higher step-up ratio, fewer components, and lower cost. The converter proposed in [24] requires two transformers and an inductor on the output side, and therefore, it is a larger and costlier system; moreover, it cannot achieve high efficiency in high-power operations, and its maximum efficiency is approximately 90.9%. Although the converter proposed in [25] has lower cost compared with that presented in [24], it is not superior in efficiency and system size. Moreover, its step-up ratio cannot be increased by a factor of ten despite the turns ratio of the transformer being identical to that of the proposed converter topology. Additionally, because there is no inductor on the output side, the converter proposed in [25] has low efficiency in high-power operations. Typically, a three-port converter cannot charge a battery by using energy from the DC bus, but the proposed converter has this capability; therefore, the proposed converter is more suitable for PV systems.

To verify the operation of the proposed isolated bidirectional three-port DC/DC converter, a 500 W hardware prototype was designed and implemented (Figure 10). The microcontroller unit dsPIC30F4011 was used as the control unit for the proposed converter topology, and the performance was measured for a light load of 50 W, half load of 250 W, and full load of 500 W. The measured waveforms of all components confirmed the feasibility and practicability of the proposed converter topology. The specifications and component parameters are presented in Table 6.

Figure 11 shows the measured efficiency curves of the proposed converter topology for operation in the three stages. The efficiencies corresponding to operations with the light load and full load in Stage 1 are 92% and 91%, respectively. However, for operation at approximately 200 W, the highest efficiency is 95.2%. Furthermore, for Stage 2, the highest efficiency is 94.2%. Finally, for Stage 3, the efficiencies for the light load and full load are 94.3% and 93.9%, respectively. For operation at 250 W, the highest efficiency reached was 97.6%.

Figure 12 shows a comparison graph of the measured step-up efficiencies of the proposed converter and the converters proposed in [21,22]. Although the efficiency of the proposed converter is greater than that of the converter proposed in [22], it is lower than the efficiency of the converter presented in [21]. In particular, the proposed converter can operate in three stages and at higher powers. Figure 13 shows another comparison graph of the proposed converter and the converters presented in [21,22] for operation in the step-down mode. Clearly, the proposed converter shows higher step-down efficiency.

Figure 14 shows a comparison graph of the efficiencies of the proposed converter topology and the converter topologies presented in [24,25] for operation in Stage 1. Clearly, the proposed converter topology is superior.

Figure 15 shows the measurement results for the proposed converter for Stage 1. Figure 15a shows the voltage waveforms of the DC bus V

_{bus}, voltage-doubling capacitor V_{Cm}, battery V_{B}, and input V_{i}. Figure 15b–d show the waveforms of the drive signal V_{gs}, switch voltage V_{ds}, and switch current i_{s}of switches S1, S3, and S4 for the full load, and Figure 15e depicts the current waveforms of the transformer (i_{pri}and i_{sec}) and inductor (i_{Lb}) for the same load. Figure 15 shows that the voltage stresses of S1, S3, and S4 are approximately 90, 150, and 90 V, respectively. The surge resulting from the leakage inductance when a switching is turmed off is suppressed, and V_{bus}is 200 V.Figure 16 shows the measurement results for the proposed converter for Stage 2. Figure 16a shows the waveforms of V

_{bus}, V_{Cm}, V_{B}, and V_{i}. Figure 16b–e show the waveforms of V_{gs}, V_{ds}, and i_{s}for switches S1, S2, S3, and S4 for the full load. Furthermore, Figure 16f presents the waveforms of i_{pri}, i_{sec}, and i_{Lb}for the full load. Figure 16 shows that V_{bus}is 200 V and that the voltage stresses of S1, S2, S3, and S4 are approximately 90, 90, 160, and 160 V, respectively.Figure 17 shows the measurement results for the proposed converter for Stage 3. Figure 17a presents the waveforms of V

_{bus}, V_{Cm}, V_{B}, and V_{i}. Furthermore, Figure 17b–d shows the waveforms of V_{gs}, V_{ds}, and i_{s}for switches S1, S3, and S4 for the full load, and Figure 17e depicts the waveforms of i_{pri}, i_{sec}, and i_{Lb}for the same load.## 5. Conclusions

This paper presents a theoretical analysis of the steady state, related considerations, and experimental results of the proposed three-port bidirectional DC/DC converter. The converter operates in a PV step-up mode, battery step-up mode, and DC bus step-down mode, reflecting the high circuit functionality of the circuit. An improved boost-flyback converter and a DC -blocking capacitor are included on the low-voltage side and high-voltage side, respectively, to increase the voltage conversion ratio and to reduce the voltage on the transformer, both of which are necessary for high step-down performance. In addition, the energy stored in the leakage inductor is recycled, which improves the overall efficiency of the system. The proposed converter offers the advantages of high efficiency, a high step-up/step-down ratio (8–10/8.33), small system size, and low cost. Finally, a 500 W three-port bidirectional converter was implemented to verify its feasibility and practicability. The highest efficiencies of the proposed converter for operation in the PV step-up stage, battery step-up stage, and step-down stage were 95.2%, 94.2%, and 97.6%, respectively, and the highest efficiencies associated with the maximum output power were 91%, 94%, and 96%, respectively.

## Author Contributions

These authors contributed equally to this work.

## Conflicts of Interest

The authors declare no conflict of interest.

## Nomenclature

L_{m} | Magnetization inductor |

Co | Output Capacitor |

N_{1} | Primary Turns |

L_{Lk1} | Primary Leakage |

V_{o} | Output Voltage |

V_{2} | Second Voltage |

V_{Lk1} | Primary Leakage Voltage |

Ts | Switch period |

G_{Vo} | Step-up voltage conversion ratio |

i_{o} | Output Current |

i_{Lb} | Charge Current |

i_{S1} | Switch S_{1} Current |

i_{S4} | Switch S_{4} Current |

V_{S2} | Switch S_{2} Voltage Stress |

V_{D1} | Diode D_{1} Voltage Stress |

L_{b} | Charger inductor |

C_{m} | Double Voltage Capacitor |

N_{2} | Second Turns |

L_{Lk2} | Second Leakage |

V_{b} | Battery Voltage |

V_{Ci} | Input Capacitor Voltage |

V_{Lk2} | Second Leakage Voltage |

T_{on} | Switch turn-on time |

G_{Vb} | Step-down voltage conversion ratio |

i_{Lm} | Magnetization inductor Current |

i_{pri} | Primary Current |

i_{S2} | Switch S_{2} Current |

i_{Lb(max)} | Charger Maximum Current |

V_{S3} | Switch S_{3} Voltage Stress |

V_{D2} | Diode D_{2} Voltage Stress |

C_{i} | Input Capacitor |

D_{o} | PV Blocking Diode |

N | Turns ratio |

V_{i} | PV input Voltage |

V_{1} | Primary Voltage |

V_{Cm} | Double Voltage Capacitor Voltage |

D | Switch Duty Cycle |

T_{off} | Switch turn-off time |

i_{in} | PV Input Current |

i_{b} | Battery Charge Current |

i_{sec} | Second Current |

i_{S3} | Switch S_{3} Current |

V_{S1} | Switch S_{1} Voltage Stress |

V_{S4} | Switch S_{4} Voltage Stress |

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**Figure 5.**Operational diagram of each mode in Stage 1: (

**a**) Mode I; (

**b**) Mode II; (

**c**) Mode III; (

**d**) Mode IV; (

**e**) Mode V; and (

**f**) Mode VI.

**Figure 7.**Operating diagram of each mode in Stage 2: (

**a**) Mode I; (

**b**) Mode II; (

**c**) Mode III; (

**d**) Mode IV; (

**e**) Mode V; and (

**f**) Mode VI.

**Figure 9.**Operating diagram of each mode in Stage 3: (

**a**) Mode I; (

**b**) Mode II; (

**c**) Mode III; (

**d**) Mode IV; (

**e**) Mode V; and (

**f**) Mode VI.

**Figure 10.**Photographs of the proposed three-port bidirectional DC/DC converter: (

**a**) the converter circuit and (

**b**) the control circuit.

**Figure 12.**Comparison graphs of the measured efficiencies of the proposed converter and existing bidirectional converters for operation in the step-up mode.

**Figure 13.**Comparison graphs of measured efficiencies of the proposed converter and existing bidirectional converters for operation in the step-down mode.

**Figure 14.**Comparison graphs of the measured efficiencies of the proposed converter and other three-port converters.

**Figure 15.**Measurement results for Stage 1 for a load of 500 W: (

**a**) output voltage waveforms; (

**b**) switch S1 waveforms; (

**c**) switch S3 waveforms; (

**d**) switch S4 waveforms; and (

**e**) current waveforms of the transformer and inductors.

**Figure 16.**Measurement results for Stage 2: (

**a**) output voltage waveforms; (

**b**) switch S1 waveforms; (

**c**) switch S2 waveforms; (

**d**) switch S3 waveforms; (

**e**) switch S4 waveforms; and (

**f**) current waveforms of the transformer and inductors.

**Figure 17.**Measurement results for Stage 3: (

**a**) output voltage waveforms; (

**b**) switch S1 waveforms, (

**c**) switch S3 waveforms; (

**d**) switch S4 waveforms; and (

**e**) current waveforms of the transformer and inductors.

Mode | Comments | Mode | Comments |
---|---|---|---|

Mode I (t _{0} ≤ t < t_{1}) | As shown in Figure 5a, when t = t_{0}, S1 and D1 turn ON, S2, S4 and D2 turn OFF; and the parasitic diode of switch S3 is also ON. The PV source stores energy in Lm, and Lb releases energy for battery charging. Simultaneously, the Cm is charged by the current induced by the transformer. In this interval, Vo is provided by the output capacitor Co. | Mode II (t _{1} ≤ t < t_{2}) | When t = t_{1}, S1 and D1 continue to be in the ON state, S2, S4, and D2 remain OFF. S3 is turned ON for synchronous rectification, and the direction of current flow is as shown in Figure 5b. Other operations are the same as those in Mode I. |

Mode III (t _{2} ≤ t < t_{3}) | When t = t_{2}, S1 and D1 remain ON, S2, S4, and D2 remain OFF. The parasitic diode on S3 is turned ON, and the direction of current flow is as shown in Figure 5c. The other operations in this mode are the same as those in Mode I. | Mode IV (t _{3} ≤ t < t_{4}) | When t = t_{3}, S1 is turned OFF, D1 and the parasitic diode on S3 remain ON, and S2, S4, and D2 continue to be OFF. The direction of current flow in this mode is shown in Figure 5d. Current flows through the L_{Lk1}, and charges the parasitic capacitor on S1. However, this charging process is stopped when the voltage of the parasitic capacitor on S1 exceeds the sum of the V_{b} and V_{Lb}. In addition, current flowing through the L_{Lk2} charges Cm, and Lb continues to release energy for battery charging. The Vo continues to be provided by Co. |

Mode V (t _{4} ≤ t < t_{5}) | When t = t_{4}, S1, S2, S3, and D2 are OFF, D1 remains ON. S4 is turned ON for synchronous rectification, and the direction of current flow is shown in Figure 5e. Energy stored in Lm is transferred to the Vo, and Co is charged by the current induced by the transformer. The Cm also releases energy to the Vo, and Lm and Lb release energy for battery charging. | Mode VI (t _{5} ≤ t < t_{6}) | When t = t_{5}, S1, S2, S3, and D2 remain OFF and D1 continues to be ON. In this mode, the direction of current flow is as shown in Figure 5f; moreover, the parasitic diode on S4 is turned ON; synchronous rectification is stopped. Other operations are the same as those in Mode V. |

Mode | Comments | Mode | Comments |
---|---|---|---|

Mode I (t _{0} ≤ t < t_{1}) | When t = t_{0}, S1 and S2 and D2 turn ON, S4 and D1 turn OFF, the direction of current flow is as shown in Figure 7a, and the parasitic diode on S3 is ON; the battery provides energy to Lm. The current induced by the transformer charges the Cm, and Co provides energy to Vo. | Mode II (t _{1} ≤ t < t_{2}) | When t = t_{1}, S1, S2, and D2 continue to be ON, and S4 and D1 remain OFF. S3 is turned ON for synchronous rectification, and the direction of current flow is as shown in Figure 7b. The other operations are the same as Mode I. |

Mode III (t _{2} ≤ t < t_{3}) | When t = t_{2}, S1, S2, and D2 remain ON and S4 and D1 continue to be in the OFF state. Synchronous rectification is stopped, and the parasitic diode on S3 is turned ON. The direction of current flow is as shown in Figure 7c. The other operations in this mode are the same as Mode I. | Mode IV (t _{3} ≤ t < t_{4}) | When t = t_{3}, S4 and D2 remain OFF, D1 and the parasitic diode on S3 are turned ON, and S1 and S2 are turned OFF. The direction of current flow is as shown in Figure 7d. Current flowing through the L_{Lk1} charges the parasitic capacitor on S1. When the voltage of the parasitic capacitor exceeds the sum of the V_{b} and V_{Lb}, this charging process is stopped. Furthermore, current flowing through the L_{Lk2} charges Cm. Lb starts releasing energy to the battery for energy recycling, and Co provides energy to Vo. |

Mode V (t _{4} ≤ t < t_{5}) | When t = t_{4}, S1, S2, S3, and D2 continue to be in the OFF state, and D1 remains ON. S4 is turned ON for synchronous rectification. The direction of current flow is as shown in Figure 7e, and the transformer induces energy from Lm to Vo and charges Co simultaneously. The energy stored in Cm is also released to Vo, and the energy stored in Lm and Lb is recycled and supplied to the battery. | Mode VI (t _{5} ≤ t < t_{6}) | When t = t_{5}, S1, S2, S3, and D2 continue to be in the OFF state and D1 remains ON. Synchronous rectification is stopped because the parasitic diode on S4 is turned ON, and the direction of current flow in this mode is as shown in Figure 7f. Other operations in this mode are the same as Mode V. |

Mode | Comments | Mode | Comments |
---|---|---|---|

Mode I (t _{0} ≤ t < t_{1}) | When t = t_{0}, S4 and D1 are turned ON, S1, S2, S3, and diode D2 are turned OFF. Furthermore, the parasitic diode on switch S1 is turned ON. The direction of current flow is shown in Figure 9a. The energy stored in the Cm and L_{Lk2} is recycled and provided to Vo, and the L_{Lk1} transfers its energy to the Ci. Simultaneously, Lb releases energy for battery charging. | Mode II (t _{1} ≤ t < t_{2}) | When t = t_{1}, S4 and D1 continue in the ON state, S1, S2, S3, and D2 remain OFF. The direction of current flow is shown in Figure 9b. The PV source starts storing energy in the Lm after the energy stored in Cm and L_{Lk2} is completely recycled. Furthermore, the current induced by the transformer starts storing energy in Lb and charging the battery simultaneously. |

Mode III (t _{2} ≤ t < t_{3}) | When t = t_{2}, S1, S2, and D2 continue to be in the OFF state and D1 remains ON. In addition, S4 is turned OFF and the parasitic diode on S3 is turned ON. The direction of current flow is as shown in Figure 9c. Current flowing through the L_{Lk2} charges Cm. The current induced by the transformer stores energy in Lb and charges the battery simultaneously. | Mode IV (t _{3} ≤ t < t_{4}) | When t = t_{3}, S2, S4, and D2 remain OFF and D1 continues to be in the ON state. Both S1 and S3 are turned ON for synchronous rectification. The direction of current flow in this mode is as shown in Figure 9d. The energy stored in Lm is induced by the transformer to charge Ci and Cm through S1 and S3, respectively. Furthermore, Lb starts releasing energy through S1 and charging the battery. |

Mode V (t _{4} ≤ t < t_{5}) | When t = t_{4}, S2, S4, and D2 remain OFF, S1, S3, and D1 continue to be in the ON state. The direction of current flow is shown in Figure 9e. Both S1 and S3 continue to perform synchronous rectification, and the entire energy stored in Lm is released. Simultaneously, Cm starts transferring energy to Lm and L_{Lk2} and induces energy by the transformer to charge Ci. Lb continues to charge the battery through S1. | Mode VI (t _{5} ≤ t < t_{6}) | When t = t_{5}, D1 remains ON, S3, S4, and D2 continue to be OFF. The direction of current flow is shown in Figure 9f. In this mode, synchronous rectification is stopped as S1 and S4 are turned OFF and their parasitic diodes are ON. Current flows through Cm and L_{Lk2}, and energy stored in Cm and L_{Lk2} is recycled to Vo. Furthermore, the energy stored in Lb is released for battery charging. |

Aspect | Proposed Converter | Converter Proposed in [21] | Converter Proposed in [22] |
---|---|---|---|

Topology | Isolated | Isolated | Isolated |

Input voltage | 24 V | 24 V | 24 V |

Output voltage | 200 V | 200 V | 200 V |

Output power | 500 W | 200 W | 1 kW |

Number of switches | 4 | 4 | 4 |

Number of diodes | 2 | 0 | 2 |

Number of inductors | 1 | 2 | 1 |

Number of capacitors | 3 | 3 | 5 |

Turns ratio | N = 3 | N = 1.5 | N = 3 |

Number of output terminals | 2 | 1 | 2 |

Maximum conversion efficiency (step up/step down) | 95.2%/97.6% | 94%/97% | 95.6%/96.3% |

Cost | Low | High | Moderate |

Number of operating modes | 3 | 2 | 2 |

Aspect | Proposed Converter | Converter Proposed in [24] | Converter Proposed in [25] |
---|---|---|---|

Topology | Isolated | Isolated | Isolated |

Input voltage | 24 V | 300–400 V | 18 V |

Battery voltage | 24 V | 160 V | 12 V |

Output voltage | 200 V | 48 V | 80 V |

Output power | 500 W | 1 kW | 200 W |

Number of switches | 4 | 4 | 4 |

Number of diodes | 2 | 4 | 2 |

Number of transformers | 1 | 2 | 2 |

Number of inductors | 1 | 1 | 0 |

Number of capacitors | 3 | 2 | 3 |

Turns ratio | N = 3 | N = 0.5 | N = 2 |

Number of output terminals | 2 | 2 | 2 |

Maximum conversion efficiency (step up) | 95.2% | 90.9% | 91.3% |

Operating modes | 3 | 2 | 2 |

Parameter | Specification | |

Input dc voltage (V_{in}) | 20–26 V | |

Battery voltage (V_{B}) | 24 V | |

Output dc voltage (V_{o}) | 200 V | |

Maximum output power (P_{o}) | 500 W | |

Switching frequency (f_{s}) | 50 kHz | |

Turns ratio of coupled inductors | N_{1}:N_{2} = 1:3 | |

Component | Model | Specification |

S1, S2 | IRFP4321 | 150 V/78 A |

S3, S4 | IRFP4868 | 300 V/70 A |

D1, D2 | MBR40200 | 200 V/40 A |

Ci | Electrolytic capacitor | 100 μF/100 V |

Co | Electrolytic capacitor | 470 μF/450 V |

Cm | Metallized polypropylene (MPP) film capacitor | 4.7 μF/250 V |

Lb | MPP ring core | 18 μH |

Lm | MPP ring core | 12 μH |

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