Analysis and Implementation of a Frequency Control DC–DC Converter for Light Electric Vehicle Applications

In order to realize emission-free solutions and clean transportation alternatives, this paper presents a new DC converter with pulse frequency control for a battery charger in electric vehicles (EVs) or light electric vehicles (LEVs). The circuit configuration includes a resonant tank on the high-voltage side and two variable winding sets on the output side to achieve wide output voltage operation for a universal LEV battery charger. The input terminal of the presented converter is a from DC microgrid with voltage levels of 380, 760, or 1500 V for house, industry plant, or DC transportation vehicle demands, respectively. To reduce voltage stresses on active devices, a cascade circuit structure with less voltage rating on power semiconductors is used on the primary side. Two resonant capacitors were selected on the resonant tank, not only to achieve the two input voltage balance problem but also to realize the resonant operation to control load voltage. By using the variable switching frequency approach to regulate load voltage, active switches are turned on with soft switching operation to improve converter efficiency. In order to achieve wide output voltage capability for universal battery charger demands such as scooters, electric motorbikes, Li-ion e-trikes, golf carts, luxury golf cars, and quad applications, two variable winding sets were selected to have a wide voltage output (50~160 V). Finally, experiments with a 1 kW rated prototype were demonstrated to validate the performance and benefits of presented converter.


Introduction
Clean renewable energies with power electronic techniques have been widely developed to generate alternative current (AC) voltage or direct current (DC) voltage on AC utility systems [1][2][3][4] or DC microgrid systems [5,6]. Normally, DC/DC pulse-width modulation converters (PWMs) and AC/DC PWM converters are adopted for solar power [7,8] and wind power [9] applications to convert unstable DC and AC voltage into a stable DC voltage on DC nanogrid or microgrid systems. The DC bus voltage on a DC microgrid may be 380, 760, or 1500 V for residential houses, light rail vehicles, or DC traction vehicles applications, respectively. DC-DC converters [10][11][12][13] with full bridge circuit topologies have been developed to convert low voltage (380 V) input into low voltage units (5, 12, or 48 V) for computers, server systems, light electric vehicle (LEVs), or telecommunication applications. Three level or multilevel converters [14][15][16] have been proposed for medium voltage (760 V) or high voltage (1500 V) input applications. Soft switching converters were researched in [17][18][19][20][21][22] to decrease switching losses on power semiconductors and high efficiency. Active clamp pulse-width modulation (PWM) and asymmetric PWM techniques were studied in [17,18] to improve the switching loss from half rated power to full power by adding an extra inductor on the primary side of a DC/DC PWM converter. However, the main problems of asymmetric PWM converters are unbalanced voltages and current stresses on power devices. Using the phase shift PWM technique on full bridge converters [19,20] can reduce switching loss and obtain a high circuit efficiency. The main disadvantage of phase shift PWM techniques is the hard switching problem on the lagging-leg switches. Resonant converters with the pulse frequency modulation technique were presented in [21,22] to realize low switching losses on power semiconductors over the whole load range. However, the main drawback of resonant converters is their narrow input voltage range.
To implement clean transportation alternatives and realize emission-free demands, a new PWM converter with the advantages of a wide voltage output capability and low switching loss is presented and verified in this paper for a universal battery charger in LEV or electric vehicle (EV) applications. A cascade resonant circuit was selected to lessen the voltage rating on active devices so that 600 V power switches are adopted for the 760 V input condition. Two resonant capacitors are used on the resonant tank to not only achieve resonant behavior but also input spilt voltage balance. Due to the variable switching frequency, active devices are turned on under zero voltage switching (ZVS), and the fast recovery diodes are turned off under zero current switching (ZCS). Therefore, the power losses are reduced in the presented converter. To implement the wide voltage output capability for universal battery charger applications in LEV or EV systems, two winding sets were selected for the low voltage side to have different voltage gains under high or low output voltage regions. The presented circuit has a more concise circuit configuration, less device counts, and an easier control strategy when using the general purpose PWM integrated circuit compared to the conventional DC converters. The basic DC microgrid system and the circuit configuration of the presented converter are provided in Section 2. The circuit operation for wide voltage operation capability is discussed in Section 3. The circuit analysis of the studied converter is provided in Section 4. Section 5 gives the design procedures and experiments of the presented circuit. In Section 6, the conclusion and future work directions are provided.

Circuit Configuration
The basic circuit blocks of DC nanogrid or microgrid systems are illustrated in Figure 1. The input voltages on a DC microgrid can be DC or AC voltage from clean renewable energies such as wind power and photovoltaic (PV) power or utility systems. Therefore, the AC/DC or DC/DC PWM converters need to change unstable AC voltage from wind turbine generators or unstable DC voltage from PV panels to stable DC voltage on DC microgrid systems. Bidirectional AC/DC or DC/DC PWM converters are needed between utility and DC microgrid systems to achieve bidirectional power flow capability. DC microgrids can supply AC motor drives, energy storage units, light rail transit systems, or battery chargers for LEVs through DC/AC inverters or DC/DC converters. The input voltage of DC transportation applications is 750 or 1500 V. However, the input voltage of a residential house or a local industry factory is 380, ±380, or 760 V. Therefore, the DC bus voltage on DC microgrids can be 380, ±380, 760, or 1500 V for different power rating requirements. High voltage PWM converters are normally adopted to supply the high power output for EV chargers or DC transportation systems. Figure 2 provides a circuit diagram of the studied high voltage DC/DC converter. The input voltage is 760 V from the DC bus of the DC microgrid. The output voltage of the studied converter is used to charge batteries for LEV applications such as scooters, electric motorbikes, Li-ion e-trikes, golf carts, luxury golf cars, and quads. Due to the wide battery voltage range in LEV applications, DC/DC converters need to have wide output voltage operation capabilities. Figure 2a provides a diagram of the presented circuit with V in = 760 V and V o = 50 V-160 V. Four active devices, two input capacitors, an isolated transformer, a resonant inductor, and two resonant capacitors are used on the high voltage side (primary side). Four fast recovery diodes, an active switch, an output capacitor, and a DC resistor are adopted on the low voltage side (secondary side). Due to the series connection of four active devices on the high voltage side, active devices with a 600 V voltage rating are adopted on the primary side to withstand 760 V input. The series resonant circuit structure with L r , C r1 , C r2 , and L m is adopted on the presented converter to have the advantage of a soft switching operation for active devices (S 1 -S 4 ) and fast  The input voltage is 760 V from the DC bus of the DC microgrid. The output voltage of the studied converter is used to charge batteries for LEV applications such as scooters, electric motorbikes, Li-ion e-trikes, golf carts, luxury golf cars, and quads. Due to the wide battery voltage range in LEV applications, DC/DC converters need to have wide output voltage operation capabilities. Figure 2a provides a diagram of the presented circuit with Vin = 760 V and Vo = 50 V-160 V. Four active devices, two input capacitors, an isolated transformer, a resonant inductor, and two resonant capacitors are used on the high voltage side (primary side). Four fast recovery diodes, an active switch, an output capacitor, and a DC resistor are adopted on the low voltage side (secondary side). Due to the series connection of four active devices on the high voltage side, active devices with a 600 V voltage rating are adopted on the primary side to withstand 760 V input. The series resonant circuit structure with Lr, Cr1, Cr2, and Lm is adopted on the presented converter to have the advantage of a soft switching operation for active devices (S1-S4) and fast recovery diodes (D1-D4). Two center-taped rectifiers with different winding turns are used on the low voltage side to extend the output voltage range operation. If switch S5 is OFF ( Figure  2b), D1 and D4 are OFF. The low voltage output is provided in the proposed converter with ns secondary winding turns. The voltage gain at the low voltage output condition ( Figure 2b) is Vo/Vin = G(fsw)ns/(4np), where G(fsw) is the voltage gain of the series resonant circuit. If switch S5 is ON (Figure 2c), D2 and D3 are OFF and the high voltage output is provided in the proposed converter with 2ns secondary winding turns. Therefore, the voltage gain at the high voltage output condition is Vo/Vin = G(fsw)ns/(2np). Thus, the studied converter can provide a low (high) voltage output range with S5 OFF (ON).   The input voltage is 760 V from the DC bus of the DC microgrid. The output voltage of the studied converter is used to charge batteries for LEV applications such as scooters, electric motorbikes, Li-ion e-trikes, golf carts, luxury golf cars, and quads. Due to the wide battery voltage range in LEV applications, DC/DC converters need to have wide output voltage operation capabilities. Figure 2a provides a diagram of the presented circuit with Vin = 760 V and Vo = 50 V-160 V. Four active devices, two input capacitors, an isolated transformer, a resonant inductor, and two resonant capacitors are used on the high voltage side (primary side). Four fast recovery diodes, an active switch, an output capacitor, and a DC resistor are adopted on the low voltage side (secondary side). Due to the series connection of four active devices on the high voltage side, active devices with a 600 V voltage rating are adopted on the primary side to withstand 760 V input. The series resonant circuit structure with Lr, Cr1, Cr2, and Lm is adopted on the presented converter to have the advantage of a soft switching operation for active devices (S1-S4) and fast recovery diodes (D1-D4). Two center-taped rectifiers with different winding turns are used on the low voltage side to extend the output voltage range operation. If switch S5 is OFF ( Figure  2b), D1 and D4 are OFF. The low voltage output is provided in the proposed converter with ns secondary winding turns. The voltage gain at the low voltage output condition ( Figure 2b) is Vo/Vin = G(fsw)ns/(4np), where G(fsw) is the voltage gain of the series resonant circuit. If switch S5 is ON (Figure 2c), D2 and D3 are OFF and the high voltage output is provided in the proposed converter with 2ns secondary winding turns. Therefore, the voltage gain at the high voltage output condition is Vo/Vin = G(fsw)ns/(2np). Thus, the studied converter can provide a low (high) voltage output range with S5 OFF (ON).

Circuit Operation
To realize the wide output voltage operation, the presented circuit has two equivalent sub-circuits, as shown in Figure 2b,c. S1-S4 are controlled with pulse switching fre-

Circuit Operation
To realize the wide output voltage operation, the presented circuit has two equivalent sub-circuits, as shown in Figure 2b,c. S 1 -S 4 are controlled with pulse switching frequency modulation (PFM). For low and high output voltage ranges, the switching signals and the main voltage and current waveforms are provided in Figure 3a,b, respectively. For the low voltage output range, the studied converter is only operated at a low voltage gain. Therefore, S 5 is off, and D 1 and D 4 are also off. In order to generate the square voltage PWM waveforms on leg voltages v ab and v bc , the gate waveforms of S 1 (S 2 ) and S 3 (S 4 ) are identical. If the proposed converter is used at the switching frequency f sw < (or >) series resonant frequency f r,1 by C r1 , C r2 , and L r , the presented resonant converter has six (or four) working modes in every switching cycle. Figure 4 shows the equivalent working modes under f sw < f r,1 . If the converter is operated at f sw > f r,1 , then only modes 1, 3, 4, and 6 are used in the proposed converter for every switching period.

Circuit Operation
To realize the wide output voltage operation, the presented circuit has two equivalent sub-circuits, as shown in Figure 2b,c. S1-S4 are controlled with pulse switching frequency modulation (PFM). For low and high output voltage ranges, the switching signals and the main voltage and current waveforms are provided in Figure 3a,b, respectively. For the low voltage output range, the studied converter is only operated at a low voltage gain. Therefore, S5 is off, and D1 and D4 are also off. In order to generate the square voltage PWM waveforms on leg voltages vab and vbc, the gate waveforms of S1 (S2) and S3 (S4) are identical. If the proposed converter is used at the switching frequency fsw < (or >) series resonant frequency fr,1 by Cr1, Cr2, and Lr, the presented resonant converter has six (or four) working modes in every switching cycle. Figure 4 shows the equivalent working modes under fsw < fr,1. If the converter is operated at fsw > fr,1, then only modes 1, 3, 4, and 6 are used in the proposed converter for every switching period.

Mode 1 [t 0~t1 ]
: v CS1 = v CS3 = 0 at time t 0 . Due to i CS3 < 0 and i CS1 < 0, the body diodes D S3 and D S1 are naturally conducting. The ZVS turn-on of S 3 and S 1 is accomplished at t 0 .
]: since f r,1 < f sw , i D2 will decrease to 0 at t 1 . Then, D 1 becomes off without reverse recovery current loss. On the primary side, L m , L r , C r2 , and C r1 are naturally resonant with the resonant frequency f r,2 = 1/2π 2(L r + L m )C r . Mode 3 [t 2~t3 ]: S 1 and S 3 are turned off at t 2 , and C S2 and C S4 are discharged. After time t 2 , i Lm > i Lr such that D 3 is conducting and v Lm becomes −(n p /n s )V o . The magnetizing current i Lm (t 2 ) can be obtained as follows:      Mode 1 [t0 ~ t1]: vCS1 = vCS3 = 0 at time t0. Due to iCS3 < 0 and iCS1 < 0, the body diodes DS3 and DS1 are naturally conducting. The ZVS turn-on of S3 and S1 is accomplished at t0. The capacitor voltage vCr2 + vCr1 = Vdc1. Due to iLr (t) > iLm(t), the fast recovery diode D2 is conducting and vLm = (np/ns)Vo. Lr, Cr2, and Cr1 are resonant in mode 1 with resonant frequency r r r Mode 2 [t1 ~ t2]: since fr,1 < fsw, iD2 will decrease to 0 at t1. Then, D1 becomes off without reverse recovery current loss. On the primary side, Lm, Lr, Cr2, and Cr1 are naturally resonant with the resonant frequency Mode 3 [t2 ~ t3]: S1 and S3 are turned off at t2, and CS2 and CS4 are discharged. After time t2, iLm > iLr such that D3 is conducting and vLm becomes −(np/ns)Vo. The magnetizing current iLm(t2) can be obtained as follows: The ZVS condition of S 4 and S 2 is given as: where C oss = C S1 = C S2 = C S3 = C S4 . However, the dead time t d between S 2 and S 1 must be greater than the discharged time of C S2 and C S4 . Therefore, the maximum value of L m can be obtained as follows: Mode 6 [t 5~Tsw +t 0 ]: S 2 and S 4 turn off at t 5 , and C S1 and C S3 are discharged. After time t 5 , i Lm < i Lr such that D 2 becomes forward-biased and v Lm = ( For the high voltage output range, the studied converter has a high voltage gain. S 5 is turned on, and the fast recovery diodes D 2 and D 3 are reverse-biased. The presented converter has six (or four) operating modes in every switching cycle if f sw < (or >) f r,1 . Figure 5 provides the equivalent circuits for f sw < f r,1 .

Circuit Analysis
For the design of the resonant converter, a general pulse frequency modulation was selected to generate the PWM signals for all power switches. Power switch S5 is controlled Mode 1 [t 0~t1 ]: at time t 0 , v CS1 = v CS3 = 0. D S1 and D S3 are conducting, v S2,ds = v S4,ds = V in /2, and v Cr1 + v Cr2 = V in /2. D 1 is conducting so that v Lm = n p V o /(2n s ).
Mode 2 [t 1~t2 ]: if f r,1 > f sw , the i D1 will be decreased to 0 at t 1 . Then, D 1 is off. Mode 3 [t 2~t3 ]: at t 2 , S 1 and S 3 turn off. Due to i Lr < i Lm , the fast recovery diode D 4 becomes forward-biased and v Lm = −n p V o /(2n s ). The magnetizing current i Lm (t 2 ) can be obtained as follows: The ZVS condition of S 4 and S 2 for the high voltage output region is obtained as follows: i In order to have ZVS operation, the maximum value of L m for the high voltage output region is given as follows:

Circuit Analysis
For the design of the resonant converter, a general pulse frequency modulation was selected to generate the PWM signals for all power switches. Power switch S 5 is controlled by using an input voltage comparator to select low (or high) winding turns on the output side under the low (or high) voltage output region. For frequency analysis under frequency modulation, the equivalent AC circuit of the proposed converter on the primary side is given in Figure 6a. In Figure 6a, V in,e and R e are the equivalent AC input voltage and AC resistor on the primary side, respectively. The voltage V in,e is a square voltage waveform with voltage values of 0 V and V in /2. One can obtain the input voltage at fundamental frequency V in,e, f = V in /( √ 2π). Components C r1 and C r2 are connected in parallel (C r,eq = C r1 + C r2 = 2C r ) under the equivalent AC resonant tank. For the low voltage output range, the turn-ratio of the isolation transformer is n p /n s (S 5 off). However, the transformer turn-ratio is n p /(2n s ) under the high voltage output range (S 5 on). One can obtain the magnetizing voltage v Lm = ±n p V o /n s (low voltage output range) or ±n p V o /(2n s ) (high voltage output range). The magnetizing voltage at the fundamental frequency can be obtained as V Lm, f = 2 √ 2n p V o /(n s π) (or √ 2n p V o /(n s π)) for the low output voltage range (or high output voltage range). The equivalent AC resistor R e on the primary side is given as follows: The voltage transfer function between the output and input sides is given in Equation (8) and shown in Figure 6b. where x = √ L r /2C r /R e , l n = L m /L r , and f n = f sw /f r,1 . From Equation (8), V o can be obtained as follows:

Design Considerations and Experiments
In this presented converter, the following are the basic input and output electric specifications: Vin = 760 V, Vo = 50~160 V, Po = 1000 W, fr,1 = 100 kHz, and ln = 7.5. For the low voltage output range, Vo = 50~90 V. For the high voltage output range, Vo is between

Design Considerations and Experiments
In this presented converter, the following are the basic input and output electric specifications: V in = 760 V, V o = 50~160 V, P o = 1000 W, f r,1 = 100 kHz, and l n = 7.5. For the low voltage output range, V o = 50~90 V. For the high voltage output range, V o is between 90 and 160 V. The circuit design procedure is based on the low voltage output range (V o = 50~90 V). The design voltage gain of the presented converter is unity at f sw = f r,1 with V in = 760 V and V o = 50 V. From Equation (8), n p /n s can be obtained as follows: The transformer T is implemented with the ferrite core (TDK EE-55) with A e = 3.54 cm 2 and ∆B = 0.4 T. The assumed minimum switching frequency f sw,min = 60 kHz at V in = 90 V. The primary turns n p,min can be obtained in Equation (11): In this laboratory prototype, the selected winding turns were n p = 32 and n s = 8. Then, DC voltage gains were obtained through Equations (12) and (13): For the high output voltage range (V o = 90~160 V), the maximum and minimum DC gains of the presented converter could be obtained as follows: Under the low output voltage range and the full load condition, R e is given in Equation (16) at V o = 90 V: The quality factor x = 0.05 and inductor ratio l n = 7.5 were selected in this design procedure, and C r1 , C r2 , L r , and L m could be derived in Equations (17)- (19).  Table 1 illustrates the circuit parameters in the laboratory prototype. Figure 7 shows a picture of the prototype circuit in the laboratory test.  The experimental results of the proposed converter for low voltage range operation are illustrated in Figures 8 and 9. Figure 8 illustrates the experiments of the presented circuit under Vin = 760 V and Vo = 50 V at a 1 kW load. Figure 8a gives the experimental PWM waveforms of S1-S4. S1 and S3 were found to have the same PWM signals. In the same manner, the PWM signals of S2 and S4 were identical. Figure 8b shows the PWM signal of S1 and leg voltages vab and vbc. If S1 and S3 were on and S2 and S4 were off, then The experimental results of the proposed converter for low voltage range operation are illustrated in Figures 8 and 9. Figure 8 illustrates the experiments of the presented circuit under V in = 760 V and V o = 50 V at a 1 kW load. Figure 8a gives the experimental PWM waveforms of S 1 -S 4 . S 1 and S 3 were found to have the same PWM signals. In the same manner, the PWM signals of S 2 and S 4 were identical. Figure 8b shows the PWM signal of S 1 and leg voltages v ab and v bc . If S 1 and S 3 were on and S 2 and S 4 were off, then the leg voltages v ab = V Cdc1 = V in /2 = 380 V and v bc = 0. When S 1 and S 3 were off and S 2 and S 4 were on, the leg voltages v ab = 0 and v bc = V Cdc2 = V in /2 = 380 V. Figure 8c illustrates the experimental waveforms of v Cr1 , v Cr2 , and i Lr . The AC voltage components of v Cr1 and v Cr2 were found to be complementary to each other, and the voltage value v Cr1 + v Cr2 = V in /2. Due to the half bridge type resonant circuit, the DC voltage values v Cr1,DC = v Cr2,DC = V in /4 = 190 V. The presented converter at V o = 50 V had a G DC,min,L = 1.05 theoretical DC voltage gain. The switching frequency was close to the resonant frequency. Therefore, the inductor current i Lr was a sinusoidal waveform. Since the presented converter at V o = 50 V was operated under the low voltage output region, S 5 was off and D 1 and D 4 were reverse-biased. Figure 8d illustrates the experimental waveforms of the secondary-side currents i D2 , i D3 , and I o and load voltage V o . The PWM signals of S 1 at P o = 200 W (20% load) and 1 kW (100% load) are given in Figure 8e,f, respectively. It can be seen in Figure 8e,f that the active device S 1 turned on at ZVS from 20% load to 100% load. In the same manner, the experiments of the presented converter under V o = 90 V and 1 kW load are illustrated in Figure 9. Following from Equations (12) and (13), the converter at V o = 90 V output had more voltage gain than at the V o = 50 V output condition. Thus, the switching frequency in Figure 9a-d (at V o = 90 V output) was less than the switching frequency in Figure 8a-d (at V o = 50 V output). For V o = 90 V output, the theoretical DC voltage gain in Equation (12) was 1.89. Therefore, it could be expected that the switching frequency was less than the series resonant frequency shown in Figure 9c and the fast recovery diodes D 2 and D 3 could be turned off at the zero current switching shown in Figure 9d. From the experiments shown in Figure 9e,f, it is clear that S 1 turned on at ZVS from 20% load to 100% load. Similarly, the experiments of the converter under high voltage range operation are provided in Figures 10 and 11. For the high voltage output operation, S 5 was on and D 2 and D 3 were reverse-biased. Figures 10  and 11 illustrate the experiments for the V o = 95 and 160 V output and 100% load conditions. Following Equations in (14) and (15), the voltage gain of converter at V o = 95 V was close to unity and the DC voltage gain at V o = 160 V was greater than unity. Therefore, the switching frequency of the presented converter at the V o = 95 V (160 V) output was greater (less) than the resonant frequency. This phenomenon can be observed in Figures 10a-d and 11a-d. For the V o = 95 V condition, it can be seen that f sw > f r,1 . In Figure 10d, the fast recovery diodes D 1 and D 4 were turned off at hard switching. On the other hand, the switching frequency f sw < f r,1 under the V o = 160 V output condition. Therefore, D 1 and D 4 were turned off at zero current switching, as shown in Figure 11d. From the test experiments in Figure 10e,f and Figure 11e,f, it can be seen that S 1 was turned off at ZVS from 20% load to full load for both V o = 95 and 160 V. Since the circuit characteristics of S 2 , S 3 , and S 4 are the same as S 1 , it could be expected that S 1 -S 4 had the soft switching turn-on operation. The experiments of output voltage V o and switch S 5 are provided in Figure 12 during the output voltage variation between 50 and 160 V. If the output voltage was less than 90 V, S 5 was off (v S5,g = 0 V). On the other hand, S 5 was on (v S5,g = 15 V) if V o > 90 V. The presented converter had 90%, 94%m and 91% efficiencies at V o = 50, 95, and 160 V, respectively, under the V in = 760 V and P o = 1 kW conditions. The proposed converter at V o = 95 V was operated at f sw > f r,1 . Therefore, power switches were operated at ZVS turn-on and rectifier diodes were operated at hard switching turn-off. Due to the f sw > f r,1 at the V o = 95 V condition, the higher switching frequency reduced the magnetizing current loss compared to the V o = 160 V condition (f sw < f r,1 ). Therefore, the circuit efficiency at the V o = 95 V condition was better than V o = 160 V. V, respectively, under the Vin = 760 V and Po = 1 kW conditions. The proposed converter at Vo = 95 V was operated at fsw > fr,1. Therefore, power switches were operated at ZVS turnon and rectifier diodes were operated at hard switching turn-off. Due to the fsw > fr,1 at the Vo = 95 V condition, the higher switching frequency reduced the magnetizing current loss compared to the Vo = 160 V condition (fsw < fr,1). Therefore, the circuit efficiency at the Vo = 95 V condition was better than Vo = 160 V. (e) v S1,g , v S1,d , and i S1 under V in = 760 V and P o = 200 W; and (f) v S1,g , v S1,d , and i S1 under V in = 760 V and P o = 1 kW. (e) v S1,g , v S1,d , and i S1 under V in = 760 V and P o = 200 W; and (f) v S1,g , v S1,d , and i S1 under V in = 760 V and P o = 1 kW. (e) (f) (e) v S1,g , v S1,d , and i S1 under V in = 760 V and P o = 200 W; and (f) v S1,g , v S1,d , and i S1 under V in = 760 V and P o = 1 kW. (e) (f)

Conclusions
A new DC resonant converter is presented and verified in this paper in DC nano-or micro-grid systems with a wide output voltage capability. The main contributions of the presented DC-DC converter are (1) a high input voltage circuit structure that uses a flying capacitor to balance two input capacitor voltages, (2) a wide output voltage operation using a control switch, and (3) a wide ZVS operation range for wide output voltage and load ranges. Since the square voltage waveforms are generated on vab and vbc, the voltage vCr1 + vCr2 = VCdc1 (if S1 and S3 are on) with d = Tsw/2 or VCdc2 (if S2 and S4 are on) with d = Tsw/2. Thus, the flying capacitors Cr1 and Cr2 can be used to balance input capacitor voltages VCdc1 = VCdc2 = Vin/2. To solve the high voltage application problems, a cascade resonant circuit is used on high voltage side to limit the voltage stress on active devices at Vin/2. Therefore, the general purpose power MOSFETs with a 600 V voltage rating can be used in the presented circuit. To realize wide output voltage requirements, such as for battery chargers for universal electric motorcycles or vehicles, the converter has two winding sets to extend the output voltage range using a control switch. In order to lessen the switching losses on active devices and increase the circuit efficiency, a series resonant converter using a LLC circuit structure was adopted for the presented converter to have a ZVS turn-on characteristic on active devices and a possible ZCS turn-off characteristic on fast recovery diodes. Therefore, the proposed converter can achieve soft switching operation from 200 to 1000 W load in the experimental test. Finally, the design procedures and experiments of the prototype circuit are provided to show the circuit performance and effectiveness of the studied converter. Further work on this circuit will consider reducing the number of circuit components and cost, as well as extending the output voltage range for universal battery charging station applications. Figure 11. Experiments of the presented converter under V o = 160 V: (a) v S1,g , v S2,g , v S3,g , and v S4,g under V in = 760 V and P o = 1 kW; (b) v S1,g , v ab , and v bc under V in = 760 V and P o = 1 kW; (c) v Cr1 , v Cr2 , and i Lr under V in = 760 V and P o = 1 kW; (d) I o , V o , i D1 , and i D4 under V in = 760 V and P o = 1 kW; (e) v S1,g , v S1,d , and i S1 under V in = 760 V and P o = 200 W; and (f) v S1,g , v S1,d , and i S1 under V in = 760 V and P o = 1 kW.

Conclusions
A new DC resonant converter is presented and verified in this paper in DC nano-o micro-grid systems with a wide output voltage capability. The main contributions of th presented DC-DC converter are (1) a high input voltage circuit structure that uses a flyin capacitor to balance two input capacitor voltages, (2) a wide output voltage operation us ing a control switch, and (3) a wide ZVS operation range for wide output voltage and load ranges. Since the square voltage waveforms are generated on vab and vbc, the voltage vCr1 vCr2 = VCdc1 (if S1 and S3 are on) with d = Tsw/2 or VCdc2 (if S2 and S4 are on) with d = Tsw/2. Thus the flying capacitors Cr1 and Cr2 can be used to balance input capacitor voltages VCdc1 = VCd = Vin/2. To solve the high voltage application problems, a cascade resonant circuit is used on high voltage side to limit the voltage stress on active devices at Vin/2. Therefore, th general purpose power MOSFETs with a 600 V voltage rating can be used in the presented circuit. To realize wide output voltage requirements, such as for battery chargers for uni versal electric motorcycles or vehicles, the converter has two winding sets to extend th output voltage range using a control switch. In order to lessen the switching losses on active devices and increase the circuit efficiency, a series resonant converter using a LLC circuit structure was adopted for the presented converter to have a ZVS turn-on charac teristic on active devices and a possible ZCS turn-off characteristic on fast recovery diodes Therefore, the proposed converter can achieve soft switching operation from 200 to 100 W load in the experimental test. Finally, the design procedures and experiments of th prototype circuit are provided to show the circuit performance and effectiveness of th studied converter. Further work on this circuit will consider reducing the number of cir cuit components and cost, as well as extending the output voltage range for universa battery charging station applications.

Conclusions
A new DC resonant converter is presented and verified in this paper in DC nano-or micro-grid systems with a wide output voltage capability. The main contributions of the presented DC-DC converter are (1) a high input voltage circuit structure that uses a flying capacitor to balance two input capacitor voltages, (2) a wide output voltage operation using a control switch, and (3) a wide ZVS operation range for wide output voltage and load ranges. Since the square voltage waveforms are generated on v ab and v bc , the voltage v Cr1 + v Cr2 = V Cdc1 (if S 1 and S 3 are on) with d = T sw /2 or V Cdc2 (if S 2 and S 4 are on) with d = T sw /2. Thus, the flying capacitors C r1 and C r2 can be used to balance input capacitor voltages V Cdc1 = V Cdc2 = V in /2. To solve the high voltage application problems, a cascade resonant circuit is used on high voltage side to limit the voltage stress on active devices at V in /2. Therefore, the general purpose power MOSFETs with a 600 V voltage rating can be used in the presented circuit. To realize wide output voltage requirements, such as for battery chargers for universal electric motorcycles or vehicles, the converter has two winding sets to extend the output voltage range using a control switch. In order to lessen the switching losses on active devices and increase the circuit efficiency, a series resonant converter using a LLC circuit structure was adopted for the presented converter to have a ZVS turn-on characteristic on active devices and a possible ZCS turn-off characteristic on fast recovery diodes. Therefore, the proposed converter can achieve soft switching operation from 200 to 1000 W load in the experimental test. Finally, the design procedures and experiments of the prototype circuit are provided to show the circuit performance and effectiveness of the studied converter. Further work on this circuit will consider reducing the number of circuit components and cost, as well as extending the output voltage range for universal battery charging station applications.