Hybrid LLC Converter with Wide Range of Zero-Voltage Switching and Wide Input Voltage Operation

: A new hybrid inductor-inductor-capacitor (LLC) converter is investigated to have wide voltage input operation capability and zero-voltage turn-on characteristics. The presented circuit topology can be applied for consumer power units without power factor correction or with long hold-up time requirement, photovoltaic energy conversion and renewable energy power transfer. To overcome the weakness of narrow voltage gain of resonant converter, the hybrid LLC converter with di ﬀ erent turns ratio of transformer is presented and the experimental investigation is provided to achieve wide voltage input capability (400 V–50 V). On the input-side, the converter can operate as full bridge resonant circuit or half bridge resonant circuit with input split capacitors for high or low voltage input region. On the output-side, the less or more winding turns is selected to overcome wide voltage input operation. According to the circuit structures and transformer turns ratio, the single stage LLC converter with wide voltage input operation capability (400 V–50 V) is accomplished. The laboratory prototype has been developed and the experimental waveforms are measured and demonstrated to investigate the e ﬀ ectiveness of the presented hybrid LLC converter.


Introduction
The renewable energy systems have been presented and established to reduce fossil fuel demand and prevent the influence of global warming.The output of renewable energy sources may be unstable voltage or current with dc or ac waveform.Power electronics play the principle of energy transfer between renewable energy sources and energy storage elements or loads.The output of photovoltaic (PV) panel or small-scale wind turbine generator is an unstable and non-constant dc voltage.Therefore, dc converters with wide voltage input capability are normally demanded to achieve the principle of energy transfer between PV panel (or small-scale wind generator) and energy storage unit or dc load.The power switches of dc-dc converters can be controlled by pulse-width modulation (PWM) [1][2][3] or frequency modulation (FM) [4][5][6].In PWM control, the switching frequency is fixed but duty cycle is variable in order to control load voltage.In FM control, the duty cycle is fixed and the switching frequency is variable to control voltage transfer function of dc converters and regulate load voltage.Gallium Nitride (GaN) FET, Insulated Gate Bipolar Transistors (IGBTs), Metal-Oxide-Semiconductor Field-Effect Transistor (MOSFET) and Silicon Carbide (SiC) devices are widely used in power electronics.The main drawback of IGBT devices is low frequency operation.SiC and GaN devices have advantages of high frequency operation and low switching loss for high power density applications.However, SiC and GaN devices have high cost problem compared to MOSFET and IGBT devices.In medium power applications, MOSFET devices have advantages of low cost and medium frequency operation capability.
High frequency and high efficiency power converters are demanded to achieve energy transfer from renewable energy sources to the energy storage units.Due to wide voltage variation of the solar panel output or low power wind generator, the soft switching converters with wide voltage input or output operation have studied.The series-connected dc-dc converter has studied in Reference [7] to accomplish wide input voltage operation for power conversion and battery charger applications.The disadvantage of the series-connected dc-dc converter is poor efficiency.The series/parallel-connected converters with wide voltage operation for vehicles were presented in References [8][9][10][11].The circuit topology needs more active and passive components so that the efficiency and reliability are decreased.Phase-shift PWM scheme with a wide voltage input/output operation were studied in References [12][13][14].The main problems of these topologies are the complicated control algorithm, high freewheeling current and hard switching operation at low load condition.Interleaved converters with frequency control have been presented in Reference [15] to have low input ripple current and achieve wide voltage input.This circuit topology is still a multistage circuit.The inductor-inductor-capacitor (LLC) converter with different winding turns was reported in Reference [16] to increase hold-up time on power units of personal computer applications.Four secondary windings are needed so that the conduction loss is increased on the transformer windings.In conventional LLC converter, the circuit can operate well at the limit voltage range to achieve low freewheeling current loss and high circuit efficiency.If the wide voltage range operation is demanded, the low inductor ratio between the magnetizing inductor and resonant inductor and low quality factor are needed in conventional LLC converter to achieve high voltage gain.However, the magnetizing inductance is decreased and the circulating current loss is increased.Therefore, the LLC converter efficiency is decreased.To solve this problem, a hybrid LLC circuit operation [17] shown in Figure 1a has been proposed to realize 4:1 wide voltage operation (V max = 4 V min ).When input voltage is in low voltage range, the full bridge LLC resonant circuit (S 1 -S 4 are active,) is operated.On the other hand, the half bridge LLC resonant circuit (S 1 and S 2 are active, S 3 OFF and S 4 ON) is used under high voltage input case.Therefore, wide input voltage operation is achieved in this circuit topology.However, there is a dc voltage level on the resonant capacitor when LLC circuit is operated at half bridge circuit structure.A hybrid LLC converter [18] has been studied to achieve 8:1 wide output voltage operation (V o,max = 8 V o,min ).This circuit topology is based on a full bridge resonant circuit and an ac switch on the primary-side and a hybrid diode rectifier on the output-side.Compares to Reference [17], the half bridge LLC circuit with split input capacitors is used in Reference [18] to avoid a dc voltage level on the resonant capacitor under low voltage output operation.In order to extend output voltage range, a hybrid output rectifier with a full-wave rectification or a half-wave rectification structure is adopted on the secondary-side in Reference [18].The main disadvantage of this circuit topology in Reference [18] is four diode conduction losses on the secondary-side when full-wave rectification structure is operated.
A new LLC converter is presented to realize wide range of input voltage operation.The LLC resonant circuit is used in the studied converter to achieve soft switching operation for all power devices due to inductive input impedance operation by pulse frequency modulation.To realize wide voltage input operation, the half bridge with split capacitors or full bridge equivalent LLC circuit can be operated on high-voltage side and the voltage doubler rectifier with different secondary turns is adopted on low-voltage side.Due to the different input voltage range, three circuit structures with different voltage gains are controlled to overcome wide range of input voltage variation.The input voltage range of the studied circuit topology is 50 V-400 V.For high voltage input (200 V-400 V), half bridge LLC circuit with fewer winding turns is used to obtain the least voltage gain and regulate load voltage.For medium voltage input (100 V-200 V), full bridge LLC circuit with fewer secondary turns is controlled to attain high voltage gain.For low voltage input (50 V-100 V), full bridge LLC circuit with more winding turns is adopted to obtain the largest voltage gain.Thus, the wide voltage operation (50 V-400 V) is realized in the studied LLC converter.Compared to the multistage converters or series/parallel converters, the proposed converter has simple circuit structure and control scheme by using the general analog integrated circuit.Compared to the 4:1 wide voltage LLC converter in Reference [17] with 4 switches (3 switches) conduction loss at low (high) voltage range, the proposed converter does not a dc voltage level on the resonant capacitor when half bridge LLC equivalent circuit structure with 4 switches conduction loss is operated and the proposed converter has 8:1 wide voltage operation capability.Compared to the 8:1 wide voltage LLC converter in Reference [18], the proposed converter has only two instead of four diode conduction losses on the secondary-side for high current output applications.The laboratory prototype has been developed and the experiments are given to investigate the presentation of the presented converter.

Circuit Configuration
The conventional LLC converters are provided in Figure 1. Figure 1a provides the converter schematic of the full bridge resonant circuit with center-tapped rectification and the voltage gain can be calculated as V o /V in = G ac (f sw )n s /n p , where G ac (f) is the transfer function of resonant tank by L m , L r and C r .The half bridge resonant circuit with center-tapped rectification is illustrated in Figure 1b.The voltage gain is calculated as V o /V in = G ac (f sw )n s /(2n p ).Therefore, the full bridge LLC converter has two times of gain of the half bridge resonant circuit.However, the rectifier diodes D 1 and D 2 has at least 2 V o voltage stress and more winding turns are needed on secondary-side.Figure 1c,d

Circuit Analysis and Principle of Operation
From the ON-OFF states of S ac1 and S ac2 in Figure 2b-d, the proposed LLC converter can be controlled under three voltage input regions: V in-H : 400 V-200 V (high voltage region), V in-M : 200 V-100 V (medium voltage region) and V in-L : 100 V-50 V (low voltage region).When V in is in the high voltage, the converter only needs low voltage gain to control load voltage.Therefore, S ac1 turns on and S ac2 , S 1 and S 2 turn off.The half bridge LLC resonant circuit and low secondary turns are used in Figure 2b.S 3 and S 4 are operated with frequency control.According to the resonant circuit by L r , C r , L m and R o , the voltage gain under high voltage input region is V o /V in-H = n s G ac (f sw )/n p , where G ac (f) is the transfer function by L r , C r , L m and R o .The LLC converter has four or six operating modes under f sw (switching frequency) > or < f r (resonant frequency).Since the converter is designed to have wide voltage gain, f sw is always less than or close to f r .Therefore, the zero-voltage switching (ZVS) of S 3 and S 4 and zero-current switching (ZCS) of D 3 and D 4 are accomplished.The PWM waveforms of LLC converter operated for high voltage region are illustrated in Figure 3a and the Figure 3b-g provide the equivalent circuits for modes 1-6 for one switching cycle.a.
Mode 1 [t 0 , t 1 ]: At t 0 , v CS4 = 0.The current i Lr flows through D S4 and S 4 turns on at this instant to have ZCS operation.The secondary current is positive so that D 3 is conducting and capacitor C o1 is charged.The components C r and L r are naturally resonant and the resonant frequency f r ≈ 1/2π Mode is conducting so that S 3 can turn on at this instant to accomplish ZVS.In mode 4, the resonant tank transfers power from input terminal V in to output terminal V o .e.
Mode      If 50 V < V in < 100 V (low voltage region, V in-L ), S ac1 turns off and S ac2 turns on.The LLC circuit and 2N s secondary turns are used in Figure 2d.For low voltage input region, the converter has gain V o /V in-L = 4G ac (f sw )n s /n p .It can observe the proposed converter under low voltage input region has the largest voltage gain compared to the high and medium voltage regions.Figure 5  Mode 6 [t 5 , T sw + t 0 ]: This mode starts at t 5 , when S 3 and S 2 turn off.i Lr (t 5 ) will discharge (charge) C S4 and C S1 and D 1 is forward biased.This mode ends at T sw + t 0 when v CS4 = v CS1 = 0.

Circuit Characteristics and Design Example
Based on the input voltage regions, S ac1 and S ac2 are controlled at on or off state.Therefore, the converter can be operated at three input voltage regions.Full bridge (half bridge) LLC converter is operated to have low (high) voltage gain for low (high) voltage input region.The equivalent resonant circuit is derived in Figure 6a.According to the PWM signals of S 1 -S 4 , the square wave voltage V ac is generated.If full bridge LLC converter with S 1 -S 4 is operated, the voltage V ac has voltage values ±V in in Figure 2c,d 6a, the input-side and output-side of the resonant tank are all square voltage waveforms.Based on the Fourier Series Analysis (FSA), the square voltage waveform is expressed as the summation of the fundamental frequency component and the higher order harmonic frequency components.Thus, the circuit analysis of the resonant circuit in Figure 6a is related to the nonlinear and nonsinusoidal circuit analysis.If the output voltage at higher order harmonic frequencies are less than the fundamental harmonic voltage, then only the fundamental harmonic of square voltage wave is considered in Figure 6b to simply the circuit analysis and design the resonant converter.The fundamental harmonic approximation (FHA) approach has acceptable design results if the LLC converter is operated at or close to the resonant frequency f r .In Figure 6b, the rms values of fundamental harmonic voltage at input and output side are given as.The ac equivalent load resistance R e,ac can be obtained according to power balance between praimary and secondary sides of trnasformer.
From Equations ( 2) and ( 3), the ac equivalent load resistance is calculated as.
From the linear circuit in Figure 6b, the voltage transfer function can be approximated by using the fundamental input and output voltages.
MOSFETs TK50J60U with 600 V/25 A ratings are used for power switches S 1 -S 4 and S ac1 .IXTP160N075T with 75 V/160 A ratings is used for power switch S ac2 .Power diodes MBR40100PT with 100 V/40 A ratings are adopted for the rectifier diodes D 1 -D 4 .The input and output capacitances are C in1 = C in2 = 300 µF and C o1 = C o2 = 540 µF.Two Schmitt voltage comparators are used to control PWM signals of S ac1 and S ac2 .The reference voltages of two comparators are designed at 200 V (between medium and high voltage input regions) and 100 V (between low and medium voltage input regions).Table 1 shows the specifications of the active and passive components in the studied LLC converter.

Experimental Verification of the Presented Converter
The hardware circuit of the presented circuit shown in Figure 7 is setup and the circuit specifications are shown in Table 1.The prototype is tested and the measured waveforms are provided to confirm the effectiveness of the presented resonant converter.The PWM signals of active switches at 50 V input (low voltage input region), 190 V input (medium voltage input region) and 400 V input (high voltage input region) and full load are shown in Figure 8a-c respectively.For V in = 50 V input (low voltage range), S ac1 is off, S ac2 is on and S 1 -S 4 are active with switching frequency f sw ≈ 65 kHz shown in Figure 8a.For V in = 190 V input (medium voltage range), S ac1 and S ac2 are off and only S 1 -S 4 are active with switching frequency f sw ≈ 102 kHz (Figure 8b).For V in = 400 V input (high voltage range), S ac1 is on and S 1 , S 2 and S ac2 are off.Only S 3 and S 4 are active with switching frequency f sw ≈ 108 kHz shown in Figure 8c.Figures 9 and 10 give the experimental waveforms of input-side and output-side, respectively, at V in = 50 V, 190 V and 400 V under full load.For 50 V input (low voltage range), S ac2 is on so that D 1 and D 2 are active and D 3 and D 4 are in the off state as shown in Figure 10a.For 190 V and 400 V input conditions, S ac2 is off so that D 1 and D 2 are reverse biased and D 3 and D 4 are active (Figure 10b,c).From the measured waveforms in Figure 10, the output capacitor voltages V Co1 and V Co2 are balanced well.Figure 11 gives the PWM signals of power switch S 1 or S 3 at 50 V, 190 V and 400 V input under 20% and 100% loads.It can observe the ZVS operation of S 1 and S 3 is realized for both 20% and 100% loads.Figure 12a shows the input voltage V in and the gating voltage v Sac2,g between V in = 50 V and 400 V.When V in ≥ or < 100 V, S ac2 is off (medium or high voltage range) or on (low voltage range).Figure 12b provides the input voltage V in and PWM voltages v Sac1 , v S1,g and v S2,g between V in = 0 V and 400 V.When V in ≥ 200 V (high voltage input region), S ac1 is turned on and S 1 and S 2 are turned off.The converter efficiencies for different input voltages and loads are given in Table 2.Under low voltage input region, the 2n s winding turns are used on the output-side.Therefore, the higher primary-side current is introduced and the conduction losses on switches and winding turns are increased compared to the n s winding turns used on the output-side under medium and high voltage input regions.Thus, the circuit efficiency at low voltage input region is less than the medium or high voltage input region under the same load.The measured switching frequency at different loads and input conditions are provided in Table 3.On each input voltage region, the higher input voltage results in the higher switching frequency under the same load.V in = 400 V & 100% load (high voltage range) 97

Conclusions
A new ZVS LLC converter is presented, analysis and realized to have wide load soft switching and wide voltage input operation.In order to accomplish wide load range of ZVS operation, the LLC resonant tank is used to accomplish ZVS turn-on for all active devices.According to the different input voltage ranges (50 V-400 V), the variable voltage gain is used to regulate load voltage constant.To realize the variable voltage gain which is related to the input voltage range (low, medium or high voltage range), two ac power switches are adopted in order to select half bridge or full bridge resonant circuit with the different turns-ratio of transformer.Thus, the wide range of input voltage operation is realized in the proposed circuit.The studied LLC converter can be applied to wide voltage input variation such as switching power converters with long hold-up time requirement, battery systems or solar panel.Compared to the other wide voltage circuit topologies, the proposed converter has no dc voltage level on the resonant converter [17] and less diode conduction loss [18].However, more power devices are still needed in the proposed converter to realize 8:1 wide voltage operation.Thus, the cost issue is the main drawback of the studied circuit for industry applications.The presented converter has been developed in the laboratory and its effectiveness was validated through experiments by a laboratory prototype circuit.
are the other two circuit configurations with voltage doubler rectification.The voltage gains of Figure 1c,d are obtained as V o /V in = G ac (f sw )(2n s )/n p and V o /V in = G ac (f sw )n s /n p , respectively.The voltage rating of D 1 and D 2 is V o and only one winding set of transformer is used on the output-side.Compared four circuit topologies in Figure 1, the full bridge LLC circuit with voltage double rectification has the largest voltage gain and the half bridge resonant circuit with center-tapped rectification has the least voltage gain.

Figure 1 .
Figure 1.Diagram of the conventional inductor-inductor-capacitor (LLC) converters (a) full bridge resonant circuit with center-tapped rectification (b) half bridge resonant circuit with center-tapped rectification (c) full bridge resonant circuit with voltage doubler rectification (d) half bridge resonant circuit with voltage doubler rectification.The proposed converter diagram is shown in Figure2ato have wide voltage input operation (50 V-400 V) such as solar power or renewable energy conversion.The proposed hybrid LLC converter includes a full bridge resonant circuit (S 1 -S 4 , L r , C r and L m ) and a half bridge resonant circuit (S 3 , S 4 , S ac1 , L r , C r and L m ) on the primary-side and the voltage doubler rectifier with N s and 2N s winding turns on the secondary-side.The converter has three different operating sub-circuits for three voltage input regions (400 V-200 V, 200 V-100 V and 100 V-50 V) as shown in Figure2b-d.If each equivalent

5 [t 4 , t 5 ]:
The diode current i D4 will naturally turn off at t 4 due to f r > f sw .In this mode, the load current I o discharges C o1 and C o2 .The current i Lm (t 5 ) is obtainedi Lm (t 5 ) ≈ −(n p V o )/(8n s L m f sw ) f.Mode 6 [t 5 , T sw + t 0 ]: S 3 turns off at the start of this mode.i Lr (t 5 ) discharge (charge) C S4 (C S3 ) and D 3 start conducting.To achieve zero-voltage switching of S 4 , |i Lm (t 5 )| must be greater than V in−H 2C oss /(L m + L r ).At the end of this mode, v CS4 = 0.

Figure 3 .
Figure 3. Presented converter under high voltage input region (a) pulse-width modulation (PWM) signals (b) mode 1 equivalent circuit (c) mode 2 equivalent circuit (d) mode 3 equivalent circuit (e) mode 4 equivalent circuit (f) mode 5 equivalent circuit (g) mode 6 equivalent circuit.When V in is in medium voltage region (V in-M : 100 V-200 V), S ac1 and S ac2 turn off.The full bridge LLC circuit and low secondary turns are used in Figure2c.The FM scheme is used to control S 1 -S 4 .The converter has gain V o /V in-M = 2G ac (f sw )n s /n p .The PWM signals and the corresponding circuits are illustrated in Figure4for medium voltage input condition. a
. If the half bridge LLC converter with S 3 and S 4 shown in Figure 2b is operated, the voltage V ac has voltage values ±V in /2.For medium and high voltage input regions shown in Figure 2b,c, S ac2 is off.It can obtain the inductor voltage V Lm = nV o /2 or -nV o /2, where n = n p /n s , if D 3 or D 4 is conducting.For low voltage input region (Figure 2d), The inductor voltage V Lm = nV o /4 or -nV o /4 if D 1 or D 2 is conducting.In Figure

Figure 6 .
Figure 6.The equivalent resonant circuit of the proposed LLC converter with (a) nonlinear nonsinusoidal circuit (b) linear sinusoidal circuit.

Figure 7 .
Figure 7. Presented LLC converter (a) picture of laboratory prototype (b) picture of the experiment setup.

Figure 8 .
Figure 8. PWM signals of switches at full load and (a) V in = 50 V (low voltage input region) (b) V in = 190 V (medium voltage input region) (c) V in = 400 V (high voltage input region).

Figure 9 .
Figure 9. Input ac voltage V ab , capacitor voltage v Cr and inductor current i Lr at full load and (a) V in = 50 V (low voltage input region) (b) V in = 190 V (medium voltage input region) (c) V in = 400 V (high voltage input region).

Figure 10 .
Figure 10.Output diode currents and capacitor voltages at full load and (a) V in = 50 V (low voltage input region) (b) V in = 190 V (medium voltage input region) (c) V in = 400 V (high voltage input region).

Figure 11 .
Figure 11.Measured PWM signals of power switch (a) S 1 at V in = 50 V (low voltage range) and 20% load (b) S 1 at V in = 50 V (low voltage range) and full load (c) S 1 at V in = 190 V (medium voltage range) and 20% load (d) S 1 at V in = 190 V (medium voltage range) and full load (e) S 3 at V in = 400 V (high voltage range) and 20% load (f) S 3 at V in = 400 V (high voltage range) and full load.

2 [t 1 , t 2 ]:
Since f r > f sw , i D3 is decreased to zero at t 1 with ZCS operation and i Lr (t 1 ) = i Lm (t 1 ).Thus, the load current I o will discharge C o1 and C o2 .The current i Lm (t 2 ) is obtained i Lm (t 2 ) ≈ (n p V o )/(8n s L m f sw ).
c. Mode 3 [t 2 , t 3 ]: Power device S 4 turns off at t 2 .i Lr (t 2 ) will discharge (charge) C S3 (C S4 ).Due to i Lr (t 2 ) < i Lm (t 2 ), D 4 will naturally conduct and capacitor voltage v Co2 is increased.d.Mode 4 [t 3 , t 4 ]: At the start of this mode, the capacitor voltage v CS3 = 0. i Lr (t 3 ) is positive and D S3

Mode 3 [t 2 , t 3 ]: At
The load current I o discharges C o2 and i D3 charges C o1 .t 2 , S 4 and S 1 turn off.i Lr (t 2 ) discharges C S3 and C S2 .Since i Lm (t 2 ) > i Lr (t 2 ), the diode D 4 is forward biased.To realize ZVS turn-on of S 3 and S 2 , the current i Lm (t 2 ) must be greater than V in,M C oss /(L m +L r ).
. Mode 1 [t 0 , t 1 ]: This mode starts at t 0 if the drain voltages v S4,d = v S1,d = 0. i Lr flows through D S4 and D S1 so that S 4 and S 1 can turn on with ZVS operation.Since i Lr (t 0 ) > i Lm (t 0 ), it can observe that D 3 conducts, v Lm ≈ (n p V o )/(2n s ).The resonant frequency by passive elements L r and C r in mode1 is f r = 1/[2π √ L r C r ].b.Mode 2 [t 1 , t 2 ]: If f sw > f r , D 3 will naturally turn off at t 1 .This mode ends at t 2 and i Lm (t 2 ) ≈ (n p V o )/(8n s L m f sw .c. d.

Mode 4 [t 3 , t 4 ]: At
t 3 , C S3 and C S2 are discharged to zero.i Lr (t 3 ) is positive so that D S2 and D s3 are conducting.Thus, the zero-voltage switching of S 3 and S 2 can be achieved after time t 3 .Owing to i Lm (t 3 ) > i Lr (t 3 ), D 4 conducts, v Lm ≈ -(V o n p )/(2n s ) and i Lm decreases.C o2 is charged.Since f r > f sw , D 4 will naturally turn off at t 4 .The current i Lm (t 5 ) is equal to −(n p V o )/(8n s L m f sw ).At t 5 , power devices S 3 and S 2 are turned off.Since i Lr (t 5 ) < 0, C S4 and C S1 discharge in this mode.To realize ZVS turn-on of S 4 and S 1 , the current |i Lm (t 5 )| must be greater than V in,M C oss /(L m + L r ).The mode 6 ends at time T sw + t 0 when v CS1 = v CS4 = 0.
f. Mode 6 [t 5 , T sw + t 0 ]: illustrates the PWM signals and the corresponding mode circuits.At time t 0 , v CS4 = v CS1 = 0. D S4 and D S1 turn on due to i Lr (t 0 ) < 0. The zero-voltage switching of S 4 and S 1 are achieved after time t 0 .i Lr (t 0 ) > i Lm (t 0 ) and D 1 is conducting.Mode 4 starts at t 3 when C S3 and C S2 are discharged to zero voltage.i Lr (t 3 ) is positive so that D S3 and D S2 are conducting.S 3 and S 2 turn on at this moment to have ZVS operation.If f r > f sw , D 2 will turn off with zero-current switching at t 4 .The load current I o discharges C o1 and C o2 .f.
b. Mode 2 [t 1 , t 2 ]: Since f r > f sw , D 1 will naturally turn off at t 1 and i Lr (t 1 ) = i Lm (t 1 ).The load current I o discharges C o1 and C o2 .C r , L m and L r are naturally resonant with frequency f p .c. Mode 3 [t 2 -t 3 ]: At t 2 , S 4 and S 1 turn off.i Lr (t 2 ) discharges C S3 and C S2 .Since i Lr (t 2 ) < i Lm (t 2 ), D 2 is forward biased and v Co2 is increased.d.Mode 4 [t 3 , t 4 ]: e. Mode 5 [t 4 , t 5 ]:

Table 1 .
Specifications of the presented converter.

Table 2 .
Measured efficiencies under input voltage and load conditions.

Table 3 .
Measured switching frequencies under input voltage and load conditions.