Design Methodology and Analysis of Five-Level LLC Resonant Converter for Battery Chargers

: This paper presents proposal of a ﬁve-level LLC resonant DC–DC converter design pro-cedure for battery chargers. The ﬁve-level inverter side of the proposed converter is connected to a transform-less LLC resonant tank to ensure operating at high frequency and achieve soft switching. The proposed converter has less weight, size, and cost. It is also much simpler in terms of implemen-tation, and has smooth energy conversion to the load. The proposed converter is designed to work within the range close to the resonant frequency, to ensure higher power density and efﬁciency. Thus, the range of operating frequency is set to be (91 kHz < fsw < 110 kHz), while the LLC parameters is designed to achieve resonant frequency fr = 100 kHz. Therefore, it is designed to achieve zero voltage switching (ZVS) for all switches, which enhances the efﬁciency as well. The theoretical analysis outcomes were conﬁrmed by simulation studies conducted using MATLAB/SIMULINK. An experimental model was also developed and validated with 100 VDC input voltage, which delivered output power of 100 W, 48 V, with efﬁciency around 96.9%. Selected ﬁndings are presented to conﬁrm the effectiveness of the suggested converter.


Introduction
With technological advancements in recent years, greenhouse gas emissions have increased, resulting in climate change and global warming. The energy technology sector is heavily reliant on fossil fuels and cannot cope with the current stringent emission requirement. As a result, green energy has become a central component of major economic policies and a major focus in international politics. Therefore, a converter is needed to convert the voltage before it can be effectively used [1]. Because of the negative effects of electricity generation from fossil sources, it is vital to switch to clean and renewable energy sources like solar, geothermal, wind, and hydropower energy [2]. The energy that reaches the Earth per hour from the Sun is close to the total energy consumed by humanity in a year [3]. As a result, utilizing this readily available resource should eventually reduce reliance on conventional fossil energy sources and contribute to decline in global warming, for a cleaner and safer world [2]. Energy storage systems (ESS) are mostly used to ensure the suitability of renewable energy generation in all circumstances.
Additional issues could also arise from the interconnection/linkage of renewable energy sources such as solar, wind, and fuel cells [4]. Scholars have developed highly efficient applications for energy conversion, owing to the recent improvements in power electronics technology. Power inverters are required for AC drives and other grid applications. Even though a traditional two-level inverter has been utilized in industry, it has significant power quality issues [5]. a 50 W/h supercapacitor. Constant voltage (CV), constant current (CC) and CC-CV mode algorithms are commonly used in fast-charging systems. The constant voltage applied to the battery is controlled using the CV technique. The CC mode differs from the CV mode in that it controls the constant current supplied to the battery [29]. Constant current (CC) and constant voltage (CV) charging strategies have been widely used to charge lithiumion batteries with great success. Battery chargers must be able to operate over a wide output voltage range based on the charging profile. LLC resonant converters are used in battery charger applications because of their soft switching and high frequency operation potentials [30].
A modulation technique for a full-bridge three-level LLC resonant converter was proposed in [15]. The researcher employed a variable duty cycle which could run at a specific frequency. A parameter called "master duty" was introduced for setting each voltage-level duty and for seamless handling of two-level, three-level, and mixed modes. LLC converter is a widely used topology among the numerous topologies of DC-DC converter because of its benefits, which include the ability of each main switch to achieve zero-voltage switching (ZVS) from zero to full load without the need for an auxiliary circuit, the ability of the rectifier diodes to achieve zero-current switching (ZCS), low electromagnetic interference (EMI) emissions, high efficiency, a wide operating range, and high power density capability [31]. The researcher also analyzed the wide-adjustable-range LLC resonant converter used in lithium-ion battery LIB charger of a plug-in hybrid EV system PHEV [31]. Different scholars [32][33][34][35][36] have also proposed a hybrid of LLC converter with a three-level conversion approach. MLIs are used to alleviate voltage stress experienced by switching devices; they are also used to increase the power range of LLC resonant converter as reported by [37,38]. Furthermore, researchers have proposed a technique that combines LLC converter with a five-level conversion topology [39].
As mentioned earlier, the conventional topologies for realizing resonant converters are half-bridge and full-bridge, and multilevel inverters are not favorable for high-frequency resonant inverters since modelling them for control design is difficult, particularly when using high-frequency switching modulation approaches. To eliminate the drawbacks associated with half-bridge [6], full-bridge [40], phase-shifted full-bridge [41], and threelevel [36] inverters integrated with resonant tank network, a five level multilevel inverter is introduced in this paper. This paper presents a five-level cascaded multilevel non-isolated LLC resonant tank DC-DC converter for battery charger application for electrical vehicle. This converter integrates the advantages of the LLC resonant converter, non-isolation, and the five-level conversion technique. The resonant network is fed by a high-frequency converter working at a proper operating frequency by applying frequency modulation control. Furthermore, the transformer-less approach adopted in this work minimizes the system's cost and weight. The validity of the proposed converter has been verified, supported by theoretical analysis, simulation results, and an experimental prototype. The design is described in detail in the next section. The selected operation range, gain, and ZVS achievement had been carefully estimated to ensure the converter able to operate with a range close to the resonant frequency, to achieve ZVS for all switches, high power density, and high efficiency. The rest of this article is presented as follows: the proposed circuit layout of the converter and its essential characteristics are described in Section 2, while Section 3 presents the explanation of the various operating modes. Section 4 explains the design consideration of the equivalent circuit, while Section 5 presents the simulation results. Section 6 presents the results of the experiments using a 100 W scaled laboratory prototype, while Section 7 concludes this paper. Finally, future challenges are introduced in Section 8. composed of an inductor (L r ), a capacitance (C r ), an inductance (L m ) in parallel, and a fullwave rectifier with a filter capacitor and a battery at the converter's output side, respectively. The switch pairs S1, S2 and S3, S4 operate in a complementary fashion and the same is true for S5, S6, S7, S8. In contrast to conventional single-stage converters, the voltage delivered to the LLC stage contains five discrete levels (see Figure 2). The key steady-state waveforms of the converter are shown in Figure 3. Summing Vin1 = Vin2 = Vdc condition, the five-level inverter can generate five output voltage (VAB) levels: +Vdc, +2Vdc, 0, −Vdc, and −2Vdc, as shown in the equivalent circuits presented in Figure 4. This paper focuses on the issue of charging the battery by CV method by MLI integrated with resonant converter and a transformer-less. The operation modes of the proposed approach are outlined in the next section. Figure 1 illustrates the complete configuration of the five-level induc capacitor (LLC) resonant DC-DC converter. The circuit comprises of two s current DC sources: each source is linked to H bridge and eight IGBTs. The composed of an inductor (Lr), a capacitance (Cr), an inductance (Lm) in paral wave rectifier with a filter capacitor and a battery at the converter's respectively. The switch pairs S1, S2 and S3, S4 operate in a complementar the same is true for S5, S6, S7, S8. In contrast to conventional single-stage c voltage delivered to the LLC stage contains five discrete levels (see Figur steady-state waveforms of the converter are shown in Figure 3. Summing Vdc condition, the five-level inverter can generate five output voltage (VAB +2Vdc, 0, −Vdc, and −2Vdc, as shown in the equivalent circuits presented in paper focuses on the issue of charging the battery by CV method by MLI in resonant converter and a transformer-less. The operation modes of the propo are outlined in the next section.   respectively. The switch pairs S1, S2 and S3, S4 operate in a the same is true for S5, S6, S7, S8. In contrast to conventional voltage delivered to the LLC stage contains five discrete lev steady-state waveforms of the converter are shown in Figure Vdc condition, the five-level inverter can generate five output +2Vdc, 0, −Vdc, and −2Vdc, as shown in the equivalent circuit paper focuses on the issue of charging the battery by CV met resonant converter and a transformer-less. The operation mod are outlined in the next section.

Modes of Operation
Based on Figure 4, the operation of the converter can be explained utilizing the transition states, which consists of five modes, as will be discussed in the following subsections. The different operational modes of the proposed converter are presented in Figure 4.

Mode1 (t0 < t < t1)
In this mode, switches in both bridges act in complementary approach; switches S1 and S4 in the first bridge are turned ON, whereas S5 and S8 in the second bridge are OFF. At this time, VAB reaches the maximum positive voltage +2Vdc, and the resonant capacitor decreases and starts to change its polarity (zero crossing) from positive to negative polarity, while the resonant current, iLr, stops decreasing. The rectifier diodes D1 and D4 are conducted. Figure 4a depicts this mode of operation.

Mode2 (t1 < t < t2)
The first step of this mode is to turn OFF switch S5, while leaving switches S1, S4 and S8 ON. In this mode, the current completes its path through freewheeling diode of S6, VAB starts to drop, and the voltage across LLC is half of the input voltage (VAB = +Vdc). The polarity of iLr begins to shift from negative to positive polarity (zero crossing). The energy stored in the resonant tank is transferred to the battery through the rectifier diodes D1 and D4. Figure 4b shows this mode of operation.

Mode3 (t2 < t < t3)
In this mode, S1 and S4 are turned OFF, while S2, S4, S6, and S8 are turned ON. VAB drops to zero, and the voltage across LLC is zero (VAB = 0) in this mode. The resonant capacitor voltage Vcr stops decreasing, and starts to rise in order to change polarity, whereas the resonant current iLr increases. Rectifier Diodes D2 and D3 are conducted and their current is increasing. Figure 4c represents this mode of operation.

Mode4 (t3 < t < t4)
In the previous stage, the positive half cycle is completed. In this mode, switches S2, S3, and S6 are switched ON and the freewheeling diode of S8 is conducted (see Figure 4d). VAB has negative polarity and the voltage across LLC is half the negative input voltage (VAB = −Vdc) in this mode. iLr reaches the maximum value and stops increasing, while resonant capacitor voltage Vcr reaches the positive polarity. Rectifier diode D2 and D3 are still conducted and their current is at the maximum value, as the energy stored in the resonant tank is transferred to the battery through them.

Modes of Operation
Based on Figure 4, the operation of the converter can be explained utilizing the transition states, which consists of five modes, as will be discussed in the following subsections. The different operational modes of the proposed converter are presented in Figure 4.

Mode1 (t0 < t < t1)
In this mode, switches in both bridges act in complementary approach; switches S1 and S4 in the first bridge are turned ON, whereas S5 and S8 in the second bridge are OFF. At this time, VAB reaches the maximum positive voltage +2Vdc, and the resonant capacitor decreases and starts to change its polarity (zero crossing) from positive to negative polarity, while the resonant current, iL r , stops decreasing. The rectifier diodes D1 and D4 are conducted. Figure 4a depicts this mode of operation.

Mode2 (t1 < t < t2)
The first step of this mode is to turn OFF switch S5, while leaving switches S1, S4 and S8 ON. In this mode, the current completes its path through freewheeling diode of S6, VAB starts to drop, and the voltage across LLC is half of the input voltage (VAB = +Vdc). The polarity of iL r begins to shift from negative to positive polarity (zero crossing). The energy stored in the resonant tank is transferred to the battery through the rectifier diodes D1 and D4. Figure 4b shows this mode of operation.

Mode3 (t2 < t < t3)
In this mode, S1 and S4 are turned OFF, while S2, S4, S6, and S8 are turned ON. VAB drops to zero, and the voltage across LLC is zero (VAB = 0) in this mode. The resonant capacitor voltage Vcr stops decreasing, and starts to rise in order to change polarity, whereas the resonant current iL r increases. Rectifier Diodes D2 and D3 are conducted and their current is increasing. Figure 4c represents this mode of operation.

Mode4 (t3 < t < t4)
In the previous stage, the positive half cycle is completed. In this mode, switches S2, S3, and S6 are switched ON and the freewheeling diode of S8 is conducted (see Figure 4d). VAB has negative polarity and the voltage across LLC is half the negative input voltage (VAB = −Vdc) in this mode. iL r reaches the maximum value and stops increasing, while resonant capacitor voltage Vcr reaches the positive polarity. Rectifier diode D2 and D3 are still conducted and their current is at the maximum value, as the energy stored in the resonant tank is transferred to the battery through them. Figure 4e shows the mode between t4 and t5. In this mode, switches S2, S3, and S6 are still ON, and S7 will operate. Resonant capacitor voltage Vcr increases, and on the way, reaches its highest point (fully charged), whereas the resonant inductor current iL r begins to reduce. VAB drops to negative 2Vdc and this voltage goes to the LLC resonant tank. At the end of the previous stage, the rectifier diodes D1 and D4 are still switched OFF, thus, the energy stored in the resonant tank is transferred through rectifier diodes D2 and D3.

Design Consideration
The best efficiency may be reached by LLC resonant converters when the resonant frequency matches the switching frequency. The design strategy for resonant elements is centered on obtaining the optimal operating point, which can be accomplished in the same manner as with a typical LLC converter, as detailed in various publications [42,43]. The parameters of the proposed converter are explained in this section based on the specifications presented in Table 1. Because the resonant tank acts as a bandpass filter, it is typical to assume that the power transmission is limited to the core components of the voltages and currents in the resonant tank. This is referred to as the first harmonic approximation (FHA). The voltage gain (M gain ) of the LLC stage can be calculated using the analogous circuit depiction in Figure 5.
where L n is the ratio of the magnetizing inductance L m to the resonant inductance L r , defined as follows: are still ON, and S7 will operate. Resonant capacitor voltage Vcr increases, and way, reaches its highest point (fully charged), whereas the resonant inductor curr begins to reduce. VAB drops to negative 2Vdc and this voltage goes to the LLC re tank. At the end of the previous stage, the rectifier diodes D1 and D4 are still sw OFF, thus, the energy stored in the resonant tank is transferred through rectifier D2 and D3.

Design Consideration
The best efficiency may be reached by LLC resonant converters when the re frequency matches the switching frequency. The design strategy for resonant elem centered on obtaining the optimal operating point, which can be accomplished in th manner as with a typical LLC converter, as detailed in various publications [42,43 parameters of the proposed converter are explained in this section based o specifications presented in Table 1. Because the resonant tank acts as a bandpass filter, it is typical to assume th power transmission is limited to the core components of the voltages and currents resonant tank. This is referred to as the first harmonic approximation (FHA). The v gain (Mgain) of the LLC stage can be calculated using the analogous circuit depic Figure 5.
where Ln is the ratio of the magnetizing inductance Lm to the resonant inducta defined as follows: The definitions of the first and second resonant frequencies are as follows: The definitions of the first and second resonant frequencies are as follows: where L r = resonant inductance, C r = resonant capacitor, and L m = magnetizing inductance. The ratio of the switching frequency to the resonant frequency is the normalized switching frequency and is defined as follows: Q is the quality factor, and introduced to illustrate different load conditions. Q is defined to be the ratio between characteristic impedance and the load resistance, which is calculated as follows: where R ac is equivalent to the load and rectifier stage. V bat and P bat denote the output voltage and power, respectively. Additionally, the output voltage is clamped by the battery and is considered the same while the charging is carried out.
The value F n determines the switching frequency in relation to the higher resonance frequency of the resonant tank f r1 . The resonant tank's typical impedance is Z o . Variables L n and Q are incorporated to make (1) independent of the actual L m , L r , and C r values. P bat represents the transferred power to the battery. The frequency fr2, at which the voltage gain M gain is maximum, is the lower resonance frequency f r2 . As shown in (1), it is used to plot the normalized dc gain against normalized frequency for various values of the quality factor (Q), as seen in Figure 6a, which depicts the voltage gain under various loading situations (changes in Q value) and varying designs (changes in L n ) as in Figure 6b at different relative switching frequencies.
The relationship between voltage gain, load, resonant and switching frequency frequencies is depicted by the curve. The optimum normalized gain value for the given inductor ratio is represented as a minimum and maximum gain value (dash lines).
Based on the operating points in Figure 6, the values of L n and Q are chosen and used to compute the L r and C r values. The gain is calculated using the lower and maximum switching frequency values to check the range of the selected switching frequency. Q value must be 0.39 to achieve the necessary output voltage of 48 Vdc at a 100 kHz resonant frequency.

Calculation of Components in Resonant Tank Circuit
To construct the resonant tank circuit, Equations (2) and (6) Following that, the value of the resonant inductor, Lr, can be estimated using: Finally, using the normalised inductance, Ln, the value of the magnetising inductor, Lm, can be calculated as follows :

Simulation Results
To validate the suggested converter, a lab model with 60-140 V input voltage and a 48 V/100 W output voltage was built for simulation. Table 1 contains the detailed design requirements. The proposed five-level LLC series-parallel resonant DC to DC converter battery charging application was simulated using MATLAB/SIMULINK to ensure that it meets the parameters specified in Tables 1 and 2.

Calculation of Components in Resonant Tank Circuit
To construct the resonant tank circuit, Equations (2) and (6) are used to calculate the inductance ratio L n and quality factor Q values. Plotting the voltage gain equation of the LLC resonant high-voltage DC-DC converter for multiple L n and Q values, as illustrated in Figure 6a,b, is the simplest way to choose L n and Q values. The graph aids in finding the L n and Q values that will meet the converter's gain requirement. As a result, the chosen values for L n and Q are 3 and 0.39, respectively, depending on the desired gain. Having chosen the L n and Q values, the sizes of LLC resonant tank components are then determined. To begin, the resonant capacitor C r can be obtained using the formula: Following that, the value of the resonant inductor, L r , can be estimated using: Finally, using the normalised inductance, L n , the value of the magnetising inductor, L m , can be calculated as follows:

Simulation Results
To validate the suggested converter, a lab model with 60-140 V input voltage and a 48 V/100 W output voltage was built for simulation. Table 1 contains the detailed design requirements. The proposed five-level LLC series-parallel resonant DC to DC converter battery charging application was simulated using MATLAB/SIMULINK to ensure that it meets the parameters specified in Tables 1 and 2. The simulated waveform results are clearly depicted in Figures 7-9. Figure 7 shows the operation of the converter at the switching frequency range mentioned in Table 1; the values of the output inverter, resonant capacitor voltage, and resonant current were set to 100 V, 40 V, and 5 A, respectively. Since the resonant current is lagging behind the tank voltage, both converter power switches were operated with ZVS.  Figure 7 the operation of the converter at the switching frequency range mentioned in Tab values of the output inverter, resonant capacitor voltage, and resonant current we 100 V, 40 V, and 5 A, respectively. Since the resonant current is lagging behind t voltage, both converter power switches were operated with ZVS. Figure 8 shows the simulation plots of the gate voltage VGE and collector VCE. The gate voltage, VGE, and VCE were simulated for switches S3, S4, S7, an the inverter to verify the occurrence of ZVS, as shown in Figure 8. It was observ the VCE had reached zero before all the switches started experiencing increase in t voltage, indicating that all the switches were switched ON at ZVS.    Figure 7 shows the operation of the converter at the switching frequency range mentioned in Table 1; the values of the output inverter, resonant capacitor voltage, and resonant current were set to 100 V, 40 V, and 5 A, respectively. Since the resonant current is lagging behind the tank voltage, both converter power switches were operated with ZVS. Figure 8 shows the simulation plots of the gate voltage VGE and collector voltage VCE. The gate voltage, VGE, and VCE were simulated for switches S3, S4, S7, and S8 of the inverter to verify the occurrence of ZVS, as shown in Figure 8. It was observed that the VCE had reached zero before all the switches started experiencing increase in the gate voltage, indicating that all the switches were switched ON at ZVS.  state of charge was purposely increased, starting from the initial value, which was 55%, until it reached stable state. With the use of the constant voltage charging method, the voltage started to increase from zero. When it reached a constant value, the current began to drop and reached a negative value. By employing a soft-switching technique, both low-and high-frequency current ripples were reduced on the battery, thus extending the battery life without compromising the charger's size and reducing switching loss.

Experimental Results
A prototype of the cascaded H-bridge multilevel inverter LLC resonant converter has been built to verify the theoretical and simulation outcomes, which is based on the specifications given in Table 1. Figure 10 shows the prototype, which has two bridges, the resonant branch, the output filter, the gate driver, and the DSP controller. The DSP control is realized using the TMS320F28379D chip, which is capable of high-speed processing of decimal data with a processing capability of 200 MHz. Table 2 shows the essential parameters coming from the proposed design approach, as well as the actual measured values. Table 3 lists the semiconductor device and major circuit components of the converter. Figure 11 shows the experimental waveforms of the voltage across the resonant tank VAB, which is a five-level voltage. The voltage across the resonant capacitor (VCr), and the resonant inductor current iLr at Vin = 100. The resonant current and the voltage across the resonant capacitor are almost inverted in this diagram, as in Figure 11. It was evident that the resonant current began with a negative polarity in each half cycle, which allowed for the achievement of the ZVS for switching frequency at the resonant frequency (100 KHz).  Figure 8 shows the simulation plots of the gate voltage VGE and collector voltage VCE. The gate voltage, VGE, and VCE were simulated for switches S3, S4, S7, and S8 of the inverter to verify the occurrence of ZVS, as shown in Figure 8. It was observed that the VCE had reached zero before all the switches started experiencing increase in the gate voltage, indicating that all the switches were switched ON at ZVS.
The output results of the battery side are presented in Figure 9. The battery output waveform included state of charge SOC%, current Ibat, and voltage Vbat waveforms. The state of charge was purposely increased, starting from the initial value, which was 55%, until it reached stable state. With the use of the constant voltage charging method, the voltage started to increase from zero. When it reached a constant value, the current began to drop and reached a negative value.
By employing a soft-switching technique, both low-and high-frequency current ripples were reduced on the battery, thus extending the battery life without compromising the charger's size and reducing switching loss.

Experimental Results
A prototype of the cascaded H-bridge multilevel inverter LLC resonant converter has been built to verify the theoretical and simulation outcomes, which is based on the specifications given in Table 1. Figure 10 shows the prototype, which has two bridges, the resonant branch, the output filter, the gate driver, and the DSP controller. The DSP control is realized using the TMS320F28379D chip, which is capable of high-speed processing of decimal data with a processing capability of 200 MHz. Table 2 shows the essential parameters coming from the proposed design approach, as well as the actual measured values. Table 3 lists the semiconductor device and major circuit components of the converter. The obtained experimental output voltage resonant capacitor and current resonant were approximately 40 V and 5 A, respectively. Furthermore, it is worth noting that these findings are consistent with the simulation results depicted in Figure 7.   The experimental voltage waveforms from collector to emitter, VCEs1, VCEs2, VCEs5, and VCEs6, and gate signal voltages, VGS1, VGS2, VGS5, and VGS6, for the output load are shown in Figure 12. The collector and gate voltages for each switch show that the VCE was zero before switching ON the switches, which allowed the turning ON of the power switches in the ZVS mode. Hence, all the switches were turned ON at ZVS for the load selected as in Figure 12, which is consistent with the simulation results reported in Figure 8. Finally, Figure 13 shows the regulated output DC voltage; the value of the   Figure 11 shows the experimental waveforms of the voltage across the resonant tank VAB, which is a five-level voltage. The voltage across the resonant capacitor (V Cr ), and the resonant inductor current iL r at Vin = 100. The resonant current and the voltage across the resonant capacitor are almost inverted in this diagram, as in Figure 11. It was evident that the resonant current began with a negative polarity in each half cycle, which allowed for the achievement of the ZVS for switching frequency at the resonant frequency (100 KHz). The obtained experimental output voltage resonant capacitor and current resonant were approximately 40 V and 5 A, respectively. Furthermore, it is worth noting that these findings are consistent with the simulation results depicted in Figure 7. The obtained experimental output voltage resonant capacitor and current resonant were approximately 40 V and 5 A, respectively. Furthermore, it is worth noting that these findings are consistent with the simulation results depicted in Figure 7.   Figure 11. Experimental waveforms of voltage across multilevel inverter VAB, resonant capacitor, VCr and resonant current, iLr, for the output load resistance of 23 Ω.
The experimental voltage waveforms from collector to emitter, VCEs1, VCEs2, VCEs5, and VCEs6, and gate signal voltages, VGS1, VGS2, VGS5, and VGS6, for the output load are shown in Figure 12. The collector and gate voltages for each switch show that the VCE was zero before switching ON the switches, which allowed the turning ON of the The experimental voltage waveforms from collector to emitter, VCEs1, VCEs2, VCEs5, and VCEs6, and gate signal voltages, VGS1, VGS2, VGS5, and VGS6, for the output load are shown in Figure 12. The collector and gate voltages for each switch show that the VCE was zero before switching ON the switches, which allowed the turning ON of the power switches in the ZVS mode. Hence, all the switches were turned ON at ZVS for the load selected as in Figure 12, which is consistent with the simulation results reported in Figure 8. Finally, Figure 13 shows the regulated output DC voltage; the value of the measured voltage was in agreement with the simulated value as in Figure 8, which is 48 V.   Table 4 shows the comparison between the conducted work and related LLC topologies for battery charger as reviewed in the literature. Based on the comparison results, the proposed converter requires less magnetic components, and less input voltage compared to previous converters reported in [6,[29][30][31]34]. Furthermore, the design consideration and selected operation range proved that the proposed configuration has significantly enhanced efficiency compared to [29,30,34]. Even though the proposed converter has a higher number of active switches (8 switches), it has been proven to have better efficiency, less weight, and less cost due to proper design consideration, parameters, and operation range selection.    Table 4 shows the comparison between the conducted topologies for battery charger as reviewed in the literature. B results, the proposed converter requires less magnetic componen compared to previous converters reported in [6,[29][30][31]34]. F consideration and selected operation range proved that the pro significantly enhanced efficiency compared to [29,30,34]. Eve converter has a higher number of active switches (8 switches), it better efficiency, less weight, and less cost due to prope parameters, and operation range selection.  Table 4 shows the comparison between the conducted work and related LLC topologies for battery charger as reviewed in the literature. Based on the comparison results, the proposed converter requires less magnetic components, and less input voltage compared to previous converters reported in [6,[29][30][31]34]. Furthermore, the design consideration and selected operation range proved that the proposed configuration has significantly enhanced efficiency compared to [29,30,34]. Even though the proposed converter has a higher number of active switches (8 switches), it has been proven to have better efficiency, less weight, and less cost due to proper design consideration, parameters, and operation range selection.

Conclusions
A five-level cascaded H-bridge LLC resonant converter for battery charger has been presented. The proposed converter is designed with proper operation region selection to obtain high power density, high efficiency, and less magnetic components to ensure the reduction on factors of size and cost. The range of operating frequency is set to be (91 kHz < fsw < 110 kHz), and LLC parameters are designed to achieve resonant frequency fr = 100 kHz to simplify the design considerations. The proposed converter performance had been compared to related works in literature, and evaluated by utilizing simulation (MATLAB/SIMULINK). An experimental prototype had been tested to achieve output power 100 W, 48 V, with efficiency around 96.9%. Thus, the evaluation of results and discussions proves that the theoretical, simulation, and experimental results are in agreement, and ensure the validity of this work.

Future Challenges
The major challenge in using the proposed converter is as a bidirectional DC-DC converter mode. This is because the diodes (full-bridge rectifier) used in proposed circuit are unidirectional and uncontrolled rectifiers. Second, if the diodes are replaced with active MOSFETs or IGBTs, then additional control strategies will be required in order to synchronously control the multilevel-inverter, resonant circuit, and the battery charger and discharger stages. By overcoming these limitations, a bidirectional behavior can be achieved by using the proposed structural, which will be the subject of future work and challenges in charger and discharger applications.