Hybrid DC-DC Converter with Low Switching Loss, Low Primary Current and Wide Voltage Operation

: A full-bridge converter with an additional resonant circuit and variable secondary turns is presented and achieved to have soft-switching operation on active devices, wide voltage input operation and low freewheeling current loss. The resonant tank is linked to the lagging-leg of the full bridge pulse-width modulation converter to realize zero-voltage switching (ZVS) characteristic on the power switches. Therefore, the wide ZVS operation can be accomplished in the presented circuit over the whole input voltage range and output load. To overcome the wide voltage variation on renewable energy applications such as DC wind power and solar power conversion, two winding sets are used on the output-side of the proposed converter to obtain the different voltage gains. Therefore, the wide voltage input from 90 to 450 V ( V in,max = 5 V in,min ) is implemented in the presented circuit. To further improve the freewheeling current loss issue in the conventional phase-shift pulse-width modulation converter, an auxiliary DC voltage generated from the resonant circuit is adopted to reduce this freewheeling current loss. Compared to the multi-stage DC converters with wide input voltage range operation, the proposed circuit has a low freewheeling current loss, low switching loss and a simple control algorithm. The studied circuit is tested and the experimental results are demonstrated to testify the performance of the resented circuit.


Introduction
For the elimination of the air pollution and emission of greenhouse gases, clean renewable energy sources with power electronic techniques have been investigated and developed for fuel cell [1,2], wind power [3,4], solar power [5,6] and battery storage system [7] applications. Solar power [8,9] is one of the most attractive clean energy sources. However, the problem of the solar panel output voltage is unstable. To solve this problem, DC-DC pulse-width modulation (PWM) converters with wide voltage variation capability were studied and researched. Full-bridge phase-shift pulse-width modulation (PSPWM) converter have been developed in [10] to have 4:1 (18 V-75 V) input voltage range operation. The synchronous rectifiers with the PSPWM scheme are adopted on the secondary side to control load voltage. The disadvantages of this circuit topology are a complicated control scheme and eight active switches. The other problems of this circuit topology are the output power and zero voltage switching (ZVS) range on active devices are related to the input voltage. In [11], the halfbridge PWM converter with multi-winding on the secondary side is proposed to control load voltage and have 2:1 (V in = 250 V-400 V) voltage operation. Three active devices, four rectified diodes, four secondary windings and two filter inductors are adopted in this converter. The more component counts used in this converter will result in high cost and circuit reliability. In [12], a cascade DC-DC converter with PSPWM modulation is proposed to achieve wide input voltage operation (V in = 600 V-800 V). However, the voltage range of this circuit topology is limited between 600 V-800 V. The conventional three-level duty cycle control or frequency control converters can achieve the same voltage range operation. In [13], an asymmetric half-bridge resonant circuit with the buck-boost circuit and resonant circuit has been presented to have 2:1 (V in = 36 V-72 V) voltage operation. The main problem of this circuit topology is the unbiased voltage stresses on active switches for both primary and secondary sides. In [14], a hybrid converter with a half-bridge PWM circuit and boost circuit was presented to achieve about 2:1 (V in = 45 V-75 V) input voltage operation. The basic structure of this circuit topology is a kind of series-connected converter. In [15], the resonant converter operated at the half bridge or full bridge resonant circuit has been studied to realize V in = 80 V-200 V voltage operation. However, the resonant frequency at low and high input voltage ranges are different due to the different resonant capacitances on half-bridge and full-bridge resonant tanks. Therefore, the wide switching frequency range will happen in this circuit topology. In [16], the PSPWM converter with two isolation transformers has been developed to have 4:1 input voltage operation. This circuit has four different operating sub-circuits under different input voltage conditions. However, the control algorithm of this circuit topology is too complicated to be implemented. In [17], the input-parallel output-series converter has been studied to have wide voltage range capability. Eight active devices and eight rectifier diodes are needed to have a 4:1 voltage operation. This circuit topology uses more power semiconductors that will reduce the circuit reliability and increase the cost.
A hybrid soft-switching DC-DC PWM converter is presented to reduce switching losses on active devices, achieve low primary current at the freewheeling state and have 5:1 (V in = 90 V-450 V) wide input voltage operation. The PSPWM modulation is used to control active devices and have zero voltage turn-on characteristic. The presented converter contains a resonant circuit and a full-bridge PWM circuit. The resonant circuit can extend the ZVS operation range and also reduce the primary freewheeling current. To extend the input voltage operation range, two secondary winding sets are used on the low voltage side. Thus, a 5:1 (V in = 90 V-450 V) wide input voltage range can be realized in the studied hybrid circuit. Compared to the conventional PWM converters, the advantages of the presented converter are low switching loss, low primary freewheeling current, wide input voltage operation and a simple control scheme. The circuit diagram of the proposed PWM circuit is discussed in Section 2. The principles of operation are provided in Section 3. In Sections 4 and 5, the circuit characteristic and experiments of the prototype circuit are discussed and presented to show the circuit performance. In Section 6, the conclusions of the studied hybrid PWM circuit are presented.

Circuit Diagram of the Proposed PWM Converter
The conventional PSPWM converter and main PWM signals are given in Figure 1a. Six power semiconductors (four active devices and two rectifier diodes), two magnetic cores (one isolation transformer and one filter inductor) and one filter capacitor are normally used in this circuit topology to realize medium or high power applications. The disadvantages of the PSPWM converter are the hard switching operation of active devices on lagging leg and high freewheeling current. The serious switching loss on active devices will result in serious switching losses at high-frequency operation and a high freewheeling current will reduce circuit efficiency. To solve these two problems, a half-bridge inductor-inductor-capacitor (LLC) converter can be added to a conventional full-bridge PSPWM converter as shown in Figure 1b remark in purple. The circuit elements of LLC converter include S 3 , S 4 , L r , C r , T 2 , D 3 , D 4 , C o,r and D r . The PSPWM converter and resonant circuit share the same active devices S 3 and S 4 . Due to the ZVS operation characteristic of LLC converter, active devices S 3 and S 4 can achieve ZVS turn-on operation with a wide load range. Therefore, the hard switching drawback is improved. Diode D r in Figure 1b is used to connect two dc voltage terminals V o,r (output terminal of LLC converter) and V R (the secondary rectified voltage). During power transfer interval (|v ab |>0), V R > V o,r and D r is reverse biased. However, D r will be forward biased at the freewheeling duration (v ab = 0 under S 1 and S 3 ON or S 2 and S 4 ON). Due to D r is conducting, the power flow at the freewheeling duration is from V o,r (LLC converter) to V o (load side) and the rectified voltage V R = V o,r . Under the freewheeling state, the primary leg voltage v ab = 0 and the secondary rectified voltage V R = V o,r . It can obtain the primary-side inductor voltage v LP = -n p1 V o,r /n s1 < 0 and the primary current i LP will be declined to zero. Hence, the high circulating current drawback is overcome. In order to overcome and achieve wide voltage operation, four secondary windings are adopted in Figure 1b remark in blue. For low voltage input conditions (V in,min~2 .2V in,min ), Q 1 turns on and Q 2 turns off ( Figure 2a). Thus, D 2 and D 3 are reverse biased. The proposed circuit has a high voltage gain with transformer turns ratio N T1 = n p1 /(n s1 +n s2 ). Under high voltage input case (2.2V in,min~5 V in,min ), the switches Q 1 turns off and Q 2 turns on ( Figure 2b). Therefore, D 1 and D 4 are reverse biased and the present circuit has low voltage gain with transformer turns ratio N T1 = n p1 /n s2 . Hence, the wide voltage operation, low freewheeling current and wide ZVS operation are realized in the presented hybrid PWM converter. freewheeling duration is from Vo,r (LLC converter) to Vo (load side) and the rectified voltage VR = Vo,r. Under the freewheeling state, the primary leg voltage vab = 0 and the secondary rectified voltage VR = Vo,r. It can obtain the primary-side inductor voltage vLP = -np1Vo,r/ns1 < 0 and the primary current iLP will be declined to zero. Hence, the high circulating current drawback is overcome. In order to overcome and achieve wide voltage operation, four secondary windings are adopted in Figure 1b remark in blue. For low voltage input conditions (Vin,min ~ 2.2Vin,min), Q1 turns on and Q2 turns off ( Figure 2a). Thus, D2 and D3 are reverse biased. The proposed circuit has a high voltage gain with transformer turns ratio NT1 = np1/(ns1+ns2). Under high voltage input case (2.2Vin,min ~ 5Vin,min), the switches Q1 turns off and Q2 turns on ( Figure 2b). Therefore, D1 and D4 are reverse biased and the present circuit has low voltage gain with transformer turns ratio NT1 = np1/ns2. Hence, the wide voltage operation, low freewheeling current and wide ZVS operation are realized in the presented hybrid PWM converter.

Operation Principle of the Present Circuit
On the basis of the input voltage range, two cub-circuits can be operated in the present converter. Figures 2a,b provide two sub-circuits under low and high input voltage regions. Under the low voltage region (Vin,min ~ 2.2Vin,min), the full-bridge PWM converter with high secondary turns is operated to obtain high DC voltage gain. To achieve high voltage gain, Q1 turns on and Q2 turns off. Thus, the transformer turns ratio in Figure 2a becomes NT1 = np1/(ns1+ns2). Under the high voltage region (2.2Vin,min ~ 5Vin,min), the present

Operation Principle of the Present Circuit
On the basis of the input voltage range, two cub-circuits can be operated in the present converter. Figure 2a,b provide two sub-circuits under low and high input voltage regions. Under the low voltage region (V in,min~2 .2V in,min ), the full-bridge PWM converter with high secondary turns is operated to obtain high DC voltage gain. To achieve high voltage gain, Q 1 turns on and Q 2 turns off. Thus, the transformer turns ratio in Figure 2a becomes N T1 = n p1 /(n s1 +n s2 ). Under the high voltage region (2.2V in,min~5 V in,min ), the present converter only needs low voltage gain to control load voltage. Therefore, Q 1 turns off and Q 2 turns on to have fewer winding turns on the secondary side. The transformer turns ratio in Figure 2b becomes N T1 = n p1 /n s2 . Due to input voltage deviation, the PSPWM modulation is selected to control active devices and regulate the duty ratio of PWM signals. The resonant circuit is used in the presented circuit in order to improve the ZVS load range for lagging-leg switches. The output voltage V o,r of resonant converter is connected to the rectified terminal voltage V R . Ths positive voltage V o,r can decrease the current i Lp to 0 at a freewheeling state. Then, the freewheeling current loss is improved. In the adopted circuit, the inductance L R << L m,T1 and the output capacitances of S 1~S4 are C S1 = C S2 = C S3 = C S4 = C oss For the low voltage input case (V in,min~2 .2V in,min ), the PWM signals are shown in Figure 2a. In the low input voltage region, Q 1 turns on and Q 2 turns off. It is obvious that the fast recovery diodes D 2 and D 3 are both reverse biased. In this equivalent operating circuit, the secondary turns of T 1 are equal to n s1 +n s2 . From Figure 2a, six states are operated in every half switching period. The equivalent state circuits for the first half switching period are provided in Figure 3.  (1) and (2).
Power transfer between input and output terminals is through a full-bridge PWM converter in this state. The resonant circuit is controlled at the resonant frequency. Since S 4 is ON, the inductor current i Lr will decrease and i Lr is less than the magnetizing current i Lm,T2 . Therefore, D 6 is forward biased and LLC converter will store energy on C o,r . State 2 [t 1~t2 ]: At t 1 , S 1 is off. Prior to t 1 , i Lp is positive. After time t 1, i Lp will discharge C S2 . If the energy (L p + N 2 then v CS2 will decline to 0 at time t 2 . The discharge time of C S2 is ∆t 12 ≈ 2V in C oss N T1 /I o . To ensure the ZVS operation, the other necessary condition is t d (dead time between S 2 and S 1 ) > ∆t 12 . LLC circuit is still operated at resonant mode ( f sw = f r = 1/2π √ L r C r ). State 3 [t 2~t3 ]: v CS2 = 0 at t 2 . Then i Lp flows through D S2 and keeps v CS2,ds = 0. Hence, S 2 can turn on to realize ZVS operation. The secondary rectified voltage V R is decreased to V o,r so that diode D r becomes forward biased and obtains V R = V o,r . The primary and secondary inductor voltages v Lp = −N T1 V o,r and v Lo = V o,r − V o < 0. It can obtain that i Lp and i Lo are both decreased in state 3.
In a traditional PSPWM converter, v Lp ≈ 0 and i Lp almost constant at the freewheeling state. From (3), i Lp in the proposed hybrid converter is decreased in this state. If the freewheeling time interval is large enough, i D1 or i Lp can be decreased to zero.
From Equation (5), one can be observed, the time ∆i Lp=0 is dependent on I o and V o,r . For full load condition, a more freewheeling time interval is needed to achieve no circulating current advantage. and D4 become forward biased. Since iD4 < iLo, Dr is conducting. The inductor voltage vLp ≈ NT1Vo,r − Vin < 0 and iLp decreases. At time t6, the diode current iD4 = iLo, iLp = -iLo/NT1 and iDr = 0. The time interval of state 6 is expressed as )] . Since Dr is conducting in this state, the duty ratio loss of PSPWM circuit at state 6 is calculated as )] The current iLo is still decreased in state 6. This state is   The proposed circuit can also be operated at high voltage input conditions (2.2Vin,min ~ 5Vin,min). The PWM waveforms for high voltage input operation are provided in Figure  2b. Under high voltage input, Q1 is controlled at OFF state and Q2 is ON. Due to Q1 is OFF, D1 and D4 become OFF. The full-bridge PWM converter has turns ratio NT1 = np1/ns2. Since Since D r is conducting in this state, the duty ratio loss of PSPWM circuit at state 6 is calculated as The current i Lo is still decreased in state 6. This state is ended at time t 6 .
The proposed circuit can also be operated at high voltage input conditions (2.2V in,min5 V in,min ). The PWM waveforms for high voltage input operation are provided in Figure 2b. Under high voltage input, Q 1 is controlled at OFF state and Q 2 is ON. Due to Q 1 is OFF, D 1 and D 4 become OFF. The full-bridge PWM converter has turns ratio N T1 = n p1 /n s2 . Since the switching frequency of LLC circuit is equal to the resonant frequency, active devices S 3 and S 4 are turned on at ZVS operation. Due to D r is connecting V o,r and V R , the primary current i Lp can be decreased to 0 at the freewheeling state. Figure 4

Circuit Analysis of the Presented Converter
The advantages of the studied hybrid PSPWM circuit are a wide load range of soft switching, less circulating current and wide input voltage operation. The hard switching drawback and high freewheeling current problem of conventional PSPWM converter are overcome by using an LLC converter in the proposed circuit. Since the switching frequency of LLC circuit is equal to the resonant frequency, active switches of the PSPWM converter can be turned on at ZVS. To realize a wide input voltage deviation problem, four winding sets and two switches are adopted on the low voltage side to control DC voltage gain. Under low input voltage conditions, the winding turns n S1 +n S2 are selected on the secondary side. Under a high input voltage case, the n S2 turns are used on the output side. In the presented circuit, the resonant circuit is controlled under constant switching frequency (equal series resonant frequency). Hence, V o,r is approximately equal to V in /(2N T2 ) = V in n S3 /(2n p2 ). In the presented phase-shift PWM converter, there is a duty loss in state 6. The inductor voltage v Lo in states 1 and 2 is equal tp where d e = d − d 6 is the effective duty cycle and N T1 = n p1 /(n s1 +n s2 ) or n p1 /n s2 for low or high voltage input condition, respectively. In a traditional PSPWM converter, the output voltage is expressed in Equation (7).
Comparison of Equations (12) and (13), it is clear that the presented hybrid PWM circuit has less output inductance L o . The output power of the resonant circuit is P LLC ≈ (0.5 − d e f f )I o V in /N T2 and the output power of the PSPWM circuit is P PW M circuit ≈ 2d e I o V in /N T1 .

Experimental Results
The presented hybrid PSPWM circuit is investigated and confirmed by a prototype circuit. The rated power of the proposed circuit is P o,max = 800 W, the input and output voltages are V in = 90 V-450 V and V o = 48 V. The switching frequency is f sw = 70 kHz. In the presented circuit, Q 1 and Q 2 are ON and OFF under 90 V ≤ V in < 200 V (low voltage input). Therefore, the transformer T 1 has turns ratio N T1 = n p1 /(n s1 + n s2 ). On the other hand, Q 1 and Q 2 are OFF and ON under 200 V < V in ≤ 450 V (high voltage input). Transformer T 1 has turns ratio N T1 = n p1 /n s2 . To present signal oscillation at V in = 200 V, a Schmitt comparator with ± 10V tolerance is adopted in the control circuit to control Q 1 and Q 2 . Hence, Q 1 is ON and Q 2 is OFF at 90 V ≤ V in ≤ 210 V, and Q 1 is OFF and Q 2 is ON at 190 V ≤ V in ≤ 450 V in the control algorithm. The general purpose PWM integrated circuit UCC3895 is selected to control S 1 -S 4 . The switches Q 1 and Q 2 are controlled by a voltage comparator. The component parameters of the laboratory prototype circuit are provided in Table 1. Figure 5a gives the picture of the prototype circuit. The control blocks of the presented converter are given in Figure 5b. The output voltage controller is based on the type III voltage control algorithm [18]. The output of the voltage controller is to regulate the phase shift angle between the leading-leg switches and lagging-leg switches. The gate drivers are used to turn on or off the switching devices S 1 -S 4 . In order to realize the wide input voltage operation, a Schmitt trigger comparator with 200 V reference voltage is adopted to select the low (Q 1 Figures 6d, 7d, 8d and 9d, respectively. Figure 10 demonstrates the test results of S 1 (at leading-leg) and S 4 (at lagging-leg) at V in = 90 V input for 20% load and full load conditions. Owing to the energy on L o is adopted to discharge the leading-leg switch S 1 , one can observe that switch S 1 in Figure 10a,b have ZVS turn-on operation. Due to the LLC circuit operation, S 4 can achieve ZVS turn-on operation shown in Figure 10c,d at 20% and 100% power conditions. Similarly, the measured waveforms of S 1 (at leading-leg) and S 4 (at lagging-leg) at V in = 190 V, 210 V and 450 V under 20% load and full load are demonstrated in Figures 11-13, respectively. Due to the phase-shift PWM operation of the DC full-bridge converter, the PWM signals of S 2 and S 3 have the same operation characteristics as S 1 and S 4 . From the experimental results in Figures 10-13, it is clear that all switches have ZVS characteristic from 20% load to full load for all input voltage range. Figure 14 provides the relationship between the signals of Q 1 and Q 2 and input voltage V in . When V in > 90 V and < 210 V (in low voltage input range), Q 1 turns on and Q 2 turns off. The n 1 + n 2 turns are selected to achieve high voltage gain. On the other hand, Q 1 turns off and Q 2 turns on, if V in > 210 V (in high voltage input range). The n 2 turns are selected to reduce voltage gain. If the input voltage V in is declined from 450 V and V in < 190 V, then Q 1 will turn on and Q 2 turns off. The measured efficiencies of the prototype circuit are about 90%, 88%, 92% and 89% at V in = 90 V, 210 V, 190 V and 450 V, respectively, under full load conditions.     v S1,g v S1,d

Conclusions
A hybrid PWM converter is discussed and implemented to improve the drawbacks of the traditional PSPWM converter. The advantages of the presented circuit are low primary current stress at the freewheeling state, low switching losses at the lagging-leg switches and more wide input voltage operation for PV power renewable energy applications. Compare to the conventional PWM converters in [10][11][12][13][14][15][16][17] with wide input voltage or output voltage operation, the proposed circuit has fewer power switches on the primary side. The LLC resonant converter is used on the lagging-leg of the conventional PSPWM converter to reduce the switching loss. Thus, the switching losses on the proposed converter are improved. The high freewheeling current loss of conventional phase-shift PWM converter is also improved by the connection of the output DC voltage of the resonant circuit to the rectified terminal of the PSPWM circuit. However, the proposed converter has more circuit components compared to the other wide input voltage PWM converters. The performance of the presented hybrid PWM circuit is provided from the experiments with an 800 W prototype circuit. Further work will consider reducing the active or passive components in the presented converter, decreasing the circuit cost and maintaining the same converter performance.