A Novel Hybrid LDC Converter Topology for the Integrated On-Board Charger of Electric Vehicles

: Recently, the integrated On-Board Charger (OBC) combining an OBC converter with a Low-Voltage DC/DC Converter (LDC) has been considered to reduce the size, weight and cost of DC-DC converters in the EV system. This paper proposes a new integrated OBC converter with V2G (Vehicle-to-Grid) and auxiliary battery charge functions. In the proposed integrated OBC converter, the OBC converter is composed of a bidirectional full-bridge converter with an active clamp circuit and a hybrid LDC converter with a Phase-Shift Full-Bridge (PSFB) converter and a forward converter. ZVS for all primary switches and nearly ZCS for the lagging switches can be achieved for all the operating conditions. In the secondary side of the proposed LDC converter, an additional circuit composed of a capacitor and two diodes is employed to clamp the oscillation voltage across rectiﬁer diodes and to eliminate the circulating current. Since the output capacitor of the forward converter is connected in series with the output capacitor of the auxiliary battery charger, the energy from the propulsion battery can be delivered to the auxiliary battery during the freewheeling interval and it helps reduce the current ripple of the output inductor, leading to a smaller volume of the output inductor. A 1 kW prototype converter is implemented to verify the performance of the proposed topology. The maximum efﬁciency of the proposed converter achieved by the experiments is 96%.


Introduction
The energy storage system in Electric Vehicles (EVs) and Plug-in Hybrid Electric Vehicles (PHEVs) normally consist of two kinds of batteries (400 V high voltage battery and 24 V or 48 V low voltage battery). One is the propulsion battery to provide a high DC voltage for the electric motor and the other is the auxiliary battery to supply a low DC voltage for the low voltage electric devices such as the lighting, entertainment, signaling circuit and audio systems. In order to charge those two batteries, in general, two different converters are required: an On-Board Charger (OBC) for the propulsion battery and a Low-Voltage DC/DC Converter (LDC) converter for the auxiliary battery. In the conventional DC-DC converter systems on EVs or PHEVs, since these two converters operate separately, the size of the DC-DC converter system becomes bulky. In order to cope with this problem, an integrated OBC has been introduced by combining the OBC converter with the LDC converter to achieve a smaller volume, lighter weight and lower cost. An integrated OBC converter can perform three different functions: (1) charge operation from Grid-to-Vehicle (G2V), (2) discharge operation from Vehicle-to-Grid (V2G) [1] and (3) charge operation from Propulsion Battery to Auxiliary Battery (P2A).
The OBC converter can be classified into two categories, the two-stage OBC converter and the single-stage OBC converter. In the two-stage OBC converter, a non-isolated boost converter is popularly used for the PFC stage [2][3][4][5]. For the DC-DC converter of This paper includes five sections: Section 1 shows the introduction of the research and Section 2 provides a description about the operating principle of the proposed converter. All the features and design considerations for the proposed converter to get soft switching are presented in Section 3. In Section 4, a prototype 1 kW converter has been implemented to verify performances of the proposed converter. Finally, the conclusion is given in Section 5.

Operation of the Proposed Integrated OBC
The circuit diagram of the LDC converter in the proposed integrated OBC is shown in Figure 2. The primary side is composed of an input capacitor Co1, an active clamp circuit with Cr1 and an active switch Q5 and a PSFB converter with switches Q1, Q2, Q3 and Q4. The transformer of the PSFB converter is TR1 with a magnetizing inductance Lm1, a leakage inductance Llk1 and a turns ratio of n1:1. The transformer of the forward converter is TR2 with a magnetizing inductance Lm2, a leakage inductance Llk2 and a turns ratio of n2:1. The secondary side of the PSFB converter includes rectifier diodes D1 and D2, a passive snubber circuit composed of Cr2, D4 and D5. The capacitor Cr2 connected in parallel with the output inductor Lo2 also plays a function as an output capacitor of the forward converter and reduces the current ripple of the output. As mentioned earlier, the focus is on explaining the operation of the LDC of the proposed integrated OBC. Here, a half of the switching cycle of the proposed converter is divided into seven modes and the operation is explained in detail since the other half is symmetric. The key waveforms and the equivalent circuits of each operation mode are shown in Figures 3 and 4, respectively. For the sake of simplicity, all of the circuit components are ideal except the output capacitance of the switch and all of the output capacitances of the switches are assumed to be same.

Operation of the Proposed Integrated OBC
The circuit diagram of the LDC converter in the proposed integrated OBC is shown in Figure 2. The primary side is composed of an input capacitor C o1 , an active clamp circuit with C r1 and an active switch Q 5 and a PSFB converter with switches Q 1, Q 2, Q 3 and Q 4 . The transformer of the PSFB converter is TR 1 with a magnetizing inductance L m1 , a leakage inductance L LK1 and a turns ratio of n 1 :1. The transformer of the forward converter is TR 2 with a magnetizing inductance L m2 , a leakage inductance L LK2 and a turns ratio of n 2 :1. The secondary side of the PSFB converter includes rectifier diodes D 1 and D 2 , a passive snubber circuit composed of C r2 , D 4 and D 5 . The capacitor C r2 connected in parallel with the output inductor L O2 also plays a function as an output capacitor of the forward converter and reduces the current ripple of the output. As mentioned earlier, the focus is on explaining the operation of the LDC of the proposed integrated OBC. Here, a half of the switching cycle of the proposed converter is divided into seven modes and the operation is explained in detail since the other half is symmetric. The key waveforms and the equivalent circuits of each operation mode are shown in Figures 3 and 4, respectively. For the sake of simplicity, all of the circuit components are ideal except the output capacitance of the switch and all of the output capacitances of the switches are assumed to be same.     At t = t 1 , Q 4 is on and Q 3 is off. Q 2 and Q 5 turn off, and their parasitic capacitors are charged. The parasitic capacitor of Q 1 is discharged and its body diode is forward biased creating ZVS turn-on condition for Q 1 . The power is transferred to load through the transformers TR 1 and TR 2 .
In the secondary side, diodes D 2 and D 5 are reverse biased. D 1 , D 3 and D 4 are forward biased. The resonant capacitor C r2 resonates with output inductor L O2 and discharges the energy to the output.
Mode 2 [t 2 -t 3 ], Figure 4b. At t = t 2 , Q 1 turns on while Q 4 is already on in mode 1. Q 2 , Q 3 and Q 5 are off. The body diode of Q 1 is reverse biased and the current flows through Q 1 . The secondary side works in the same fashion as in mode 1. Due to the resonance between C r2 and L O2 , the current flowing through D 4 decreases to zero and achieves ZCS turn-off condition for D 4 at the end of this mode.   Figure 4a. At t = t1, Q4 is on and Q3 is off. Q2 and Q5 turn off, and their parasitic capacitors are charged. The parasitic capacitor of Q1 is discharged and its body diode is forward biased creating ZVS turn-on condition for Q1. The power is transferred to load through the transformers TR1 and TR2.
In the secondary side, diodes D2 and D5 are reverse biased. D1, D3 and D4 are forward biased. The resonant capacitor Cr2 resonates with output inductor LO2 and discharges the energy to the output.
Mode 2 [t2-t3], Figure 4b. At t = t2, Q1 turns on while Q4 is already on in mode 1. Q2, Q3 and Q5 are off. The body diode of Q1 is reverse biased and the current flows through Q1. The secondary side works in the same fashion as in mode 1. Due to the resonance between Cr2 and LO2, the current flowing through D4 decreases to zero and achieves ZCS turn-off condition for D4 at the end of this mode.
Mode 3 [t3-t4], Figure 4c. . At t = t 3 , Q 1 and Q 4 are on. Q 2 , Q 3 and Q 5 are off. The input power is delivered to the output by both converters. In the forward converter, the primary current i pri2(t) flows through the transformer TR 2 and charges the resonant capacitor C r2 in the secondary side. In the PSFB converter, the primary current i pri1 flows through switches Q 1 , Q 4 and transformer TR 1 . The currents i pri1 and i pri2 (t) are determined as below.
where V pri1 is the primary winding voltage of the transformer TR 1 , V aux_bat is the auxiliary battery voltage, i pri1 is the primary current of the forward converter, i pr2 is the primary current of the PSFB converter, i Lo2 is the current flow through the output inductor L o2 , and V Cr2 is the voltage of capacitor C r2 . In the secondary side, diode D 4 is reverse biased and D 5 is forward biased to charge the resonant capacitor C r2 and the output capacitor C o2 , respectively. The rectifier bridge voltage V rec (t) can be calculated as in (3).
where V rec is the rectifier voltage of the PSFB converter. Mode 4 [t 4 -t 5 ], Figure 4d. At t = t 4 , the switch Q 4 turns off. The parasitic capacitor of Q 3 is discharged. In the forward converter, the current of transformer TR 2 flows through the active clamp circuit including capacitor C r1 and the body diode of Q 5 . The capacitor C r1 resonates with the leakage inductor L LK2 . Since the parasitic capacitor of Q 4 is charged and that of Q 3 is discharged, the ZVS turn-on condition for Q 3 is achieved. The voltage V Coss_Q3 across Q 3 can be found as shown in (4).
In the secondary side, the current commutation occurs from D 1 to D 4 and resonant capacitor C r2 . The current i D1 flowing through D 1 can be determined as follows.
Mode 5 [t 5 -t 6 ], Figure 4e. At t = t 5 , Q 3 and Q 5 are turned on while Q 1 is on. The body diodes of Q 3 and Q 5 and diodes D 1 , D 3 and D 4 are forward biased. The other diodes are reverse biased. The resonant capacitor C r2 is discharged through the diodes D 1 and D 4 . In the forward converter, the resonance between the capacitor C r1 and the leakage inductor L LK2 continues. The power is transferred to the secondary side through the transformer TR 2 of the forward converter.
The primary current i pri1 (t) of the PSFB converter can be expressed using (8).
where i m1 is the magnetizing current of the PSFB converter. The current through diode D 1 can be expressed using (9).
Mode 6 (t 6 -t 7 ), Figure 4f. At t = t 6 , the diode D 1 is reverse biased as the current commutation from D 1 to D 4 is completed. In the primary side of the PSFB converter, Q 1 is on and the body diode of Q 3 is forward biased. The primary current of the PSFB converter is circulating and kept constant. The forward converter operates the same as in mode 5.
In the secondary side, diodes D 3 and D 4 are forward biased. Due to the discharge of L O2 , the current through D 3 decreases to zero gradually. At the end of this mode, the diode D 3 is turned off with ZCS. Mode 7 (t 7 -t 8 ), Figure 4g. At t = t 7 , the body diode of Q 5 is reverse biased and the current flows through the Q 5 . In the forward converter, the capacitor C r1 resonates with the inductance of the transformer TR 2 and the resonant current resets it, thereby eliminating the need for tertiary winding of the forward converter. The primary current of the PSFB converter is still circulated through the Q 1 and the body diode of Q 3 . In this mode, there is no power transferred to the secondary side. As in mode 6, the diode D 4 is still forward biased and the energy in the capacitor C r2 is discharged to the load. After mode 7 the other half of the switching cycle operates in a symmetric fashion. The proposed LDC converter is a combination of a forward converter and a PSFB converter with a high step-down voltage conversion ratio as compared to the converter introduced in [24]. In order to explain the characteristics of the proposed LDC converter in terms of voltage conversion ratio its simplified circuit model is shown in Figure 5.

Features and Design Consideration
completed. In the primary side of the PSFB converter, Q1 is on and the body diode of Q3 is forward biased. The primary current of the PSFB converter is circulating and kept constant. The forward converter operates the same as in mode 5.
In the secondary side, diodes D3 and D4 are forward biased. Due to the discharge of LO2, the current through D3 decreases to zero gradually. At the end of this mode, the diode D3 is turned off with ZCS.
Mode 7 (t7-t8), Figure 4g. At t = t7, the body diode of Q5 is reverse biased and the current flows through the Q5. In the forward converter, the capacitor Cr1 resonates with the inductance of the transformer TR2 and the resonant current resets it, thereby eliminating the need for tertiary winding of the forward converter. The primary current of the PSFB converter is still circulated through the Q1 and the body diode of Q3. In this mode, there is no power transferred to the secondary side. As in mode 6, the diode D4 is still forward biased and the energy in the capacitor Cr2 is discharged to the load. After mode 7 the other half of the switching cycle operates in a symmetric fashion.

High Step-Down Voltage Conversion Ratio
The proposed LDC converter is a combination of a forward converter and a PSFB converter with a high step-down voltage conversion ratio as compared to the converter introduced in [24]. In order to explain the characteristics of the proposed LDC converter in terms of voltage conversion ratio its simplified circuit model is shown in Figure 5. Based on the equivalent circuit shown in Figure 5a. the output voltage Vo of the proposed converter can be derived as (10). Based on the equivalent circuit shown in Figure 5a. the output voltage V o of the proposed converter can be derived as (10).
where D eff is the effective duty cycle of the PSFB converter and D' is the time period from mode 4 to mode 6 (t 4 -t 6 ) when the resonant capacitor C r2 resonates with the leakage inductor L LK2 of the transformer TR 2 as shown in (11).
The voltage conversion ratio of the proposed converter is decided by the turns ratio for n 1 and n 2 as shown in Figure 6.
The voltage applied to the primary winding of each transformer can be expressed as in (12) and (13), respectively.
where V pri2 is the primary winding voltage of the transformer TR 2 and V Pri1 is the primary winding voltage of the transformer Tr 1. By combining (12) and (13), the voltage conversion ratio M proposed of the proposed converter can be calculated as in (14).
Energies where Deff is the effective duty cycle of the PSFB converter and D' is the time period from mode 4 to mode 6 (t4-t6) when the resonant capacitor Cr2 resonates with the leakage inductor LLK2 of the transformer TR2 as shown in (11).
The voltage conversion ratio of the proposed converter is decided by the turns ratio for n1 and n2 as shown in Figure 6. The voltage applied to the primary winding of each transformer can be expressed as in (12) and (13), respectively.

Vin
where Vpri2 is the primary winding voltage of the transformer Tr2 and VPri1 is the primary winding voltage of the transformer Tr1. By combining (12) and (13), the voltage conversion ratio Mproposed of the proposed converter can be calculated as in (14).

Elimination of the Circulating Current
In the conventional PSFB converter, the circulating current losses in the freewheeling interval reduces the power conversion efficiency, especially when the converter works with small effective duty, D. In contrast, the proposed converter can eliminate circulating current due to the additional snubber circuit composed of Cr2, D4 and D5. Figure 7 shows the difference between the circulating current in the conventional PSFB and the proposed converter. As explained in the operation of mode 4 and mode 5, since the resonant capacitor Cr2 is discharged and the diodes D1 and D4 are forward biased the primary current is reduced quickly during the freewheeling period thereby reducing the circulating current. Consequently, ZCS turn-off can nearly be achieved for the lagging leg switches of the PSFB converter.

Elimination of the Circulating Current
In the conventional PSFB converter, the circulating current losses in the freewheeling interval reduces the power conversion efficiency, especially when the converter works with small effective duty, D. In contrast, the proposed converter can eliminate circulating current due to the additional snubber circuit composed of C r2 , D 4 and D 5 . Figure 7 shows the difference between the circulating current in the conventional PSFB and the proposed converter. As explained in the operation of mode 4 and mode 5, since the resonant capacitor C r2 is discharged and the diodes D 1 and D 4 are forward biased the primary current is reduced quickly during the freewheeling period thereby reducing the circulating current. Consequently, ZCS turn-off can nearly be achieved for the lagging leg switches of the PSFB converter.

Small Output Current Ripple
In the conventional PSFB converter, the amplitude of the voltage applied to the output inductor in the powering period is the same as in the freewheeling period, as shown

Small Output Current Ripple
In the conventional PSFB converter, the amplitude of the voltage applied to the output inductor in the powering period is the same as in the freewheeling period, as shown in Figure 8a. However, in the proposed converter, due to the operation of the forward converter, as explained in mode 5 to mode 7, the voltage applied to the output inductor during the freewheeling period is reduced since the diode D5 is reverse biased, as shown in Figure 8b. As a result, the current ripple of the output inductor can be reduced significantly.

Small Output Current Ripple
In the conventional PSFB converter, the amplitude of the voltage applied to the output inductor in the powering period is the same as in the freewheeling period, as shown in Figure 8a. However, in the proposed converter, due to the operation of the forward converter, as explained in mode 5 to mode 7, the voltage applied to the output inductor during the freewheeling period is reduced since the diode D5 is reverse biased, as shown in Figure 8b. As a result, the current ripple of the output inductor can be reduced significantly. In the conventional PSFB converter, the ripple current ∆I1(Conventional_PSFB) of the output inductor can be determined using (15). In the conventional PSFB converter, the ripple current ∆I 1(Conventional_PSFB) of the output inductor can be determined using (15).
The current ripple ∆I 2(Proposed_Converter) of the output inductor in the proposed converter can be calculated as in (16).
The ratio R ripple of the ripple of the current between these two cases can be calculated as in (17). Figure 9 illustrates the ratio of current ripple between two cases with different values of the effective duty cycles. When the voltage of the propulsion battery is 420 V, the duty of the LDC is 0.9 and the current ripple of the output inductor L O2 of the proposed converter is just 29% of that of the conventional PSFB converter. When the voltage of the propulsion battery is 250 V, the current ripple of the output inductor L O2 is just 47% of that of the conventional PSFB converter. Therefore, the value and size of the output filter inductor L O2 can be significantly reduced. Figure 9 illustrates the ratio of current ripple between two cases with different values of the effective duty cycles. When the voltage of the propulsion battery is 420 V, the duty of the LDC is 0.9 and the current ripple of the output inductor Lo2 of the proposed converter is just 29% of that of the conventional PSFB converter. When the voltage of the propulsion battery is 250 V, the current ripple of the output inductor Lo2 is just 47% of that of the conventional PSFB converter. Therefore, the value and size of the output filter inductor Lo2 can be significantly reduced.

ZVS Conditions for All of the Switches in the PSFB over the Full Load Range
In order to achieve ZVS turn-on for MOSFETs, the magnetizing current Im1 of TR1 needs to be large enough. The magnetizing current and the energy stored in the magnetizing inductance of TR1 can be determined as follows.

ZVS Conditions for All of the Switches in the PSFB over the Full Load Range
In order to achieve ZVS turn-on for MOSFETs, the magnetizing current i m1 of TR 1 needs to be large enough. The magnetizing current and the energy stored in the magnetizing inductance of TR 1 can be determined as follows. (19) where I m1,peak is the peak value of the magnetizing current of TR 1 . T S is the switching period and D eff_min is the minimum effective duty cycle of the PSFB converter.
To ensure the ZVS condition over the whole load range, the energy stored in the magnetizing inductance should be larger than that stored in the parasitic capacitors C oss of MOSFETs.
The required magnetizing inductance of TR1 can be calculated as in (21).
In addition to the condition in (21), the dead-time for the MOSFETs also needs to be satisfied (22).

Design of the Clamp and Resonant Capacitors
As explained earlier in the operation of mode 7, the capacitor C r1 resonates with the leakage inductance L LK2 and the magnetizing inductance L m2 of TR 2 . Hence, the value of clamp capacitor C r1 can be calculated as in (23). However, since the voltage applied to the switches of PSFB converter increases due to the resonance, the resonant frequency needs to be selected much lower than the switching frequency in order to reduce it.
The resonant capacitor C r2 is the output capacitor of the forward converter. Thus, its value can be calculated using (24).

Experimental Results
In order to verify the performance of the proposed topology, a prototype converter was implemented. The specification of the proposed converter is shown in Table 1. All of the parameters for the transformer, inductor and capacitor are illustrated in Table 2. This paper focuses on illustrating the experimental results regarding function III of the proposed integrated OBC.
DSSK 60-0045B Figures 10 and 11 show the measured waveforms at the lagging switches and the leading switches with 400 V input, 25 V output and 1 kW output power. In Figure 10, it can be observed that the MOSFET Q 1 turns on with ZVS and turns off with nearly ZCS. There is a small negative current flowing through its body diode to maintain the zero voltage during the turn-on period. The turn-off current of Q 1 is just 0.5 A, thus turn-off losses of the lagging leg MOSFETs are minimized. Figure 11 shows that MOSFET Q 3 is also turned on with ZVS condition. DSSK 60-0045B Figures 10 and 11 show the measured waveforms at the lagging switches and the leading switches with 400 V input, 25 V output and 1 kW output power. In Figure 10, it can be observed that the MOSFET Q1 turns on with ZVS and turns off with nearly ZCS. There is a small negative current flowing through its body diode to maintain the zero voltage during the turn-on period. The turn-off current of Q1 is just 0.5 A, thus turn-off losses of the lagging leg MOSFETs are minimized. Figure 11 shows that MOSFET Q3 is also turned on with ZVS condition.

V Q1 (100 V/div) I Q1 (2 A/div) ZVS turn-on
Nearly ZCS turn-off t: 5 µ/div Figure 10. Voltage and current of switch Q1 at 100% of load. Figure 10. Voltage and current of switch Q 1 at 100% of load.
Energies 2021, 14, x FOR PEER REVIEW 13 of 19 Figure 11. Voltage and current of switch Q3 at 100% of load. Figure 12 shows the waveform of MOSFET Q1 at the light load condition (10% load) with an input voltage of 330 V and output power of 100 W. It can be clearly observed from Figure 12 that the lagging leg switches can maintain ZVS turn-on and nearly ZCS turn-off at light load condition.  Figure 12 shows the waveform of MOSFET Q 1 at the light load condition (10% load) with an input voltage of 330 V and output power of 100 W. It can be clearly observed from Figure 12 that the lagging leg switches can maintain ZVS turn-on and nearly ZCS turn-off at light load condition. Figure 11. Voltage and current of switch Q3 at 100% of load. Figure 12 shows the waveform of MOSFET Q1 at the light load condition (10% load) with an input voltage of 330 V and output power of 100 W. It can be clearly observed from Figure 12 that the lagging leg switches can maintain ZVS turn-on and nearly ZCS turn-off at light load condition.  Figure 13 represents the voltage and current waveforms of the transformer TR1. It can be observed that there is nearly no circulating current in the primary side of the transformer TR1 during freewheeling interval.  The measured waveforms in Figure 14 depict the waveforms at the primary side of the transformer TR2. We can see that the current of transformer TR2 is reset by the active clamp circuit Q5 and Cr1. Figure 15 shows that both ZVS turn-on and ZCS turn-off can be achieved at the secondary rectifier diodes, hence there is no reverse recovery. In addition, the voltage at the rectifier diode is clamped around 80 V so that the diode with a lower voltage rating can be used. The measured waveforms in Figure 14 depict the waveforms at the primary side of the transformer TR 2 . We can see that the current of transformer TR 2 is reset by the active clamp circuit Q 5 and C r1 . Figure 15 shows that both ZVS turn-on and ZCS turn-off can be achieved at the secondary rectifier diodes, hence there is no reverse recovery. In addition, the voltage at the rectifier diode is clamped around 80 V so that the diode with a lower voltage rating can be used.
The measured waveforms in Figure 14 depict the waveforms at the primary side of the transformer TR2. We can see that the current of transformer TR2 is reset by the active clamp circuit Q5 and Cr1. Figure 15 shows that both ZVS turn-on and ZCS turn-off can be achieved at the secondary rectifier diodes, hence there is no reverse recovery. In addition, the voltage at the rectifier diode is clamped around 80 V so that the diode with a lower voltage rating can be used.  The measured waveforms at diode D3 are shown in Figure 16. It can achie ZVS turn-on and ZCS turn-off. The voltage across the diode D3 oscillates due to t age inductance of the transformer TR2. Figure 17 shows the current and voltag forms at the active clamp switch Q5 with ZVS turn-on. The measured waveforms at diode D 3 are shown in Figure 16. It can achieve both ZVS turn-on and ZCS turn-off. The voltage across the diode D 3 oscillates due to the leakage inductance of the transformer TR 2 . Figure 17 shows the current and voltage waveforms at the active clamp switch Q 5 with ZVS turn-on.
The measured waveforms at diode D 4 are shown in Figure 18. Both ZVS turn-on and ZCS turn-off are achieved at diode D 4 , thus there is no recovery loss at this diode.
The efficiency during function III operation with a wide range of input voltage variation is measured and shown in Figure 19 with the wide input voltage variation from 290 V to 400 V. The proposed converter shows a high efficiency all over the load range and the maximum efficiency is 96.03% at 500 W, which is much higher than those of conventional ones in [19,22,25]. The measured waveforms at diode D3 are shown in Figure 16. It can achieve both ZVS turn-on and ZCS turn-off. The voltage across the diode D3 oscillates due to the leakage inductance of the transformer TR2. Figure 17 shows the current and voltage waveforms at the active clamp switch Q5 with ZVS turn-on.  The measured waveforms at diode D4 are shown in Figure 18. Both ZVS turn-on and ZCS turn-off are achieved at diode D4, thus there is no recovery loss at this diode.  The measured waveforms at diode D4 are shown in Figure 18. Both ZVS turn-on and ZCS turn-off are achieved at diode D4, thus there is no recovery loss at this diode. The efficiency during function III operation with a wide range of input voltage variation is measured and shown in Figure 19 with the wide input voltage variation from 290 V to 400 V. The proposed converter shows a high efficiency all over the load range and the maximum efficiency is 96.03% at 500 W, which is much higher than those of conven- The efficiency during function I and function II operations is also shown in Figures  20 and 21, respectively. In function I, the maximum efficiency is 98.2% when VDC = 400 V, VDC = 420 V and PO_I = 2.3 kW. In function II, the maximum efficiency is 97.58% when VDC = 400 V, VPro_Bat = 420 V and PO_II = 1.8 kW.  The efficiency during function I and function II operations is also shown in Figures 20 and 21, respectively. In function I, the maximum efficiency is 98.2% when V DC = 400 V, V DC = 420 V and P O_I = 2.3 kW. In function II, the maximum efficiency is 97.58% when V DC = 400 V, V Pro_Bat = 420 V and P O_II = 1.8 kW. The efficiency during function I and function II operations is also shown in Figures  20 and 21, respectively. In function I, the maximum efficiency is 98.2% when VDC = 400 V, VDC = 420 V and PO_I = 2.3 kW. In function II, the maximum efficiency is 97.58% when VDC = 400 V, VPro_Bat = 420 V and PO_II = 1.8 kW.  The efficiency during function I and function II operations is also shown in Figures  20 and 21, respectively. In function I, the maximum efficiency is 98.2% when VDC = 400 V, VDC = 420 V and PO_I = 2.3 kW. In function II, the maximum efficiency is 97.58% when VDC = 400 V, VPro_Bat = 420 V and PO_II = 1.8 kW.

Conclusions
This paper has introduced a novel hybrid LDC converter for the integrated OBC using a combination of a PSFB converter and a forward converter. The proposed converter has a high voltage conversion ratio and a high efficiency characteristic due to the hybrid structure. The power is shared by two converters and hence the conduction loss is reduced. The efficiency of the proposed converter can be further improved due to the soft-switching characteristics throughout the load range. The proposed integrated OBC shows over 97% efficiency during function I operation, over 96% efficiency during function II operation and over 95% efficiency during function III operation almost throughout the load range, respectively. In addition, the size of the output inductor can be reduced significantly due to the cascaded output capacitors of the proposed LDC converter. The circulating current is reduced by using the passive snubber circuit, which also helps clamp the voltage applied to the rectifier diodes. The cost and the volume of the OBC can be significantly reduced due to the integrated structure of the proposed converter, and the fuel economy of the electric vehicles can be improved due to its high efficiency characteristics throughout the load range.