A Si-FET-Based High Switching Frequency Three-Level LLC Resonant Converter

This paper highlights the proposed silicon field-effect transistor (Si-FET)-based high switching frequency three-level (TL) LLC resonant converter. It provides a detailed operational analysis of the converter; the multilevel (ML) organization of cells; voltage-balancing principles; current-balancing principles; loss comparison between Si-FETs and gallium-nitride (GaN)-FETs; and an optimal design consideration based on loss analysis. This analysis reveals that the switching losses of all power switches can be considerably reduced as the voltage across each switch can be set to half of the input voltage without an additional circuit or control strategy. Moreover, the current of each resonant inductor is automatically balanced by a proposed integrated magnetic (IM)-coupled inductor. Therefore, the operating frequency can be easily increased to near 1 MHz without applying high-performance switches. In addition, the resonant tanks of the converter can be a group of cells for multilevel operation, which indicates that the voltage across each switch is further reduced as more cells are added. Based on the results of the analysis, an optimal design consideration according to the resonant tank and switching frequency is discussed. The proposed converter was validated via a prototype converter with an input of 390 V, an output of 19.5 V/18 A, and a frequency of 1 MHz.


Introduction
Size minimization of modern power systems, including those for communications, TV, and electric vehicle (EV) systems, has become an increasingly important objective. Size reduction of converters is oftentimes accompanied by high-frequency minimization of magnetics, i.e., inductors and transformers. Unfortunately, during the process, the high switching loss causes a serious thermal problem and a large heat sink, which can be resolved by the application of a soft-switching technique or the use of high-performance switches.
Some studies have proposed certain kinds of high-frequency DC-DC topologies with softswitching [1-7]. Practically, employing soft-switching methods on the main switches can reduce the switching loss and achieve higher efficiency, higher power density, and lower system costs. Nonetheless, these cannot achieve zero-voltage switching (ZVS) in all load conditions, aside from the fact that an additional circuit potentially degrades the power density.
A number of LLC resonant converters can easily solve the abovementioned problems [8,9]. Under all load conditions, all switches can operate with the ZVS without additional circuits. Moreover, zero-current switching (ZCS) of output rectifier diodes can be ensured, thereby removing the need for snubber circuits to prevent voltage spikes and reverse recovery problems. The LLC resonant converter achieves high efficiency during high-frequency operation. However, with these converters, increasing the switching frequency is restrained by the higher turned-off loss of silicon field-effect transistors (Si-FETs). To improve the performance of the high-frequency operation, high-performance switches, i.e., gallium-nitride field-effect transistors ((GaN)-FETs) can be employed instead of Si-FETs [10][11][12][13]. These switches feature low on-state resistance, fast rise-and-fall time, and small drain-source capacitance, which favor the dramatic decrease in the switching and conduction losses. On the other hand, they are costly and require an additional gate driver and a voltage source near 5 V.
Among the various means of switching losses reduction, various three-level (TL) resonant converters with half of the input voltage for the voltage across the main switches have been utilized in some studies [14][15][16][17][18][19]. The principle behind the practicality of such converters is the possible utilization of high-performance switches due to the reduced rated voltage on the switches, mainly as a consequence of the lower drain-source resistance relative to the lower breakdown voltage [18]; moreover, the lower drain-source voltage can reduce the switching power loss [19]. In [14], although the voltage stress of all switches was found to be half the input voltage, additional dead time was required, leading to more complex gate driving, smaller effective duty, and larger additional circulating current. A fixed-frequency TL LLC resonant converter was presented in [15]. For this converter, although the voltage across switches was half of the input voltage, two clamp diodes were required. Moreover, the main switches cannot be guaranteed to be at half the input voltage when there is a slight difference in the parasitic capacitance of switches or the gate-driving signal. Figure 1 shows a conventional TL LLC resonant converter [17]. It consists of two half-bridge (HB) LLC resonant converters arranged in series and sharing a transformer and a resonant inductor. Switches Q 1 and Q 3 are operated complementarily to Q 2 and Q 4 at a constant duty (D = 0.5). When Q 1 and Q 3 are turned on, the current flows through the input capacitor C in1 are as shown in Figure 1a, and the voltage is determined by the equation: (1) Alternatively, when Q 2 and Q 4 are on, the current flows through the other input capacitor, C in2 , are as shown in Figure 1b, and the voltage is determined by the equation: Therefore, the voltage across the switches is set to half of the input voltage. Moreover, capacitors C r1 and C r2 are implemented as the resonant capacitors.
However, given different capacitance, the resonant capacitor currents i Cr1 and i Cr2 are imbalanced due to the impedance difference of each resonant capacitor. Thus, the voltages of each input capacitor V Cin1 and V Cin2 cannot be guaranteed to be at half the input voltage because of the current imbalance. Consequently, the voltage across the switches cannot be guaranteed to be at half the input voltage. To confirm these issues, the Power SIM (PSIM) (Rockville, MD, USA) simulation results are presented in Figure 2, considering the following parameters: 1.
Turn ration: n = 5 Figure 2 displays the PSIM simulation waveforms of the voltage across input capacitors, the currents through resonant capacitor, and the drain-source voltage of the conventional TL LLC resonant converter. Assuming different capacitance for the resonant capacitors, the voltage of the input capacitor gets imbalanced, together with the drain-source voltage of switches. Furthermore, switches Q 1 and Q 2 operate under hard switching because they have higher voltage stress than Q 3 and Q 4 . Therefore, the conventional TL LLC resonant converter cannot ensure the voltage balance and ZVS of all switches.   Figure 2 displays the PSIM simulation waveforms of the voltage across input capacitors, the currents through resonant capacitor, and the drain-source voltage of the conventional TL LLC resonant converter. Assuming different capacitance for the resonant capacitors, the voltage of the input capacitor gets imbalanced, together with the drain-source voltage of switches. Furthermore, switches Q1 and Q2 operate under hard switching because they have higher voltage stress than Q3 and Q4. Therefore, the conventional TL LLC resonant converter cannot ensure the voltage balance and ZVS of all switches. This paper proposes a Si-FET-based high switching frequency TL LLC resonant converter, with a discussion of its mode analysis, voltage-balancing principles, current balancing principles, and optimal design considerations for high efficiency. Structurally, it consists of four stacked switches in series, two resonant tanks, and a balancing capacitor connected to each resonant tank. Switches Q1 and Q3 in the converter operate with a constant duty ratio, with the other pair (Q2 and Q4) complementing their operation. This converter is especially important as it ensures the balance of the voltage across each switch and the current through the IM-coupled inductor. First, the balancing capacitor ensures that the voltage across each switch is half that of the input voltage, without any additional dead time or clamp diodes required. Here, because only the balancing capacitor is required to ensure the balance in voltage, there is no need for an additional circuit or digital IC. Moreover, the offset voltage of the resonant capacitors is a quarter of the input voltage, allowing the use of lowvoltage rating capacitors. Secondly, the IM-coupled inductor in the converter ensures the current balance of each resonant inductor without adding a balancing circuit or control strategy.
The rest of the paper is structured as follows. The detailed circuit operation is presented in Section 2. An analysis of the voltage balancing principles, current balancing principles, voltage conversion ratio, performance comparison between Si-FETs and GaN-FETs, and optimal design consideration based on loss comparison is given in Section 3. The validity of the TL LLC resonant converter is confirmed through the experimental results of a prototype converter from an input of 390 V and an output of 19.5 V/18 A, as shown in Section 4. Conclusions are provided in Section 5. This paper proposes a Si-FET-based high switching frequency TL LLC resonant converter, with a discussion of its mode analysis, voltage-balancing principles, current balancing principles, and optimal design considerations for high efficiency. Structurally, it consists of four stacked switches in series, two resonant tanks, and a balancing capacitor connected to each resonant tank. Switches Q 1 and Q 3 in the converter operate with a constant duty ratio, with the other pair (Q 2 and Q 4 ) complementing their operation. This converter is especially important as it ensures the balance of the voltage across each switch and the current through the IM-coupled inductor. First, the balancing capacitor ensures that the voltage across each switch is half that of the input voltage, without any additional dead time or clamp diodes required. Here, because only the balancing capacitor is required to ensure the balance in voltage, there is no need for an additional circuit or digital IC. Moreover, the offset voltage of the resonant capacitors is a quarter of the input voltage, allowing the use of low-voltage rating capacitors. Secondly, the IM-coupled inductor in the converter ensures the current balance of each resonant inductor without adding a balancing circuit or control strategy.
The rest of the paper is structured as follows. The detailed circuit operation is presented in Section 2. An analysis of the voltage balancing principles, current balancing principles, voltage conversion ratio, performance comparison between Si-FETs and GaN-FETs, and optimal design consideration based on loss comparison is given in Section 3. The validity of the TL LLC resonant converter is confirmed through the experimental results of a prototype converter from an input of 390 V and an output of 19.5 V/18 A, as shown in Section 4. Conclusions are provided in Section 5.  Figure 3 is a schematic representation of the proposed TL LLC resonant converter, which consists of four stacked switches in parallel and two resonant tanks. The converter's key waveforms in a steady state are depicted in Figure 4; here, Q 1 and Q 3 are operated with a constant duty ratio (D = 0.5), with the complementary Q 2 and Q 4 . Thus, the converter has six switching period modes, with each detailed operating circuit illustrated in Figure 5. To illustrate the operation of the proposed converter, several assumptions are considered:

1.
Output capacitor C O is large enough to maintain a constant voltage.

2.
The four resonant capacitors, C r1 , C r2 , C r3 , and C r4 , have the same capacitance of C r . 3.
The balancing capacitor C B is larger than the resonant capacitors (C r1 , C r2 , C r3 , and C r4 ) to be considered as a voltage source of V IN /2.

4.
The output capacitance of all Si-FETs is the same as the capacitance of C ds . 5.
All Si-FETs and main switches have parasitic diodes of D Q1 , D Q2 , D Q3 , and D Q4 . 6.
The two resonant inductors, L r1 and L r2 , have the same inductance L r . 7.
The two magnetizing inductors, L m1 and L m2 , have the same inductance L m . 8.
The turn ratio of the main transformer T is n = N p /N s .

Circuit Operation in a Steady State
Mode 1 [t 0 − t 1 ]: This mode begins when Q 1 and Q 3 are turned on with ZVS at t 0 . Figure 6a shows the equivalent circuit during Mode 1. Resonant inductors L r1 and L r2 have a positive current flow with sinusoidal waveform by the resonance of C r1 , C r2 , C r3 , and C r4 , as follows:   As the rectifier diode D 1 is conducting, the output voltage V O is reflected in the turn ratio, n. Therefore, the primary voltage of transformer V T (t) is set to nV O . Subsequently, the magnetizing inductor L m is built up, and its current i Lm (t) is linearly raised from negative to positive: Voltages across the main switches V DSQ2 and V DSQ4 are set to half the input voltage. The offset voltage of the resonant capacitor can also be set to half the input voltage, and the balancing capacitor has a constant voltage of half the input voltage. The equivalent circuit during Mode 1 is shown in Figure 6a, with the equations of voltage being: where V T1 = (z 1 i Lr (t 0 )) 2 2L r C r , and z 1 = L r /(2C r ). Mode 2 [t 1 − t 2 ]: At the end of Mode 2, the primary currents i Lr1 (t) and i Lr2 (t) reach the same current level as the magnetizing currents i Lm1 (t) and i Lm2 (t). Thus, the magnetizing inductors L m1 and L m2 take part in the resonance with two resonant tanks: Accordingly, the voltages across the main switches, V DSQ2 and V DSQ4 , are set to half of the input voltage, with the offset voltage of the resonant capacitor also set to half of the same input voltage, as the balancing capacitor has a constant voltage that is half of the input voltage. In addition, output rectifier D 1 is turned off with ZCS and becomes reverse-biased. During this mode, with the output being higher than the secondary voltage of the transformer, it is separated from the transformer primary side. Therefore, the output capacitor supplies to the load: where : This mode begins when Q 1 and Q 3 are turned off. Subsequently, the voltages of parasitic capacitors C ds1 and C ds3 are charged from 0 to half of the input voltage V IN /2; linearly, the voltages of parasitic capacitor C ds2 and C ds4 are discharged from V IN /2 to 0 by i Lm (t 1 ). After i Lm fully charges C ds1 and C ds3 , and discharges C ds2 and C ds3 , D b2 and D b4 , the anti-parallel diodes of Q 2 and Q 4 , are forward-biased, and then the resonant tank of C r1 , C r2 , and L r1 and another resonant tank of C r3 , C r4 , and L r2 conduct simultaneous resonance. In addition, as the rectifier diode D 2 is conducting, the primary voltage of transformer V T (t) decreases to −nV O . The current and voltage for this mode are expressed as follows: Figure 5d-f describes the operation principle of Modes 4-6, the equivalent circuit, and the operating circuit, which are the same as for Modes 1-3; thus, no further description is needed.

Principle of Voltage Balancing
To ensure that the voltage across the main switches is half of the input voltage, the balancing capacitor voltage VCB, should be maintained. The TL LLC resonant converter can maintain VCB at half of the input voltage with no additional components or digital control. Nonetheless, the capacitance

Principle of Voltage Balancing
To ensure that the voltage across the main switches is half of the input voltage, the balancing capacitor voltage VCB, should be maintained. The TL LLC resonant converter can maintain VCB at half of the input voltage with no additional components or digital control. Nonetheless, the capacitance of the balancing capacitor should be 10 times that of the converter, large enough to not have any effect on the resonance: Figure 7 shows the voltage-balancing principle according to the switch operation. Firstly, as soon as Q1 and Q3 are turned on, the current flow through the balancing capacitor CB is as shown in Figure

Principle of Voltage Balancing
To ensure that the voltage across the main switches is half of the input voltage, the balancing capacitor voltage V CB , should be maintained. The TL LLC resonant converter can maintain V CB at half of the input voltage with no additional components or digital control. Nonetheless, the capacitance of the balancing capacitor should be 10 times that of the converter, large enough to not have any effect on the resonance: Figure 7 shows the voltage-balancing principle according to the switch operation. Firstly, as soon as Q 1 and Q 3 are turned on, the current flow through the balancing capacitor C B is as shown in Figure 7a, with its voltage represented by: Secondly, when Q 2 and Q 4 are turned on, the current flow through the balancing capacitor is as shown in Figure 7b, with the voltage as follows: Finally, the sum of voltages of all the resonant capacitor are the same as the input voltage:

Principle of Voltage Balancing
To ensure that the voltage across the main switches is half of the input voltage, the balancing capacitor voltage VCB, should be maintained. The TL LLC resonant converter can maintain VCB at half of the input voltage with no additional components or digital control. Nonetheless, the capacitance of the balancing capacitor should be 10 times that of the converter, large enough to not have any effect on the resonance: Figure 7 shows the voltage-balancing principle according to the switch operation. Firstly, as soon as Q1 and Q3 are turned on, the current flow through the balancing capacitor CB is as shown in Figure  7a, with its voltage represented by:  Therefore, the voltage across the main switches is set to half the input voltage. In addition, when the capacitance of the resonant capacitors has a different value in the conventional TL LLC resonant converter mentioned in earlier sections, the voltage across the main switches cannot be guaranteed to be half of the input voltage. However, with the proposed voltage-balancing method, the offset voltage of each resonant capacitor is fixed at nV O regardless of the value of the resonant capacitor. Consequently, as the voltage of the balancing capacitor clamps the voltage across the main switches, the voltage across them is also set to half of the input voltage.
Moreover, the proposed TL LLC resonant converter can be extended into a multi-level (ML) LLC resonant converter by organization of the cells. Figure 8a shows an example of the proposed TL LLC resonant converter with the organization of cells. The basic cell unit consists of two switches, Q 2k and Q 2k−1 ; a resonant capacitor, C r_2k and C r_2k−1 ; a resonant inductor, L r_k ; and a transformer, T k , as shown in Figure 8b. The ML LLC resonant converter with n cells is shown in Figure 9. It consists of n cells and n balancing capacitors, which are connected to the node A n of each cell. Through the addition of the n cells, the voltage across the main switches can be proportionally reduced to V IN /n according to the number of cells.
When group Q 2k−1 is turned on, V Cbm is the same as the sum of V cr_2k−1 and V cr_2k . In contrast, when the group Q 2k is turned on, V Cbm is the same for the sum of V Cr_2x, V Cr_2x−1 , V Cr_4x, and V Cr_4x−1 for m = 1, 2, · · · n − 2, n − 1, and k = 1, 2, · · · n − 1, n. V Cbm is obtained as follows: Based on Equation (25), all balancing capacitors can implement the voltage balancing of each cell. In addition, the sum of all resonant capacitor voltages is the same for the input voltage. Considering Equation (24), the voltage stress of all switches is set to V IN /n: the capacitance of the resonant capacitors has a different value in the conventional TL LLC resonant converter mentioned in earlier sections, the voltage across the main switches cannot be guaranteed to be half of the input voltage. However, with the proposed voltage-balancing method, the offset voltage of each resonant capacitor is fixed at nVO regardless of the value of the resonant capacitor. Consequently, as the voltage of the balancing capacitor clamps the voltage across the main switches, the voltage across them is also set to half of the input voltage. Moreover, the proposed TL LLC resonant converter can be extended into a multi-level (ML) LLC resonant converter by organization of the cells. Figure 8a shows an example of the proposed TL LLC resonant converter with the organization of cells. The basic cell unit consists of two switches, Q2k and Q2k-1; a resonant capacitor, Cr_2k and Cr_2k−1; a resonant inductor, Lr_k; and a transformer, Tk, as shown in Figure 8b. The ML LLC resonant converter with n cells is shown in Figure 9. It consists of n cells and n balancing capacitors, which are connected to the node An of each cell. Through the addition of the n cells, the voltage across the main switches can be proportionally reduced to VIN/n according to the number of cells.   the capacitance of the resonant capacitors has a different value in the conventional TL LLC resonant converter mentioned in earlier sections, the voltage across the main switches cannot be guaranteed to be half of the input voltage. However, with the proposed voltage-balancing method, the offset voltage of each resonant capacitor is fixed at nVO regardless of the value of the resonant capacitor. Consequently, as the voltage of the balancing capacitor clamps the voltage across the main switches, the voltage across them is also set to half of the input voltage.
Moreover, the proposed TL LLC resonant converter can be extended into a multi-level (ML) LLC resonant converter by organization of the cells. Figure 8a shows an example of the proposed TL LLC resonant converter with the organization of cells. The basic cell unit consists of two switches, Q2k and Q2k-1; a resonant capacitor, Cr_2k and Cr_2k−1; a resonant inductor, Lr_k; and a transformer, Tk, as shown in Figure 8b. The ML LLC resonant converter with n cells is shown in Figure 9. It consists of n cells and n balancing capacitors, which are connected to the node An of each cell. Through the addition of the n cells, the voltage across the main switches can be proportionally reduced to VIN/n according to the number of cells.

Principle of Current Balancing
The resonant inductor is generally implemented with the leakage inductor of the transformer. Nonetheless, the transformer of the proposed TL LLC resonant converter has a very small leakage inductance due to the strongly coupled structure of the planar transformer; therefore, the proposed converter must have either two additional inductors or one differential mode (DM)-coupled inductor at each primary side of the transformer. In this case, the current balance of each primary side is very important. Assuming that the parameters of the transformer and gate signals are ideal, the average current of magnetizing inductors <i Lm1 > and <i Lm2 > is 0. Subsequently, the magnetizing current of each resonant tank assumes the same value: In addition, the primary currents i Lr1 and i Lr2 have the same value: However, with every component having a tolerance, each resonant capacitor yields a different practical value. Likewise, the practical impedance of each DM-coupled inductor becomes different, because of the different wire lengths and air gaps. Therefore, when the abovementioned conditions occur, the current of each resonant inductor, as well as the current stress of each power switch, becomes imbalanced. Figure 10 shows the PSIM simulation waveforms of each resonant inductor and drain-source voltage. When the value of each resonant capacitor has ±10% tolerance, the drain-source voltage can be set to half of the input voltage by the balancing capacitor, but the current of each resonant inductor is imbalanced, as shown in Figure 10a. In addition, when the value of the resonant inductor has ±10% tolerance, the voltage across each power switch can be set to half of the input voltage with the aid of the balancing capacitor. On the other hand, Figure 10b shows that the current of each resonant inductor is imbalanced, which can induce destruction of the switches and thermal imbalance problems.
inductance due to the strongly coupled structure of the planar transformer; therefore, the proposed converter must have either two additional inductors or one differential mode (DM)-coupled inductor at each primary side of the transformer. In this case, the current balance of each primary side is very important. Assuming that the parameters of the transformer and gate signals are ideal, the average current of magnetizing inductors <iLm1> and <iLm2> is 0. Subsequently, the magnetizing current of each resonant tank assumes the same value: = . (28) In addition, the primary currents iLr1 and iLr2 have the same value: However, with every component having a tolerance, each resonant capacitor yields a different practical value. Likewise, the practical impedance of each DM-coupled inductor becomes different, because of the different wire lengths and air gaps. Therefore, when the abovementioned conditions occur, the current of each resonant inductor, as well as the current stress of each power switch, becomes imbalanced. Figure 10 shows the PSIM simulation waveforms of each resonant inductor and drain-source voltage. When the value of each resonant capacitor has 10% tolerance, the drainsource voltage can be set to half of the input voltage by the balancing capacitor, but the current of each resonant inductor is imbalanced, as shown in Figure 10a. In addition, when the value of the resonant inductor has 10% tolerance, the voltage across each power switch can be set to half of the input voltage with the aid of the balancing capacitor. On the other hand, Figure 10b shows that the current of each resonant inductor is imbalanced, which can induce destruction of the switches and thermal imbalance problems.  To overcome the practical current imbalance, common-mode (CM)-coupled inductor is inserted into the transformer primary side as shown in Figure 11. Despite the existence of the tolerances of resonant capacitors and inductors, the CM-coupled inductor achieves the exact current balance of each transformer primary side. However, as there is the need for an additional CM-coupled inductor besides the resonant inductors, the converter size may be increased. To overcome the practical current imbalance, common-mode (CM)-coupled inductor is inserted into the transformer primary side as shown in Figure 11. Despite the existence of the tolerances of resonant capacitors and inductors, the CM-coupled inductor achieves the exact current balance of each transformer primary side. However, as there is the need for an additional CM-coupled inductor besides the resonant inductors, the converter size may be increased.  Figure 12a shows the proposed single-core IM capable of replacing the CM-coupled inductor and resonant inductors. Apparently, the proposed IM-coupled inductor not only achieves current balance, but also makes the converter more compact. The reluctance model of the proposed IMcoupled inductor based on the proposed structure of the CM-coupled inductor in Figure 12a is as shown in Figure 12b. The equivalent circuit is similar to the magnetic structure; the magneto-motive force (mmf) F [A] is produced by n-turn windings, NLr1 and NLr2 and carrying currents of iLr1 and iLr2,  Figure 12a shows the proposed single-core IM capable of replacing the CM-coupled inductor and resonant inductors. Apparently, the proposed IM-coupled inductor not only achieves current balance, but also makes the converter more compact. The reluctance model of the proposed IM-coupled inductor based on the proposed structure of the CM-coupled inductor in Figure 12a is as shown in Figure 12b. The equivalent circuit is similar to the magnetic structure; the magneto-motive force (mmf) F [A] is produced by n-turn windings, N Lr1 and N Lr2 and carrying currents of i Lr1 and i Lr2 , and each reluctance of R = µ o g + /µ r /A H −1 , where is the magnetic path length, g is the air-gap of the core, µ r is the relative permeability, µ o is the absolute permeability, and A is the cross-sectional area of the core. Figure 13a illustrates the proposed IM inductance model based on the reluctance model. Also, the IM inductance model in Figure 13a can be represented as the equivalent IM-coupled inductor, as shown in Figure 14.  Figure 12a shows the proposed single-core IM capable of replacing the CM-coupled inductor and resonant inductors. Apparently, the proposed IM-coupled inductor not only achieves current balance, but also makes the converter more compact. The reluctance model of the proposed IMcoupled inductor based on the proposed structure of the CM-coupled inductor in Figure 12a is as shown in Figure 12b. The equivalent circuit is similar to the magnetic structure; the magneto-motive force (mmf) F [A] is produced by n-turn windings, NLr1 and NLr2 and carrying currents of iLr1 and iLr2, and each reluctance of where ℓ is the magnetic path length, ℓ is the air-gap of the core, is the relative permeability, is the absolute permeability, and A is the cross-sectional area of the core. Figure 13a illustrates the proposed IM inductance model based on the reluctance model. Also, the IM inductance model in Figure 13a can be represented as the equivalent IM-coupled inductor, as shown in Figure 14.   Figure 12a shows the proposed single-core IM capable of replacing the CM-coupled inductor and resonant inductors. Apparently, the proposed IM-coupled inductor not only achieves current balance, but also makes the converter more compact. The reluctance model of the proposed IMcoupled inductor based on the proposed structure of the CM-coupled inductor in Figure 12a is as shown in Figure 12b. The equivalent circuit is similar to the magnetic structure; the magneto-motive force (mmf) F [A] is produced by n-turn windings, NLr1 and NLr2 and carrying currents of iLr1 and iLr2, and each reluctance of where ℓ is the magnetic path length, ℓ is the air-gap of the core, is the relative permeability, is the absolute permeability, and A is the cross-sectional area of the core. Figure 13a illustrates the proposed IM inductance model based on the reluctance model. Also, the IM inductance model in Figure 13a can be represented as the equivalent IM-coupled inductor, as shown in Figure 14.  The equivalent inductance model of the proposed IM-coupled inductor can be expressed as the equivalent two resonant inductors, L r1 and L r2 , and CM-coupled inductor. Consequently, the inductance of each inductor can be derived as follows: where A c is the cross section area of the core outer legs, A g is the cross section area of the core center leg, c is the air gap of the core outer legs, and g is the air gap of the core center leg.
Assuming that the air gaps of the outer legs are 0, as the value of inductors L c1 and L c2 are very large, the current of each resonant is almost same. However, small air gaps is required to prevent the core saturation and improve the core loss. Therefore, there is a current difference between each resonant inductor according to the ratio of L c and L g . Figure 13b shows the current difference ratio according to the normalized L (=L c /L g ), where the current difference ratio is defined as |i Lr1 − i Lr2 |/[(i Lr1 + i Lr2 )/2]. As shown in Figure 13b, the larger normalized L produces a smaller current difference. In summary, the resonant inductors L r1 and L r2 are implemented as the leakage inductances of the proposed IM inductor, which can be adjusted mainly by the air gap of the core center leg. In addition, the CM-coupled inductor for current balancing is implemented as the magnetizing inductance of the proposed IM inductor, which is determined mainly by the reluctance of core outer legs.

Voltage Conversion Ratio
If the TL LLC resonant converter operates at the resonant frequency, the resonant current flow follows a nearly sinusoidal waveform pattern, which would validate the use of fundamental harmonic approximation to derive the DC characteristic of the LLC resonant converter. The simplified equivalent circuit, which includes a single resonant tank to easily obtain the voltage conversion ratio, is derived as shown in Figure 15.
where is the cross section area of the core outer legs, is the cross section area of the core center leg, is the air gap of the core outer legs, and is the air gap of the core center leg. Assuming that the air gaps of the outer legs are 0, as the value of inductors Lc1 and Lc2 are very large, the current of each resonant is almost same. However, small air gaps is required to prevent the core saturation and improve the core loss. Therefore, there is a current difference between each resonant inductor according to the ratio of Lc and Lg. Figure 13b shows the current difference ratio according to the normalized L (=Lc/Lg), where the current difference ratio is defined as |iLr1 − iLr2|/[(iLr1 + iLr2)/2]. As shown in Figure 13b, the larger normalized L produces a smaller current difference. In summary, the resonant inductors Lr1 and Lr2 are implemented as the leakage inductances of the proposed IM inductor, which can be adjusted mainly by the air gap of the core center leg. In addition, the CM-coupled inductor for current balancing is implemented as the magnetizing inductance of the proposed IM inductor, which is determined mainly by the reluctance of core outer legs.

Voltage Conversion Ratio
If the TL LLC resonant converter operates at the resonant frequency, the resonant current flow follows a nearly sinusoidal waveform pattern, which would validate the use of fundamental harmonic approximation to derive the DC characteristic of the LLC resonant converter. The simplified equivalent circuit, which includes a single resonant tank to easily obtain the voltage conversion ratio, is derived as shown in Figure 15.

Vin_F (t)/2 Vo_F (t)
2Cr Lr Rac Lm  To obtain mathematical expressions, the total load resistance, R ac , and the fundamental input and output voltage of the resonant tank can be expressed as follows: Meanwhile, the voltage conversion ratio of the proposed converter can be expressed as follows: where

ZVS Condition
To ascertain that the ZVS operates in a steady state, the stored energy of the magnetizing inductors L m1 and L m2 should be sufficiently large to charge and discharge the parasitic capacitors of switches during Modes 3 and 6. In the TL LLC resonant converter, because the voltage stress of switches is half the input voltage, the ZVS condition can be expressed as follows:

Maximum Gain Analysis
With the TL LLC resonant converter operating at the resonant frequency, the magnetizing current during the dead time i Lm and the limit of the magnetizing inductance can be expressed as follows: Accordingly, when the TL LLC resonant converter is utilized behind the power factor correction (PFC) converter, the resonant tank is designed to operate at the resonant frequency for load regulation under a nominal operation. Nonetheless, the link voltage of PFC converter is lower during the hold-up time, but the TL LLC resonant converter requires a higher voltage gain. Thus, the resonant tank is designed to obtain an output voltage under the minimum link voltage, V link_min , and frequency ratio, F min . From Equation (35), the resonant tank can be expressed as follows: where, F min = f sw_min / f r . Furthermore, for assumed values of parameters F min , and L n , the resonant tank of L m , L r , and C r is determined from Equations (39)-(41). For example, if P o = 350 W, n = 5, V link_min = 320 V, L n = 7, and F min = 0.7, the output voltage can be obtained under the minimum link voltage according to the switching and resonant frequency from Figure 16.   Table 1 provides a list of equations for the maximum voltage of a Si-FET and a resonant capacitor between the TL LLC resonant converter and a HB LLC resonant converter. Here, although the TL LLC resonant converter has two additional switches, compared with the HB LLC resonant converter for the same resonant tank condition, the voltage across the main switches is half that of the HB LLC resonant converter.

Comparison of Switch Loss
To elucidate the distinction between the TL LLC resonant converter with Si-FETs and the HB LLC resonant converter with GaN-FETs, their switch losses are compared under the same resonant tank conditions and specifications of main parameters, as shown in Table 2. When the LLC resonant converter operates at the resonant frequency with enough dead time, the main switches can operate with ZVS. Thus, turned-on and output capacitor losses are not considered.
Consequently, the loss sources of FETs deal with the conduction, turned-off, dead time, and gate driver losses. From Table 3, the Si-FETs, with lower rated voltage than the GaN-FETs, have similar  Table 1 provides a list of equations for the maximum voltage of a Si-FET and a resonant capacitor between the TL LLC resonant converter and a HB LLC resonant converter. Here, although the TL LLC resonant converter has two additional switches, compared with the HB LLC resonant converter for the same resonant tank condition, the voltage across the main switches is half that of the HB LLC resonant converter.

Comparison of Switch Loss
To elucidate the distinction between the TL LLC resonant converter with Si-FETs and the HB LLC resonant converter with GaN-FETs, their switch losses are compared under the same resonant tank conditions and specifications of main parameters, as shown in Table 2. When the LLC resonant converter operates at the resonant frequency with enough dead time, the main switches can operate with ZVS. Thus, turned-on and output capacitor losses are not considered. Consequently, the loss sources of FETs deal with the conduction, turned-off, dead time, and gate driver losses. From Table 3, the Si-FETs, with lower rated voltage than the GaN-FETs, have similar drain-source resistance, fall time, and input capacitance as the GaN-FETs. This demonstrates that the proposed converter is applicable to high-performance switches, which has a lower rated voltage compared with switches for the HB LLC resonant converter.  Figure 17 displays the switch losses in both converters under the assumption that the resonant converter operates at the resonant frequency. Figure 17a,b illustrates the specified switch losses of the TL LLC resonant converter and two switches of the HB LLC resonant converter based on Table 3, at the point of switching frequency f sw = 1 MHz. Apparently, as the TL LLC resonant converter has four switches in series, the conduction loss is approximately twice that of the HB LLC resonant converter. Overall, a slight difference of 0.7 W is obtained for the total switch losses of each converter. Figure 17c displays the single-switch losses of the GaN-FET and Si-FET. Although both increase with a higher switching frequency, the single-switch losses of the TL LLC are smaller than those of the HB LLC, mainly due to its reduced turned-off loss and the lower voltage across the switches. Moreover, the proposed TL LLC resonant converter can use the high-performance Si-FETs, which have lower drain-source resistance and source-drain diode forward voltage, owing to the low rated voltage. Therefore, each switch of the TL LLC resonant converter exhibits lower conduction, turned-off, and dead time losses, indicating its ability to reduce the heat sink size of the main switches, as switch losses can be dissipated.
converter operates at the resonant frequency. Figure 17a,b illustrates the specified switch losses of the TL LLC resonant converter and two switches of the HB LLC resonant converter based on Table 3, at the point of switching frequency fsw = 1 MHz. Apparently, as the TL LLC resonant converter has four switches in series, the conduction loss is approximately twice that of the HB LLC resonant converter. Overall, a slight difference of 0.7 W is obtained for the total switch losses of each converter.

Optimal Design through the Loss Analysis
A converter's efficiency is relative based on various components, including the value of resonant tank, switching frequency, and so on. Based on the loss analysis, the highest efficiency for the TL LLC resonant converter can be obtained via optimization of the design components. General loss sources include Si-FET conduction loss and switching loss, copper and core loss of magnetics, and conduction loss of synchronous (SR)-FETs. The parameters used are based on the assumption that the converter operates at a switching frequency in a steady state.

Loss Analysis for Switching Components
The switching components of the TL LLC resonant converter are four Si-FETs in the primary side and two SR-FETs in the secondary side as the output rectifier. For the comparison of power losses, equations for the main parameters can be consulted as provided in Table 4. Table 4. Equations for calculation of switching loss.

Components Symbols Equations
Si-FET I Q_rms Most of the Si-FET loss components for the TL resonant converter are similar to those of a conventional HB LLC resonant converter. Switches for the TL LLC and HB LLC resonant converters operate with ZVS as mentioned in the switch loss comparison; as such, the loss sources of Si-FETs deal with the conduction, turned-off, dead time, and gate driver losses. The conduction loss of Si-FETs can be expressed as follows: where R ds(on) is the drain-source on-state resistance, whereas I Q_rms is the drain-source root mean square (RMS) current of the switch.
The drain-source voltage of the converter and the maximum current of the magnetizing inductance at the turned-off point affect the transition loss of Si-FETs. Moreover, because the drain-source voltage of the converter is half the input voltage, the turn-off loss of each switch is equally reduced. Therefore, the TL LLC resonant converter can reduce the heat sink size. Relatively, the turned-off loss of Si-FETs can be expressed as follows: where V Q_peak is the drain-source voltage, I Lm_peak is the maximum current of the magnetizing inductor, f sw is the switching frequency, and t f is the fall time. The gate driver loss of Si-FETs can be expressed as follows: where C iss is the parasitic input capacitance of SR-FETs, and V drv is the gate driver supply voltage. The dead time loss of Si-FETs can be expressed as follows: where V SD is the source-drain diode forward voltage drop, and t dead is the dead time of Si-FETs. The SR-FETs operate with ZVS and ZCS due to the 0 current at the turned-on and turned-off point. Thus, the loss sources of SR-FETs are the conduction and gate driver losses. The conduction loss of SR-FETs can be expressed as follows: where R SR_ds(on) is the drain-source on-state resistance of SR-FETs, and I do_rms is the drain-source RMS current of SR-FET. The gate driver loss of SR-FETs can be expressed as follows: where C iss is the parasitic input capacitance of SR-FETs, and V drv is the supply voltage of the gate driver.

Loss Analysis for Magnetics
The magnetics of the TL LLC resonant converter are two resonant inductors and two transformers in the primary side. For comparison of the magnetics losses, the RMS and peak current equations for the magnetics of the TL LLC resonant converter are shown in Table 5. Table 5. Equations for the calculation of magnetic loss.

Components Symbols Equations
Resonant inductor I Lr_rms A common technique for calculating loss is employed in the magnetic analysis. The waveforms of the TL LLC resonant converter are first assumed to be sinusoidal. Cores used for magnetics of the TL Energies 2019, 12, 3082 18 of 24 LLC resonant converter are planar cores aimed at minimizing the converter height. The core losses of the inductor and the transformer are determined by the modified Steinmetz equation [20,21]: where k, α, and β are the core loss coefficients, dependent on the core material; B T_pk is the peak flux density of the transformer; and V e_T is the volume of the transformer. The total wire loss of the transformer is expressed as follows: where R pri is the primary winding resistance, I pri_rms is the rms current of the primary wire, R sec is the secondary winding resistance, and I sec_rms is the RMS current of the secondary wire. Inductor core loss is determined by using the modified Steinmetz equation: where k, α, and β are the core loss coefficients, dependent on the core material; B Lr_pk is the peak flux density; and V e_Lr is the volume of the transformer. Inductor wire loss is expressed as follows: where R Lr is the winding resistance of the resonant inductor and I Lr_rms is the rms current of the resonant inductor.

Results of Loss Analysis
To achieve a highly efficient converter, the optimal design based on loss analysis is required to choose a switching frequency and a value of the resonant tank that minimizes power losses. In this respect, the power losses are relative to the switching frequency and L n of the converter.
For the loss analysis, the following parameters are used and the proposed converter is designed to be operated at f r , as the resonant frequency:
Output power: The TL LLC resonant converter operates at f r as the switching frequency; thus, the transformer turn ratio can be obtained using the following equation, where the voltage conversion gain is equal to 1: Using Equation (52), the turn ratio is about 5. Moreover, the proposed converter requires two inductors and two transformers. To reduce its size, a planar transformer and a resonant inductor with 12 layers are selected with the two oz. width for the pattern, relative to the core window area. Also, two resonant tanks, a transformer and an inductor, consist of a single core with winding of the pattern on the outer legs of the core, thereby increasing the power density, as shown in Figure 18.  Figure 20 shows a comparison of the total losses based on Figure 19. A comparison of total loss by Ln and fsw is shown in Figure 20a, mainly describing a slight difference in total losses when fsw > 1 MHz and Ln > 7. Figure 20b described a detailed total loss comparison according to the switching frequency with Ln = 7 and Po = 350 W. Note that when the switching frequency is higher than 1 MHz, the value of the total losses is almost the same value as at 16 W. Consequently, the switching frequency is set to 1 MHz, and Ln to 7. The transformer has five primary turns and one secondary turn, whereas by core saturation, the resonant inductor has five turns. Figure 19a shows the total switch losses, including those of Si-FETs and SR-FETs according to L n (= L m /L n ) and the switching frequency. Here, when L n is higher than 7 with a high switching frequency, there is a slight difference in the switch losses. Figure 19b depicts the total magnetic component losses according to L n and the switching frequency; there is also a slight difference in the switch losses when L n is higher than 7 with a high switching frequency.  Figure 20 shows a comparison of the total losses based on Figure 19. A comparison of total loss by Ln and fsw is shown in Figure 20a, mainly describing a slight difference in total losses when fsw > 1 MHz and Ln > 7. Figure 20b described a detailed total loss comparison according to the switching frequency with Ln = 7 and Po = 350 W. Note that when the switching frequency is higher than 1 MHz, the value of the total losses is almost the same value as at 16 W. Consequently, the switching frequency is set to 1 MHz, and Ln to 7.  Figure 20 shows a comparison of the total losses based on Figure 19. A comparison of total loss by L n and f sw is shown in Figure 20a, mainly describing a slight difference in total losses when f sw > 1 MHz and L n > 7. Figure 20b described a detailed total loss comparison according to the switching frequency with L n = 7 and P o = 350 W. Note that when the switching frequency is higher than 1 MHz, the value of the total losses is almost the same value as at 16 W. Consequently, the switching frequency is set to 1 MHz, and L n to 7.

Experimental Results
A 350 W prototype of the TL LLC resonant converter was created to verify its performance, operational principle, and advantages, as depicted in Figure 21. Table 3 and 6 provide the main parameters and components of the prototype.

Experimental Results
A 350 W prototype of the TL LLC resonant converter was created to verify its performance, operational principle, and advantages, as depicted in Figure 21. Tables 3 and 6 provide the main parameters and components of the prototype.

Experimental Results
A 350 W prototype of the TL LLC resonant converter was created to verify its performance, operational principle, and advantages, as depicted in Figure 21. Table 3 and 6 provide the main parameters and components of the prototype.

Practical Voltage Balancing under Soft Start-Up Conditions
Under soft start-up conditions, the voltage of all capacitors in the TL LLC resonant converter was discharged to 0. The voltage of the balancing capacitor V B was also 0 at the point of start-up, as shown in Figure 22a. After Q 1 and Q 3 were turned on during Mode 1 or Q 2 and Q 4 were turned on during Mode 4, the input voltage was applied to the balancing capacitor and the resonant capacitor, C r3 and C r4 by the Kirchhoff voltage law. At this moment, the sums of the resonant capacitor voltages, V cr1 + Vc r2 and V cr3 + V cr4 , could not be guaranteed to be half of the input voltage. Figure 23a shows the experimental waveforms at V IN = 200 V, which are due to the voltage imbalance. From Figure 22a, the switches could be destroyed at V IN = 390 V due to the voltage imbalance. Air gap of center leg 0.55 mm

Practical Voltage Balancing under Soft Start-Up Conditions
Under soft start-up conditions, the voltage of all capacitors in the TL LLC resonant converter was discharged to 0. The voltage of the balancing capacitor VB was also 0 at the point of start-up, as shown in Figure 22a. After Q1 and Q3 were turned on during Mode 1 or Q2 and Q4 were turned on during Mode 4, the input voltage was applied to the balancing capacitor and the resonant capacitor, Cr3 and Cr4 by the Kirchhoff voltage law. At this moment, the sums of the resonant capacitor voltages, Vcr1 + Vcr2 and Vcr3 + Vcr4, could not be guaranteed to be half of the input voltage. Figure 23a shows the experimental waveforms at VIN = 200 V, which are due to the voltage imbalance. From Figure 22a, the switches could be destroyed at VIN = 390 V due to the voltage imbalance. These problems could be resolved by configuring the current flow path with pre-charging resistors to charge the balancing capacitor before soft start-up. However, as the location of resistors enables the current and included input voltage to assume a steady state, as depicted in Figure 22b, the power consumption can be as large as the standby power. However, if the value of the resistors was too high to reduce the standby power, the pre-charging time would be longer. As such, the resistor value has to be properly selected. With the resistors of 1 MOhm, the voltage across switches can be set to half the input voltage at the point of soft start-up, as shown in Figure 23b. Air gap of center leg 0.55 mm

Practical Voltage Balancing under Soft Start-Up Conditions
Under soft start-up conditions, the voltage of all capacitors in the TL LLC resonant converter was discharged to 0. The voltage of the balancing capacitor VB was also 0 at the point of start-up, as shown in Figure 22a. After Q1 and Q3 were turned on during Mode 1 or Q2 and Q4 were turned on during Mode 4, the input voltage was applied to the balancing capacitor and the resonant capacitor, Cr3 and Cr4 by the Kirchhoff voltage law. At this moment, the sums of the resonant capacitor voltages, Vcr1 + Vcr2 and Vcr3 + Vcr4, could not be guaranteed to be half of the input voltage. Figure 23a shows the experimental waveforms at VIN = 200 V, which are due to the voltage imbalance. From Figure 22a, the switches could be destroyed at VIN = 390 V due to the voltage imbalance. These problems could be resolved by configuring the current flow path with pre-charging resistors to charge the balancing capacitor before soft start-up. However, as the location of resistors enables the current and included input voltage to assume a steady state, as depicted in Figure 22b, the power consumption can be as large as the standby power. However, if the value of the resistors was too high to reduce the standby power, the pre-charging time would be longer. As such, the resistor value has to be properly selected. With the resistors of 1 MOhm, the voltage across switches can be set to half the input voltage at the point of soft start-up, as shown in Figure 23b. These problems could be resolved by configuring the current flow path with pre-charging resistors to charge the balancing capacitor before soft start-up. However, as the location of resistors enables the current and included input voltage to assume a steady state, as depicted in Figure 22b, the power consumption can be as large as the standby power. However, if the value of the resistors was too high to reduce the standby power, the pre-charging time would be longer. As such, the resistor value has to be properly selected. With the resistors of 1 MOhm, the voltage across switches can be set to half the input voltage at the point of soft start-up, as shown in Figure 23b.

Waveforms
The experimental balancing waveforms of each resonant component with P O = 350 W and normalized L = 30 in the steady state are shown in Figure 24. In Figure 24a, the voltage of the balancing capacitor V CB and the sum of the resonant capacitor voltage V Cr1 + V Cr2 and V Cr3 + V Cr4 maintained Energies 2019, 12, 3082 22 of 24 the voltage balance. Moreover, the resonant currents i Lr1 and i Lr2 exhibited the same waveforms and maintained the current balance for each resonant tank, as depicted in Figure 24b. Figure 24 shows the main experimental waveforms with output power from P O = 0 W to 350 W, with drain-to-source voltage V Q_ds , gate-to-source voltage V Q_gs , and resonant current i Lr in Figure 25a-d. All switches were turned on with ZVS under every load condition, and the voltage stress was set to half the input voltage. The experimental balancing waveforms of each resonant component with PO = 350 W and normalized L = 30 in the steady state are shown in Figure 24. In Figure 24a, the voltage of the balancing capacitor VCB and the sum of the resonant capacitor voltage VCr1 + VCr2 and VCr3 + VCr4 maintained the voltage balance. Moreover, the resonant currents iLr1 and iLr2 exhibited the same waveforms and maintained the current balance for each resonant tank, as depicted in Figure 24b. Figure 24 shows the main experimental waveforms with output power from PO = 0 W to 350 W, with drain-to-source voltage VQ_ds, gate-to-source voltage VQ_gs, and resonant current iLr in Figure 25a-d. All switches were turned on with ZVS under every load condition, and the voltage stress was set to half the input voltage.  Figure 26a shows efficiency comparison curves between the theoretical and experimental results measured by YOKOGAWA : Tokyo, Japan, WT1804E under different load conditions. The efficiency curves were the same, with slight differences. The maximum efficiency was 95.43% at an 80% load condition. Figure 26b is an efficiency comparison between TL LLC with Si-FETs and HB LLC with GaN-FETs. From Figure 26b, we see that, while the proposed TL LLC resonant converter has the higher efficiency under the light load condition, the difference in efficiency is further reduced as the output power increases. Because the turn-off loss forms a large portion of the total loss at a light load, the proposed TL LLC resonant converter has less total loss due to low voltage across the switches. The experimental balancing waveforms of each resonant component with PO = 350 W and normalized L = 30 in the steady state are shown in Figure 24. In Figure 24a, the voltage of the balancing capacitor VCB and the sum of the resonant capacitor voltage VCr1 + VCr2 and VCr3 + VCr4 maintained the voltage balance. Moreover, the resonant currents iLr1 and iLr2 exhibited the same waveforms and maintained the current balance for each resonant tank, as depicted in Figure 24b. Figure 24 shows the main experimental waveforms with output power from PO = 0 W to 350 W, with drain-to-source voltage VQ_ds, gate-to-source voltage VQ_gs, and resonant current iLr in Figure 25a-d. All switches were turned on with ZVS under every load condition, and the voltage stress was set to half the input voltage.  Figure 26a shows efficiency comparison curves between the theoretical and experimental results measured by YOKOGAWA : Tokyo, Japan, WT1804E under different load conditions. The efficiency curves were the same, with slight differences. The maximum efficiency was 95.43% at an 80% load condition. Figure 26b is an efficiency comparison between TL LLC with Si-FETs and HB LLC with GaN-FETs. From Figure 26b, we see that, while the proposed TL LLC resonant converter has the higher efficiency under the light load condition, the difference in efficiency is further reduced as the output power increases. Because the turn-off loss forms a large portion of the total loss at a light load, the proposed TL LLC resonant converter has less total loss due to low voltage across the switches.  Figure 26a shows efficiency comparison curves between the theoretical and experimental results measured by YOKOGAWA : Tokyo, Japan, WT1804E under different load conditions. The efficiency curves were the same, with slight differences. The maximum efficiency was 95.43% at an 80% load condition. Figure 26b is an efficiency comparison between TL LLC with Si-FETs and HB LLC with GaN-FETs. From Figure 26b, we see that, while the proposed TL LLC resonant converter has the higher efficiency under the light load condition, the difference in efficiency is further reduced as the output power increases. Because the turn-off loss forms a large portion of the total loss at a light load, the proposed TL LLC resonant converter has less total loss due to low voltage across the switches. However, as the conduction loss gets larger as the output power increases, the difference in efficiency is further reduced because the drain-source on-state resistance of GaN-FETs is smaller than that of the proposed converter. As a result of the efficiency comparison, although the proposed TL LLC resonant converter uses relatively low-cost Si-FET switches, it shows superior performance to the conventional HB LLC resonant converter using the high-performance and high-cost GaN-FETs. However, as the conduction loss gets larger as the output power increases, the difference in efficiency is further reduced because the drain-source on-state resistance of GaN-FETs is smaller than that of the proposed converter. As a result of the efficiency comparison, although the proposed TL LLC resonant converter uses relatively low-cost Si-FET switches, it shows superior performance to the conventional HB LLC resonant converter using the high-performance and high-cost GaN-FETs.

Conclusions
A proposed Si-FET-based high switching frequency three-level LLC resonant converter was presented herein, with a discussion of its operational principle, loss analysis, and optimal design consideration based on loss analysis to those of another converter. The results demonstrated the ability of the proposed converter to achieve voltage balancing without additional circuits or control strategy and to ensure that the voltage across the main switches is set to half of the input voltage. Moreover, it was shown that the voltage stress of the main switches could be further reduced via the application of a modular system. Based on analysis and design considerations, the converter's operating frequency was over 1 MHz, with a high efficiency of 95.33% under a full load (350 W).

Conclusions
A proposed Si-FET-based high switching frequency three-level LLC resonant converter was presented herein, with a discussion of its operational principle, loss analysis, and optimal design consideration based on loss analysis to those of another converter. The results demonstrated the ability of the proposed converter to achieve voltage balancing without additional circuits or control strategy and to ensure that the voltage across the main switches is set to half of the input voltage. Moreover, it was shown that the voltage stress of the main switches could be further reduced via the application of a modular system. Based on analysis and design considerations, the converter's operating frequency was over 1 MHz, with a high efficiency of 95.33% under a full load (350 W).