Tolerance Considerations for MHMIC Manufacturing Process at Millimeter-Wave Band

This paper investigates the manufacturing uncertainties at a 60 GHz millimeter-wave band for the monolithic hybrid microwave integrated circuits (MHMIC) fabrication process. It specifically deals with the implementation tolerances of thin-film gold microstrip transmission lines, titanium oxide thin-layer resistors, microstrip quarter-wavelength radial stubs, and active device implementation using the gold-bonding ribbons. The impacts of these manufacturing tolerances are assessed and experimentally quantified through prototyped MHMIC circuits. This allows us, on one hand, to identify the acceptable amount of dimensional variation enabling reasonable performances. On the other hand, it aims to establish a relationship between the manufacturing tolerances and the circuit parameters to provide more flexibility for the tolerance compensation and accuracy enhancement of the MHMIC fabrication processes.


Introduction
Since its introduction in 1960, the technology of Miniature Hybrid Microwave Integrated Circuits (MHMICs) has constantly advanced, and it has widely been used in many RF/microwave applications [1][2][3][4][5].It has been particularly useful for small-scale production and prototyping purposes, thus resulting in a significant reduction in production costs and development times.MHMICs are regarded as a matured technology, integrating various transmission lines (coplanar waveguide, microstrip, slot-lines) [6,7], as well as a large number of passive components typically found in gallium arsenide monolithic microwave integrated circuits (GaAs MMICs).Those passive components include, but are not limited to, thin-film resistors, air-bridges, metal insulator metal (MIM) capacitors, and planar inductors [8,9].They are mostly implemented on less expensive and low-loss thin-film alumina ceramic substrates [9].However, active devices or elements are connected to the circuits separately with wire or ribbon bonds since they are not formed in or on the substrate [10][11][12][13].
Although MHMIC technology today offers well-proven performance at frequencies above 60 GHz due to its high-density integration potentials [2], it remains relatively sensitive to manufacturing tolerances.Several MHMICs have been reported in the literature in recent years, covering a wide range of frequencies and applications [1,14,15].The vast majority of proposed works were design-oriented.and there is no specific study that has dealt with the manufacturing process tolerance and its limits.That kind of study is, in fact, required to enable the quantification of process inaccuracies, which allows MHMICs' performance improvement.For instance, the implementation of MHMIC termination loads Sensors 2024, 24, 2486 2 of 10 and isolation resistors is sensitive to fabrication tolerances, resulting in phase and amplitude imbalance, especially when it comes to multiport circuit design.Those termination loads are typically integrated using a 100 Ω per square titanium oxide thin layer, along with quarter-wavelength radial stubs to enable proper impedance matching while avoiding metallization via holes.
In this paper, the sensitivity to the tolerances of the MHMIC manufacturing process is investigated at the millimeter-wave frequency band.The tolerance analysis specifically addresses the implementation tolerances of thin-film gold microstrip transmission lines, titanium oxide thin-layer resistors, microstrip quarter-wavelength radial stubs, and active device implementation using the gold-bonding ribbons.It turns out that manufacturing tolerance optimization provides enhanced performances in terms of amplitude and phase balances.Therefore, tolerance quantification enables fabrication process improvement, as well as the establishment of a relationship between the manufacturing tolerances and the design parameters.This study aims to provide more flexibility for the tolerance compensation and accuracy enhancement of the MHMIC fabrication process.

On-Wafer Calibration Techniques and Standards
To achieve an accurate assessment of manufacturing uncertainties in the designed MHMIC circuits as part of this study, it is necessary to select an appropriate calibration technique.This task leads to effectively correcting on-wafer measurement errors due to the non-ideal characteristics of the cables and probes at the operating millimeter-wave frequency band.The most commonly used calibration techniques for millimeter-wave on-wafer measurements are LRM (Line-Reflect-Match), LLRM (Line-Line-Reflect-Match) and TRL (Thru-Reflect-Line).LRM and LLRM calibrations require match standards, which are difficult to achieve accurately at millimeter-wave frequencies.We then chose the TRL calibration technique by implementing our calibration kit with the circuits to be characterized instead of using an expensive commercial impedance standard substrate (ISS).
To connect the Ground Signal Ground (GSG) 150 µm probes to the TRL standard ports, a microstrip line transition to a coplanar line was designed and implemented at each standard access.In the adopted approach, an RF short circuit was made by a quarter-wave sector, avoiding via holes metallization that is difficult to repeatably achieve at millimeterwave frequencies [16].
The photograph of the on-wafer two-port circuit characterization, the microphotographs of the TRL calibration standards, and the GSG 150 µm probe on the access of the microstrip line/coplanar line transition are, respectively, illustrated in Figure 1a-c, shown above.The TRL calibration kit consists of a line thru (T), two open circuits (reflect R), and a line (L).Due to the fragility of the thin layer of gold (1 µm), multiple identical standards were manufactured on the same 2.54 cm by 2.54 cm alumina ceramic substrate (ε r = 9.5, h = 127 µm) to ensure the repeatability and success of calibrations and measurements.
The calibration was carried out by connecting the Cascade Microtech Infinity GSG 150 µm probe to the three standards, starting with a direct connection (Thru) between the two ports that had the property of fixing the reference plane in the middle of its length.Then, we connected an open-circuit line (Reflect), and we ended with a transmission line (Line).The length of the latter limits the measurement frequency band, and it is estimated by the relationship in (1).This is relative to the "Thru" line, which is considered to have zero length.It is noteworthy that our measurement setup is designed to operate in the frequency band of 60 to 90 GHz.Under those considerations, the corresponding wavelengths are, respectively, λ g1 = 1.9 mm and λ g2 = 1.23 mm [2].
where ∆L is the length difference between the "Line" and "Thru" standards.In that respect, we have selected lengths of 1.305 mm for the "Thru" line, and a length of 1.798 mm for wave sector, avoiding via holes metallization that is difficult to repeatably achieve at millimeter-wave frequencies [16].
The photograph of the on-wafer two-port circuit characterization, the microphotographs of the TRL calibration standards, and the GSG 150 µm probe on the access of the microstrip line/coplanar line transition are, respectively, illustrated in Figure 1a-c, shown above.The TRL calibration kit consists of a line thru (T), two open circuits (reflect R), and a line (L).Due to the fragility of the thin layer of gold (1 µm), multiple identical standards were manufactured on the same 2.54 cm by 2.54 cm alumina ceramic substrate (εr = 9.5, h = 127 µm) to ensure the repeatability and success of calibrations and measurements.

Manufacturing Tolerance Analysis of the Gold Thin-Film Microstrip Line
The loss assessment of the gold thin-film microstrip line is an important operation to optimize the manufacturing process of MHMIC circuits at millimeter-wave bands.It should be noted that most substrate manufacturers do not provide information covering the millimeter-wave range.Quantifying losses over the operating frequency band between 60 and 70 GHz is then performed by measuring the insertion losses of two gold thin-film microstrip lines with, respectively, lengths of 2.486 mm and 2.914 mm.
Figure 2a,b show, respectively, the microphotographs of both manufactured gold thin-film microstrip lines and the additional attenuation in the 60 GHz to 70 GHz frequency range.The considered frequency range starts from 60 GHz instead of 57 GHz (the starting frequency of the unlicensed 60 GHz frequency spectrum) due to the capability of measurement equipment (Keysight Technologies E8362B (VNA) connected with E-band extension modules, operating in the 60-90 GHz band).The calibration was carried out by connecting the Cascade Microtech Infinity GSG 150 µm probe to the three standards, starting with a direct connection (Thru) between th two ports that had the property of fixing the reference plane in the middle of its length Then, we connected an open-circuit line (Reflect), and we ended with a transmission lin (Line).The length of the latter limits the measurement frequency band, and it is estimated by the relationship in (1).This is relative to the "Thru" line, which is considered to hav zero length.It is noteworthy that our measurement setup is designed to operate in th frequency band of 60 to 90 GHz.Under those considerations, the corresponding wave lengths are, respectively, λg1 = 1.9 mm and λg2 = 1.23 mm [2].
where ΔL is the length difference between the "Line" and "Thru" standards.In that re spect, we have selected lengths of 1.305 mm for the "Thru" line, and a length of 1.798 mm for the "Line" line.Then, ΔL is equal to 0.493 mm, ranging between λg1/4 = 0.475 mm and λg2/2 = 0.615 mm.Therefore, the "Thru" and "Line" line lengths enable it to cover the entir frequency band, from 60 to 90 GHz, without a phase ambiguity problem.

Manufacturing Tolerance Analysis of the Gold Thin-Film Microstrip Line
The loss assessment of the gold thin-film microstrip line is an important operation to optimize the manufacturing process of MHMIC circuits at millimeter-wave bands.I should be noted that most substrate manufacturers do not provide information covering the millimeter-wave range.Quantifying losses over the operating frequency band be tween 60 and 70 GHz is then performed by measuring the insertion losses of two gold thin-film microstrip lines with, respectively, lengths of 2.486 mm and 2.914 mm.The obtained results illustrated in Figure 2b show that the two curves are almos identical.The conductor losses of the 1 µm gold layer undergoing metallization do no exceed 0.05 dB/mm in the considered frequency range, from 60 GHz to 70 GHz.Given that the physical length of a quarter-wave line at the central frequency of 61 GHz is abou The obtained results illustrated in Figure 2b show that the two curves are almost identical.The conductor losses of the 1 µm gold layer undergoing metallization do not exceed 0.05 dB/mm in the considered frequency range, from 60 GHz to 70 GHz.Given that the physical length of a quarter-wave line at the central frequency of 61 GHz is about Sensors 2024, 24, 2486 4 of 10 0.46 mm, we may conclude that the loss level is very low.These results demonstrate the success of the design and calibration, as well as the appropriate choice of substrate material.

Manufacturing Tolerance Analysis of the Thin-Film Resistive Layer
To experimentally assess the manufacturing tolerances of a thin-layer titanium oxide resistor (100 Ω per square), an MHMIC 50 Ω termination is implemented on an alumina ceramic substrate (ε r = 9.5, h = 127 µm).This 50 Ω termination is a part of the circuit's layout presented in Figure 1a, which includes some typical MHMICs such as 90-degree hybrid couplers, Wilkinson power dividers, and several identical TRL standard calibration kits.
Figure 3a,b show an implemented MHMIC 50 Ω termination on an alumina ceramic substrate, and the results of on-wafer input impedance measurement (real and imaginary parts) versus frequency.As can be observed, the value of the real part of the input impedance ranges from 48.5 Ω to 49.8 Ω over the considered frequency range of 60 GHz to 70 GHz.However, the value of the imaginary part is very low around 61 GHz (the resonant frequency), and it increases with frequency to reach a value of 10 at 70 GHz.This increase in the value of the imaginary part is due to the presence of significant parasitic inductive effects at high millimeter-wave frequencies.
0.46 mm, we may conclude that the loss level is very low.These results demonstrate the success of the design and calibration, as well as the appropriate choice of substrate material.

Manufacturing Tolerance Analysis of the Thin-Film Resistive Layer
To experimentally assess the manufacturing tolerances of a thin-layer titanium oxide resistor (100 Ω per square), an MHMIC 50 Ω termination is implemented on an alumina ceramic substrate (εr = 9.5, h = 127 µm).This 50 Ω termination is a part of the circuit's layout presented in Figure 1a, which includes some typical MHMICs such as 90-degree hybrid couplers, Wilkinson power dividers, and several identical TRL standard calibration kits.
Figure 3a,b show an implemented MHMIC 50 Ω termination on an alumina ceramic substrate, and the results of on-wafer input impedance measurement (real and imaginary parts) versus frequency.As can be observed, the value of the real part of the input impedance ranges from 48.5 Ω to 49.8 Ω over the considered frequency range of 60 GHz to 70 GHz.However, the value of the imaginary part is very low around 61 GHz (the resonant frequency), and it increases with frequency to reach a value of 10 at 70 GHz.This increase in the value of the imaginary part is due to the presence of significant parasitic inductive effects at high millimeter-wave frequencies.To estimate the maximum error during the implementation of a 100 Ω per square titanium oxide-based thin-layer resistor, we consider the fractional uncertainty of the area (A) in the 50 Ω rectangular resistor layer in Figure 3a.By considering that x and y are, respectively, the width and length of the titanium oxide rectangular covered area (A), the fractional uncertainty can then be expressed as follows [17,18]: where the area of the rectangle is described by the relation A = x•y, and the initial area is represented by A0.Under those considerations, the final area is obtained by A = A0 ± ∆A.
In the case of a 100 Ω per square resistor, Equation (2) will be further simplified since y = x = W, where W is the microstrip line width.Then, Equation (2) becomes It should be noted that the values of A0 and W are, respectively, set at 0.0161 mm 2 and 0.127 mm, taking account into the employed alumina substrate parameters.To estimate the maximum error during the implementation of a 100 Ω per square titanium oxide-based thin-layer resistor, we consider the fractional uncertainty of the area (A) in the 50 Ω rectangular resistor layer in Figure 3a.By considering that x and y are, respectively, the width and length of the titanium oxide rectangular covered area (A), the fractional uncertainty can then be expressed as follows [17,18]: where the area of the rectangle is described by the relation A = x•y, and the initial area is represented by A 0 .Under those considerations, the final area is obtained by A = A 0 ± ∆A.
In the case of a 100 Ω per square resistor, Equation (2) will be further simplified since y = x = W, where W is the microstrip line width.Then, Equation (2) becomes It should be noted that the values of A 0 and W are, respectively, set at 0.0161 mm 2 and 0.127 mm, taking account into the employed alumina substrate parameters.
As can be seen from the Formula (3), the tolerance of the metallization process creates a variation ∆W in the microstrip line width, which, in turn, affects the tolerance level of a thin-layer resistor area ∆A.Therefore, by optimizing the accuracy of the gold metallization process, we may achieve a reduced error in the microstrip line width and, accordingly, enhance the implementation accuracy of a 100 Ω per square thin-layer resistor.
The relationship in (3) allows for plotting curves of the thin-layer resistor area variation versus the relative deviation of the microstrip line width, as shown in Figure 4.The thinlayer resistor area variation increases linearly with the relative variation in microstrip line width.In general, the total thin-layer resistor area change does not exceed 0.0005 mm 2 for a maximum relative variation in a microstrip line width of 15%.
Sensors 2024, 24, x FOR PEER REVIEW 5 of 1 As can be seen from the Formula (3), the tolerance of the metallization process create a variation ∆W in the microstrip line width, which, in turn, affects the tolerance level of thin-layer resistor area ∆A.Therefore, by optimizing the accuracy of the gold metallization process, we may achieve a reduced error in the microstrip line width and, accordingly enhance the implementation accuracy of a 100 Ω per square thin-layer resistor.
The relationship in (3) allows for plotting curves of the thin-layer resistor area varia tion versus the relative deviation of the microstrip line width, as shown in Figure 4. Th thin-layer resistor area variation increases linearly with the relative variation in microstrip line width.In general, the total thin-layer resistor area change does not exceed 0.0005 mm for a maximum relative variation in a microstrip line width of 15%.

Manufacturing Tolerance Analysis of a Quarter-Wavelength Radial Stub
Radial stubs are widely used in various microwave circuits such as filters, matching networks, biasing lines, and even grounding RF circuits.They provide a low impedanc level at well-specified insertion points in a wide frequency band, contrary to the conven tional straight stubs, which exhibit a reduced bandwidth and an increase in circuit siz [19][20][21][22][23][24].Now, we examine the accuracy improvement in the gold-metalized microstrip line that allows for manufacturing error reduction in the millimeter-wave microstrip quar ter-wavelength radial stub.This radial stub technique enables performance optimization of the millimeter-wave grounding while avoiding via holes metallization [25][26][27].
Figure 5a,b show the quarter-wavelength radial stub configuration including geo metrical parameters and the microphotograph of the prototyped radial stub.The equiva lent circuit of the quarter-wavelength radial stub with and without a 50 Ω termination i shown in Figure 5c.As can be observed, the radial stub with a 50 Ω load can be modelled by a series combination of a 50 Ω resistor, an inductor Lrs, and a capacitor Crs, whereas th radial stub without a 50 Ω load is equivalent to an inductor Lrs and capacitor Crs in series According to Figure 5a, the geometrical parameters of the quarter-wavelength radia stub are as follows: P is the penetration depth, W is the microstrip width, and θ is th angle subtended by the stub, which is limited to a range of 10° ≤ θ ≤ 170°.For the proposed radial stub design, the values of those parameters have been set to P = 63.5 µm, W = 12 µm, and θ = 90°.

Manufacturing Tolerance Analysis of a Quarter-Wavelength Radial Stub
Radial stubs are widely used in various microwave circuits such as filters, matching networks, biasing lines, and even grounding RF circuits.They provide a low impedance level at well-specified insertion points in a wide frequency band, contrary to the conventional straight stubs, which exhibit a reduced bandwidth and an increase in circuit size [19][20][21][22][23][24].Now, we examine the accuracy improvement in the gold-metalized microstrip line that allows for manufacturing error reduction in the millimeter-wave microstrip quarter-wavelength radial stub.This radial stub technique enables performance optimization of the millimeter-wave grounding while avoiding via holes metallization [25][26][27].
Figure 5a,b show the quarter-wavelength radial stub configuration including geometrical parameters and the microphotograph of the prototyped radial stub.The equivalent circuit of the quarter-wavelength radial stub with and without a 50 Ω termination is shown in Figure 5c.As can be observed, the radial stub with a 50 Ω load can be modelled by a series combination of a 50 Ω resistor, an inductor L rs , and a capacitor C rs , whereas the radial stub without a 50 Ω load is equivalent to an inductor L rs and capacitor C rs in series.
According to Figure 5a, the geometrical parameters of the quarter-wavelength radial stub are as follows: P is the penetration depth, W is the microstrip width, and θ is the angle subtended by the stub, which is limited to a range of 10  The relative variation in the angle subtended by the stub and relative variation in the microstrip line width is governed by Equation ( 4) [28,29].
where W0 is the manufactured microstrip line width (W0 = 1.27 × 10 −4 m) It can be noted that the relative variation in the angle subtended by the stub could be optimized by minimizing the relative deviation in the microstrip line width.Using this equation, we can, therefore, plot the curves characterizing the relative variation in the angle as a function of the relative deviation of the microstrip line width, as shown in Figure 6 below.As a linear curve with a positive coefficient, it shows that for a maximum variation of 4% in the microstrip line width, the relative variation in the angle does not exceed 2.5%.In order to verify the performances of short-circuit and 50 Ohm impedance matching for the designed quarter-wavelength sector, along with the 50 Ohm termination, a Smith Chart is employed to plot both the simulated and the measured results over the frequency range extending from 60 GHz to 65 GHz, as shown in Figure 7.The relative variation in the angle subtended by the stub and relative variation in the microstrip line width is governed by Equation ( 4) [28,29].
It can be noted that the relative variation in the angle subtended by the stub could be optimized by minimizing the relative deviation in the microstrip line width.Using this equation, we can, therefore, plot the curves characterizing the relative variation in the angle as a function of the relative deviation of the microstrip line width, as shown in Figure 6 below.As a linear curve with a positive coefficient, it shows that for a maximum variation of 4% in the microstrip line width, the relative variation in the angle does not exceed 2.5%.The relative variation in the angle subtended by the stub and relative variation in th microstrip line width is governed by Equation (4) [28,29].
where W0 is the manufactured microstrip line width (W0 = 1.27 × 10 −4 m) It can be noted that the relative variation in the angle subtended by the stub could b optimized by minimizing the relative deviation in the microstrip line width.Using thi equation, we can, therefore, plot the curves characterizing the relative variation in the an gle as a function of the relative deviation of the microstrip line width, as shown in Figur 6 below.As a linear curve with a positive coefficient, it shows that for a maximum varia tion of 4% in the microstrip line width, the relative variation in the angle does not excee 2.5%.In order to verify the performances of short-circuit and 50 Ohm impedance matchin for the designed quarter-wavelength sector, along with the 50 Ohm termination, a Smit Chart is employed to plot both the simulated and the measured results over the frequenc range extending from 60 GHz to 65 GHz, as shown in Figure 7.In order to verify the performances of short-circuit and 50 Ohm impedance matching for the designed quarter-wavelength sector, along with the 50 Ohm termination, a Smith Chart is employed to plot both the simulated and the measured results over the frequency range extending from 60 GHz to 65 GHz, as shown in Figure 7. Mathematically, an ideal short circuit has typically zero impedance (zero resistanc and reactance).Nonetheless, in reality, it is impossible to achieve that; the resistive part i usually so small and is approximated by zero Ohms.The measured and simulated shor circuit impedances at frequencies ranging from 60 GHz to 65 GHz are plotted on the le of the chart, at the intersection of the resistance and the reactance axes.However, by usin the quarter-wave transformer, we can perform matching to a resistive load impedance o 50 Ohms, corresponding to the normalized load impedance of 1 at the center of Smit Chart.It should be noted that our design is optimized to operate around 61 GHz to enabl experimental characterization with the available measurement equipment in our RF la boratory at INRS-EMT.As can be observed, a slight shift toward the inductive region oc curred at the center.This parasitic inductive effect results from the microstrip transmis sion line acting as a series inductor.Overall, a reasonable agreement between measure ments and simulations is achieved.

Manufacturing Tolerance Analysis of the Gold-Bonding Ribbon
At millimeter-wave frequencies, the interconnection between the chip devices an the RF circuits using the ribbon bonding technique has been identified as one of the ke challenges due to the discontinuity introduced by the bond ribbon that can significantl affect the performance of the entire transceiver at the millimeter-wave frequency band However, the ribbon bonding technique remains a very attractive solution in the electroni packaging domain since it is robust and cost-effective.Several studies on the transmissio performance of ribbon bonding interconnection have been reported for microstrip an coplanar configurations.The reported studies specify that a bond ribbon could form series inductor, contributing to drastically increasing losses, especially when the fre quency or the bond ribbon length increases.For that reason, many efforts have focuse on reducing the length of the bond ribbon, as well as reducing the chip-to-package gap t improve the interconnectivity performance at millimeter-wave frequencies [10][11][12].
Figure 8a illustrates the ribbon bonding configuration of a highly integrate TGA4600 low-noise amplifier (LNA) chip with a 16-element antenna array in a 60 GH MHMIC six-port RF front-end receiver.In this configuration, a 0.127 mm × 1 mm ribbo bond is employed to ensure an RF signal connection between the antenna array and th LNA RFin, as well as between the LNA RFout and the six-port down converter input.Mathematically, an ideal short circuit has typically zero impedance (zero resistance and reactance).Nonetheless, in reality, it is impossible to achieve that; the resistive part is usually so small and is approximated by zero Ohms.The measured and simulated short-circuit impedances at frequencies ranging from 60 GHz to 65 GHz are plotted on the left of the chart, at the intersection of the resistance and the reactance axes.However, by using the quarter-wave transformer, we can perform matching to a resistive load impedance of 50 Ohms, corresponding to the normalized load impedance of 1 at the center of Smith Chart.It should be noted that our design is optimized to operate around 61 GHz to enable experimental characterization with the available measurement equipment in our RF laboratory at INRS-EMT.As can be observed, a slight shift toward the inductive region occurred at the center.This parasitic inductive effect results from the microstrip transmission line acting as a series inductor.Overall, a reasonable agreement between measurements and simulations is achieved.

Manufacturing Tolerance Analysis of the Gold-Bonding Ribbon
At millimeter-wave frequencies, the interconnection between the chip devices and the RF circuits using the ribbon bonding technique has been identified as one of the key challenges due to the discontinuity introduced by the bond ribbon that can significantly affect the performance of the entire transceiver at the millimeter-wave frequency band.However, the ribbon bonding technique remains a very attractive solution in the electronic packaging domain since it is robust and cost-effective.Several studies on the transmission performance of ribbon bonding interconnection have been reported for microstrip and coplanar configurations.The reported studies specify that a bond ribbon could form a series inductor, contributing to drastically increasing losses, especially when the frequency or the bond ribbon length increases.For that reason, many efforts have focused on reducing the length of the bond ribbon, as well as reducing the chip-to-package gap to improve the interconnectivity performance at millimeter-wave frequencies [10][11][12].
Figure 8a illustrates the ribbon bonding configuration of a highly integrated TGA4600 low-noise amplifier (LNA) chip with a 16-element antenna array in a 60 GHz MHMIC six-port RF front-end receiver.In this configuration, a 0.127 mm × 1 mm ribbon bond is employed to ensure an RF signal connection between the antenna array and the LNA RF in , as well as between the LNA RF out and the six-port down converter input.Figure 9 shows transmission loss as a function of frequency for a 1 mm ribbon bon length in different lateral deviation scenarios during ribbon bond implementation.It ca be seen from the figure that in the case of a perfect implementation (∆WR = 0%), the losse start from 0.6 dB/mm at 60 GHz to achieve a maximum of about 1 dB/mm around 70 GHz However, a horizontal deviation of 5% in ribbon bond implementation generates extr losses of about 0.1 dB at 60 GHz and 0.35 dB at 70 GHz compared to the perfect scenari in Figure 8c.Overall, the transmission losses are more severe at high frequencies and sig nificantly increase with the deviation tolerance to exhibit 1.15 dB/mm at 60 GHz and 2. dB/mm for a tolerance level of ∆WR = 15%.Figure 9 shows transmission loss as a function of frequency for a 1 mm ribbon bond length in different lateral deviation scenarios during ribbon bond implementation.It can be seen from the figure that in the case of a perfect implementation (∆W R = 0%), the losses start from 0.6 dB/mm at 60 GHz to achieve a maximum of about 1 dB/mm around 70 GHz.However, a horizontal deviation of 5% in ribbon bond implementation generates extra losses of about 0.1 dB at 60 GHz and 0.35 dB at 70 GHz compared to the perfect scenario in Figure 8c.Overall, the transmission losses are more severe at high frequencies and significantly increase with the deviation tolerance to exhibit 1.15 dB/mm at 60 GHz and 2.4 dB/mm for a tolerance level of ∆W R = 15%.

Conclusions
The implementation tolerances are critical parameters affecting the change in isolation, amplitude and phase balance for millimeter-wave MHMIC circuits.In this paper, a tolerance analysis is performed to investigate the sensitivity of the MHMIC fabrication process to manufacturing tolerances over the unlicensed 60 GHz Industrial-Scientific-Medical (ISM) band.The results have demonstrated that the MHMIC fabrication technique enables reasonable tolerances at millimeter-waves for passive MHIMICs.However, further optimization of these tolerances is feasible by optimizing the implementation accuracy of thin-film gold microstrip transmission lines.This study provides key inputs for the improvement of the MHMIC manufacturing process and enables designers to consider the critical parameters affecting the performances of the designed circuits at millimeter-wave frequencies.

Figure 1 .
Figure 1.On-wafer two-port circuit characterization in (a), the microphotographs of the manufactured TRL calibration standards in (b), and the GSG 150 µm Infinity probe on the access of the microstrip line/coplanar line transition in (c).

Figure 1 .
Figure 1.On-wafer two-port circuit characterization in (a), the microphotographs of the manufac tured TRL calibration standards in (b), and the GSG 150 µm Infinity probe on the access of the mi crostrip line/coplanar line transition in (c).
Figure 2a and b show, respectively, the microphotographs of both manufactured gold thin-film microstrip lines and the additional attenuation in the 60 GHz to 70 GHz fre quency range.The considered frequency range starts from 60 GHz instead of 57 GHz (th starting frequency of the unlicensed 60 GHz frequency spectrum) due to the capability o measurement equipment (Keysight Technologies E8362B (VNA) connected with E-band extension modules, operating in the 60-90 GHz band).

Figure 2 .
Figure 2. Microphotographs of both manufactured gold thin-film microstrip lines in (a) and the ad ditional attenuation in the 60 GHz to 70 GHz frequency range in (b).

Figure 2 .
Figure 2. Microphotographs of both manufactured gold thin-film microstrip lines in (a) and the additional attenuation in the 60 GHz to 70 GHz frequency range in (b).

Figure 3 .
Figure 3. Microphotographs of the implemented MHMIC 50 Ω termination on an alumina substrate in (a) and the additional attenuation in the 60 GHz to 70 GHz frequency range in (b).

Figure 3 .
Figure 3. Microphotographs of the implemented MHMIC 50 Ω termination on an alumina substrate in (a) and the additional attenuation in the 60 GHz to 70 GHz frequency range in (b).

Figure 4 .
Figure 4.The thin-layer resistor area variation versus the relative deviation of microstrip line width

Figure 4 .
Figure 4.The thin-layer resistor area variation versus the relative deviation of microstrip line width.

10 Figure 5 .
Figure 5. Millimeter-wave microstrip quarter-wavelength radial stub: (a) a quarter-wavelength radial stub with geometric parameters, (b) a microphotograph of the implemented prototype, and (c) the equivalent circuits with and without a 50 Ω load.

Figure 6 .
Figure 6.Relative variation in angle subtended by stub versus relative variation in microstrip line width for εr = 9.5 and h = 127 µm.

Figure 5 .
Figure 5. Millimeter-wave microstrip quarter-wavelength radial stub: (a) a quarter-wavelength radial stub with geometric parameters, (b) a microphotograph of the implemented prototype, and (c) the equivalent circuits with and without a 50 Ω load.

Sensors 2024 , 1 Figure 5 .
Figure 5. Millimeter-wave microstrip quarter-wavelength radial stub: (a) a quarter-wavelength ra dial stub with geometric parameters, (b) a microphotograph of the implemented prototype, and (c the equivalent circuits with and without a 50 Ω load.

Figure 6 .
Figure 6.Relative variation in angle subtended by stub versus relative variation in microstrip lin width for εr = 9.5 and h = 127 µm.

Figure 6 .
Figure 6.Relative variation in angle subtended by stub versus relative variation in microstrip line width for ε r = 9.5 and h = 127 µm.

Figure 7 .
Figure 7. Measured and simulated impedance values on the Smith chart for the designed qua ter−wavelength sector with and without the 50 Ω termination resistor for the 60 GHz to 65 GH frequency range.

Figure 7 .
Figure 7. Measured and simulated impedance values on the Smith chart for the designed quarter−wavelength sector with and without the 50 Ω termination resistor for the 60 GHz to 65 GHz frequency range.

Sensors 2024 , 1 Figure 8 .
Figure 8. Microphotograph of the implemented TGA4600 low-noise amplifier (LNA) chip using rib bon bonding in (a).Typical ribbon bonding implementation in (b,c).Implementation imperfection in (d,e).The equivalent circuit model of ribbon bond in (f).

Figure 8 .
Figure 8. Microphotograph of the implemented TGA4600 low-noise amplifier (LNA) chip using ribbon bonding in (a).Typical ribbon bonding implementation in (b,c).Implementation imperfections in (d,e).The equivalent circuit model of ribbon bond in (f).Typical ribbon bonding implementation is shown in Figure8b,c.However, the possible scenarios of imperfections in the ribbon bond implementation are shown in Figures8d and 8e, respectively.The equivalent circuit model of the ribbon bond is shown in Figure 8f.It consists of resistance R, inductance L, and two distinct capacitors C 1 and C 2 .The resistance R and the inductance L are related to the parameters of the bonding ribbon itself.However, the equivalent capacitances C 1 and C 2 at the ends of the bonding ribbon are not identical since the two ends of the bonding ribbon are connected to different materials.To accurately estimate the equivalent circuit model parameters, a circuit co-simulation is performed using the simulation tools of Keysight ADS software.The optimized parameters of this circuit model are selected as follows: R = 0.0061 Ω, L = 38.31nH, C 1 = 140.82pF, and C 2 = 124.61pF.Figure9shows transmission loss as a function of frequency for a 1 mm ribbon bond length in different lateral deviation scenarios during ribbon bond implementation.It can be seen from the figure that in the case of a perfect implementation (∆W R = 0%), the losses start from 0.6 dB/mm at 60 GHz to achieve a maximum of about 1 dB/mm around 70 GHz.However, a horizontal deviation of 5% in ribbon bond implementation generates extra losses of about 0.1 dB at 60 GHz and 0.35 dB at 70 GHz compared to the perfect scenario in Figure8c.Overall, the transmission losses are more severe at high frequencies and significantly increase with the deviation tolerance to exhibit 1.15 dB/mm at 60 GHz and 2.4 dB/mm for a tolerance level of ∆W R = 15%.

Figure 9 .
Figure 9. Transmission loss as a function of frequency for a 1 mm ribbon bond at different lateral deviation tolerances during ribbon bond implementation.