Just as human beings acquire information about their environment through their senses and process such information using their brain, electronic systems perform such functions by means of sensors and processing digital devices such as microcontrollers (μC) or microprocessors (μP). Nowadays, such small but smart devices have become essential in many fields (industrial, automobiles, aircraft, medical devices, consumer electronics, home appliances, etc.) so that it is hard to imagine a society without them.

Figure 1(a) shows the classical block diagram of a sensor electronic interface [1]. First of all, the sensor transforms a signal from a given energy domain (such as thermal, magnetic, mechanical, chemical or radiant) to the electrical domain by changing—for example—its electrical resistance or capacitance. Afterwards, the signal conditioning circuit, which generally relies on operational amplifiers (OpAmp), performs some or all of the following tasks in the analogue domain: sensor output-to-voltage conversion, amplification, filtering, linearization and/or demodulation. The resulting analogue signal is then digitized via an analogue-to-digital converter (ADC). Finally, a digital system (e.g., a μC) acquires, stores, processes, controls, communicates (to other devices or systems) and/or displays the digital value with information about the measurand.

Nowadays, several blocks, as shown in Figure 1(a), can be embedded into the same integrated circuit (IC). There are commercially available ICs that integrate the processing digital system, the ADC and/or some signal conditioning circuit; for instance, MSC1210 from Texas Instruments (TI). There are also commercial ICs that integrate the sensor, its signal conditioning circuit and/or the ADC; for example, ADXL103 and ADXL312 accelerometers from Analog Devices; such chips are usually known as integrated smart sensors [2]. Other commercial ICs integrate the required signal conditioning circuit and the ADC to measure a specific type of sensor; for instance, ADS1232 from TI for bridge-type resistive sensors and AD7745 from Analog Devices for single and differential capacitive sensors.

For some sensors, however, the block diagram in Figure 1(a) can be simplified to that shown in Figure 1(b) [3], where the sensor is directly connected to the digital system without using either the signal conditioning circuit or the ADC; these circuits were initially proposed in application notes of μC’s manufacturers [4,5,6]. In such a circuit topology (so-called direct interface circuit), the μC appropriately excites the analogue sensor to get a signal (usually, a time-modulated signal) that can be directly measured in the digital domain, for instance, using an embedded digital timer; such output signals are also known as quasi digital since the analogue information (e.g., period or time interval) can be directly measured by a digital system. In comparison with the sensor electronic interface shown in Figure 1(a), a direct interface circuit is simpler and needs less components; actually, it just needs a common low-cost general-purpose 8-bit μC. Therefore, a direct interface circuit has advantages in terms of cost, physical space and power consumption, which is of major interest, for instance, in battery-powered measurement systems such as autonomous sensors. Furthermore, as will be shown along this paper, the performance of such circuits is quite remarkable taking into account their simplicity.

This paper reviews most of the work carried out about direct interface circuits and is organized as follows. Section 2 describes the operating principle of such circuits. Section 3 and Section 4 explain how these circuits can be used to measure different topologies of resistive and capacitive sensors, respectively, and then discuss their performance. Section 5 deals with the power consumption and proposes some design guidelines to reduce the current consumption. Finally, Section 6 takes some conclusions and forecasts the future research work about this topic.

Two measurement methods have been proposed so far to build the direct interface circuit shown in Figure 1(b):

Most of the research work about direct interfaces has been focused on those based on RC circuits, whose basics are explained by means of Figure 2.

If C is initially discharged and a step of amplitude V₁ is applied at the input, the output voltage is [see Figure 2(b)]: (1) and the time required to charge C from 0 to a given high threshold voltage (V_TH) is: (2) which is proportional to R and C. On the other hand, if C is already charged to V₁ and a step towards ground is applied at the input, the output voltage is [see Figure 2(c)]: (3) and the time needed to discharge C from V₁ to a given low threshold voltage (V_TL) is: (4) which again is proportional to R and C. According to (4), if C, V₁ and V_TL are known and T_d is measured, then the resistance of the RC circuit could be estimated by R = T_d/(Cln(V₁/V_TL)).

The RC circuit in Figure 2(a) can be directly connected to a μC using the circuit topologies shown in Figure 3(a) and Figure 4(a). In Figure 3(a), R has been replaced by R_x (i.e., a resistive sensor) and C by C_d (i.e., a capacitor whose nominal value is known), whereas in Figure 4(a), C has been replaced by C_x (i.e., a capacitive sensor) and R by R_d (i.e., a resistor whose nominal value is known). Pins 1 and P are two input/output digital port pins. Pin 1, which is in charge of monitoring the exponential charging or discharging voltage, usually includes a Schmitt trigger (ST) buffer (with two threshold voltages) and should be associated to an external interrupt or a capture module. The circuits in Figure 3(a) and Figure 4(a) can measure either the charging time or the discharging time of the RC circuit, but the measurement of the latter is more recommended since it has lower variability. This is because the discharging-time measurement uses the low threshold voltage (V_TL) of the ST buffer, which is less noisy [9] than the high threshold voltage (V_TH) used for the charging-time measurement.

The circuit in Figure 3(a), which is for the measurement of resistive sensors, involves two operation stages: charging stage, and discharging and measurement stage. During the charging stage [see Figure 3(b)], Pin 1 is set as an output providing a digital “1”, whereas Pin P is set as an input offering high impedance (HZ). Therefore, the capacitor C_d is quickly charged to the analogue output voltage (V₁) corresponding to a digital “1”, which is generally equal to the supply voltage (V_DD). During the discharging and measurement stage [see Figure 3(c)], Pin 1 is set as a HZ input and Pin P is set as an output providing a digital “0” and, consequently, C_d is discharged towards ground through R_x while the embedded timer measures the time interval required to do so; such a measurement is carried out using a high-frequency oscillator (with a period T_S) as a reference. When the exponential discharging voltage crosses V_TL, the timer is read and a digital number proportional to R_x [see Equation (4)] is achieved. The resulting waveform of the voltage across C_d is similar to that shown in Figure 2(c).

For the measurement of a capacitive sensor (C_x), it is recommended to swap the position of the resistance and the capacitance, as shown in Figure 4(a), but again two operation stages are required. During the charging stage [see Figure 4(b)], both pins are set as an output: Pin 1 provides a digital “1”, whereas Pin P provides a digital “0”. Therefore, C_x is rapidly charged to V₁. During the discharging and measurement stage [see Figure 4(c)], Pin 1 is set as a HZ input and Pin P does not change its state and, consequently, C_x is discharged towards ground through the resistor R_d while the timer is running. When the voltage-threshold crossing is detected [see Figure 2(c)], the timer is read and a digital number proportional to C_x [see Equation (4)] is achieved.

The operating principle explained in Figure 3 can be applied to measure different topologies of resistive sensor. Before discussing the proposed circuit for each sensor topology, following are a few general remarks on the components required:

Different topologies of capacitive sensor can be directly measured by a μC using the operating principle explained in Figure 4. Again, before discussing the proposed circuits, following are a few general remarks on the components required:

For those applications in which the sensor is not read continuously but periodically (i.e., every T_T seconds), the average current consumption of a direct interface circuit is (13) where I_act is the average current consumption in active mode (i.e., when the μC is working to perform the measurement), T_act is the time interval required to complete one measurement in active mode, N is the number of times that the measurement is repeated in active mode to average N measurements, I_sleep is the current consumption of the μC in sleep (or standby) mode, T_sleep is the time interval in which the μC is in sleep mode, and T_T = N·T_act + T_sleep. The value of I_sleep specifically depends on the low-power features of the μC, but it is generally smaller than 1 μA (even it can be smaller than 100 nA). On the other hand, the value of I_act clearly depends on the configuration of the resources embedded into the μC during the measurement. To reduce the current consumption, it is advisable to set the CPU (Central Processing Unit) and the timer as indicated in Table 10 [29]; the rest of peripherals should be switched off. In the charging stage, the charging time should be controlled by the CPU running at low frequency (e.g., tens or hundreds of kHz); two remarks are as follows: (i) most of the current μCs have a prescaler to divide the master clock frequency, and (ii) a low-accuracy charging time (due to a low-frequency clock) is not a problem at all since the information is in the discharging time. In the discharging stage, the discharging time should be measured by the timer running at high frequency (e.g., units or tens of MHz) to have a good resolution, whereas the CPU should be off whenever this does not stop the operation of the interrupt system and the timer. In the processing stage, the CPU should run at high frequency (e.g., units or tens of MHz) to compute as fast as possible the resistance or capacitance of the sensor by means of, for example, (5) or (8).

The average current consumption in active mode in measurements involving three charging stages, three discharging stages and one processing stage (as happens, for example, in the direct interface circuits shown in Figure 5(a) and Figure 6(a)) can be estimated by [29] (14) where C is the capacitance being charged [i.e., C_d in Figure 5(a) and C_x in Figure 6(a)], T_c is the time interval required to charge C, T_d is the discharging time to be measured, T_p is the time interval required to process the discharging times, and I_int1, I_int2 and I_int3 are the average current consumed by the internal electronics of the μC during the charging, discharging and processing stages, respectively. Note that in Equation (14) T_d is assumed to be the same for the three measurements; this assumption does not make sense in terms of the result obtained from (5) or (8), but it does in terms of current consumption analysis. According to (14), the current consumed by the internal electronics of the μC plays an important role on the overall current consumption. Furthermore, the value of I_int1, I_int2 and I_int3 clearly depends on the supply voltage (V_DD) and operating frequency (f_clk) of the μC. Table 11 shows, for example, some experimental values of such currents for an AVR μC running at different operating conditions when the configuration proposed in Table 10 is applied.

Direct interface circuits for capacitive sensors are expected to be less power demanding than those for resistive sensors. This is because the first two components in Equation (14), which correspond to the charging stages, are almost negligible when measuring capacitive sensors in the picofarad range or smaller. Actually, the current consumption in active mode of direct interface circuits for capacitive sensors is equal, in a first approximation, to I_int2. Experimental tests carried out at 3 V – 4 MHz show that the current consumption when measuring capacitances (177 pF) is about 2–3 smaller than that when measuring resistances (1 kΩ), as shown in Figure 7. In terms of measuring time and energy, a complete measurement of a 1-kΩ resistive sensor requires 1.6 ms and 7 μJ for C_d = 220 nF, and 5.8 ms and 27 μJ for C_d = 1 μF.

Direct interface circuits are able to measure different topologies of resistive and capacitive sensors just using a common low-cost general-purpose 8-bit μC with an embedded digital timer but without any on-chip ADC, OpAmp or analogue comparator. In spite of their simplicity and low cost, these circuits perform remarkably and therefore they could be very attractive for medium-accuracy, medium-resolution applications. Note, however, that such a satisfactory performance is to be expected when measuring resistive sensors in the kiloohm range and capacitive sensors in the picofarad range. The measurement of low-value resistive sensors (say, lower than 100 Ω) is feasible, but the discharging-time measurement would suffer from a significant offset effect due to R_s, whereas the measurement of high-value resistive sensors (say, higher than 10 MΩ) could be affected by the parasitic resistance of the input port pins of the μC. On the other hand, the measurement of capacitive sensors with a nominal capacitance smaller than 1 pF seems impracticable because of the high effects of the parasitic capacitances of the input port pins of the μC. Direct interface circuits implemented with different commercial μCs from different manufacturers show similar results and, therefore, the design of such circuits does not depend on any specific device or integrated circuit from any manufacturer. A measuring time of about units or tens of millisecond can be a limitation of such circuits if the quantity to be measured changes quite fast.

The field of direct interface circuits is still under research and many interesting ideas could be developed in the near future. From the author’s point of view, future work on direct interface circuits could be focused on the following two directions: